MOTOROLA Order this document by AN1669/D SEMICONDUCTOR APPLICATION NOTE AN1669 MC44603 in a 110 W Output SMPS Application (80-140 Vrms and 180-280 Vrms Mains Voltages) by Joel Turchi Power Management Products Operation Application Laboratory, Motorola, Toulouse, France The purpose of this application note is to present a way of designing the MC44603 in a fly-back dedicated to a 110 W output power supply in two distinct cases: 110 Vrms mains and 220 Vrms mains. For this range of power, the discontinuous mode must be chosen as it limits the stress on the power switch and the output diodes. This kind of working can be guaranteed, thanks to the demagnetization arrangement of the MC44603. This application note considers both high and low mains voltages. -- In the high voltage a.c. line case, it deals with both MOSFET and BIPOLAR transistor use. -- In the low mains voltage case, only the MOSFET solution is considered as the inductor peak current is high. The MC44603 is a high flexibility SMPS controller. It offers a wide variety of protection (accurate maximum current limitation, Vcc overvoltage detection, fold-back, demagnetization, soft start). In addition, distinct working modes can be used with this circuit: -- a fixed frequency working mode Here, the frequency is given by the circuit oscillator. -- a variable frequency working mode This kind of working can be obtained when the chosen oscillator period is shorter than the cycle time for transformer magnetization and complete demagnetization. -- a stand-by mode This reduces the working frequency during this phase. This method reduces the stand-by losses that are mainly proportional to the switching frequency, and avoids the unstable very high frequency working that is generally associated with traditional free frequency systems. In the MC44603 design proposed here, the variable frequency mode is not used. Indeed, as shown later, the fixed frequency mode enables a more accurate control of the power that is drawn from the mains. I -- THE APPLICATIONS OUTPUT: 110 W In the following calculations, the maximum input power will be considered equal to 135 W ( 80%). [ Application 1: 110 V mains 80 Vrms 140 Vrms The outputs: 120 V 28 V 15 V 8V 0.5 A 1.0 A 1.0 A 1.0 A Consequences for the Transformer This output voltage values are obtained using four secondary windings of the transformer. Now, as a diode is located between the output capacitor and the winding, this diode voltage must be taken into account in the choice of the number of turns of each of the windings. So, if the diode voltage is considered equal to 1 V, the desired voltages on the windings are nearly: 121 V 29 V 16 V 9V To obtain the output voltage values of the specification with a good accuracy, we need at least 3 turns for the 9 V and then: 9V 3 turns 16 V 5 turns 29 V 10 turns 121 V 40 turns This solution seems to be correct, since with three turns, the 9 V should be well coupled; also, 40 turns (for the 120 V) is not too large a value (no ferrite saturation). In our application, in order to obtain a very well regulated high voltage output, an opto coupler is used. By this means, the regulation uses feedback drawn from the 120 V output. II -- GENERAL DESIGN EQUATIONS Calculation of the Main SMPS Parameters II-1 -- Fixed frequency working mode: To obtain a fixed frequency discontinuous working mode (the discontinuous mode is guaranteed by the demagnetization section), the transformer primary inductor magnetization and demagnetization cycle must be shorter than the oscillator period (that is the chosen working period) for any working point. This condition can be expressed by the following inequality: Ton ) Toff v Tosc (ineq 1) Application 2: European mains 180 Vrms 280 Vrms Motorola Data Motorola, Applications Inc. 1998 1 AN1669 where: Ton is the on-time, Toff is the off-time II-3 -- Power switch on-time losses Tosc is the MC44603 oscillator period II-3-1 -- MOSFET: The current increases linearly during the on-time and decreases linearly in the inductor for the off-time. + VinL Ipk + NVo L Ipk So, and, where: Ton Rdson being the MOSFET on-time resistor, the on-time losses can be calculated from: (eqn 1) Pon Toff (eqn 2) Ipk is the primary inductor peak current where Vin is the rectified a.c. line voltage (fly-back input voltage) So, as: L is the primary inductor value NVo is the output voltage considered in the primary side (N: turn ratio) ) Toff + L ) Vin NVo (eqn 3) Vin NVo In addition, the energy drawn through the transformer during one cycle is: So, Ton E + 12 So, Pin (input power) is: 1 Pin 2 + where Ipk Ipk 2 L condition required to work in a fixed frequency mode: v2 1 Pin Vin Vin ) NVo NVo + 13 Pon where 2 (ineq 2) Iin + 12 (eqn 7) d is the duty cycle Ton Now, +d + 12 Ipk + Ipk 2 2 L fosc Vin 2 + (Pin)max (Vin)min Rdson where (eqn 16) (Pin)max L fosc dmax is the maximum duty cycle. II-3-2 -- BIPOLAR TRANSISTOR + Ton 1 Tosc V CE I T dt + VCE IT Pon + V Iin CE Pon So: where (eqn 17) (eqn 18) (eqn 19) Iin is the input current Pon + VCE Pin Vin (eqn 20) Pin fosc maximum BIPOLAR transistor on-time losses (eqn 9) (Pon)max (eqn 10) Thus, Peak inductor current: (Ipk)max (eqn 15) Consequently, L Using equations 6 and 9: 3 (eqn 8) Using equations 1, 7 and 8: Iin 2 +2 So: Tosc Pin fosc 0 (eqn 6) d L maximum MOSFET on-time losses Thus: Ipk Pin Vin Rdson Consequently, Pon Iin is the input current and where Iin (eqn 13) Ton 3 (eqn 14) (Tosc L2 ) Vin 2 Rdson + 2 3 2 Pin is given by the following expression: + Vin 0 Consequently, using equations 1, 5 and 14, the following expression can be written: II-2 -- Peak inductor current expression Pin (eqn 12) + Using equations 3 and 5 and inequality 1, fosc dt Using the (Ipk)max value, the following equation could also be written: 1 Rdson (Ipk)max 2 dmax (Pon)max 3 fosc is the MC44603 oscillator frequency L I 2 T Rdson + Pon (eqn 5) fosc IT is the MOSFET current. t I Vin T L (Pon)max Ipk 2 L (eqn 4) + Ton 1 Tosc + VCE L (Pin)max fosc (eqn 21) Ton + Tosc (eqn 22) II-4 -- Maximum duty cycle The duty cycle, d, is equal to: 2 (Pin)max (Vin)min d (eqn 11) Motorola Applications Data Ton Now, d so, AN1669 +L +L Ipk Vin Ipk Vin fosc N is the transformer turn ratio between the 120 V output winding and the primary inductor (eqn 23) (eqn 24) (VD )max: the maximum voltage the 120 V output diode must face: Thus, using equation 10: d + 2 (V )max D Pin L fosc Vin 2 (eqn 25) So, maximum duty cycle: dmax + 2 (Pin)max L fosc (Vin)min 2 (eqn 26) (VT )max: the maximum voltage the power switch must face: + (2 where: (Vin)max) ) (N 120) (V) (eqn 27) (Vin)max is the maximum rms a.c. line voltage (140 V or 280 V according to the line) ) 120 (V) (eqn 28) To avoid any risk of saturation in the transformer, the inductor peak current must be lower than (ni/ np), where np is the turns number of the primary inductor. np where The other parameters that must be taken into account are: (Vin)max N (ni): the transformer ferrite saturation parameter Now, II-5 -- Other design parameters (V T)max + 2 + N x n120 V (eqn 29) n120V is the turns number of the 120 V winding ni N x n x Ipk (eqn 30) So, 120V These parameters are the main elements that have to be taken into account as they allow you to choose the power switch, the diodes, the transformer and the working frequency. As shown by the preceding calculations, the design parameters depend on some elements like N or the value of (fosc x L). + 120 is the maximum output voltage SUMMARY Condition Required to Work in a Fixed Frequency Mode L Maximum Peak Inductor Current Maximum Power Mosfet On-Time Losses Motorola Applications Data v2 (Pon)max dmax + 13 Vin Vin 1 Pin + (Ipk)max Maximum Bipolar Transistor On-Time Losses Maximum Duty Cycle fosc 2 L Rdson (Pon)max + VCE + (Pin)max 2 ) NVo NVo 2 (Pin)max fosc (Ipk)max 2 dmax (Pin)max (Vin)min L fosc (Vin)min 2 3 AN1669 that would cut the voltage spikes due to the leakage inductor (refer to Figure 1). Consequently, N must be chosen lower than 1.25 (VTmax = 350 V). III -- APPLICATION 1: 110 V INPUT III-1 -- Choice of the transformer One way to use the above design equations, consists of drawing up a table showing how the main SMPS parameters vary with the value of the turn ratio. To calculate these values, it is necessary to know the input power level. This value is taken equal to (135 W) in our application (135 W corresponds to an efficiency equal to about 80%. The application results will show that this assumption ensures a desirable margin with the nominal input voltage). On the other hand, the parameters calculation shows that (L x fosc)max is the (L x fosc) value that results in the lowest (Ipk)max and (Pon)max ones (refer to Ipk or Pon expressions). This (L x fosc) value is the maximum one that guarantees a fixed frequency working for any working point (refer to section II-1). The SMPS parameters given in the following table are calculated using this threshold value. Choice criteria and definition of the transformer: As shown by the following table, the higher the turn ratio (N) is, the lower the peak current is. Now, the (ni)max is proportional to N and the voltage the transistor must face, increases when N rises. That is why an optimal N value must be chosen. In fact, there are three main choice criteria: -- the peak current and the on-time losses. N must be as large as possible in order to reduce the peak current and the on-time losses -- the voltage the power MOSFET must face. Indeed, this voltage must be as low as possible to reduce its cost and in order to decrease the Rdson. That is why a MOSFET 400 V should be used. It is necessary to have a safety voltage margin, to avoid the need to incorporate a lossy and costly clamping network Finally, in order to use a ferrite (ni = 180, AL = 250 nH/ turns2), (N = 0.75) seems to be a suitable value. Indeed, this value should result in a well coupled transformer with a low leakage inductor value. On the other hand, if we do not take into account the turning off spikes, then the theoretical highest value the power MOSFET must face, is 290 V. Consequently, with a 400 V power switch, only a low loss clamping arrangement is required. The chosen MOSFET, is the MTP10N40E (Rdson = 0.55 W, 400 V). Consequently, Lp + AL x (N x 40)2 Lp 225 H and the optimal working frequency is: L p x fosc 9.3 fosc 41.3 kHz So, the following values can be chosen: Lp = 225 H fosc 40 kHz and then: (Rref = 10 k, CT = 1nF) Ipk 5.4A N (L.fosc)max (Ipk)max (A) (VT)max (V) (VD)max (V) MOSFET on losses/ Rdson (W/W) (ni)max 0.50 5.6 6.9 260 520 5.7 139 0.75 9.3 5.4 290 390 4.3 162 0.90 11.0 5.0 300 340 4.1 180 1.00 12.5 4.6 320 320 3.7 184 1.25 14.9 4.3 350 280 3.5 215 1.50 17.3 4.0 380 250 3.2 240 21.9 3.5 440 220 2.8 281 2.00 NOTE: 4 -- the transformer must be well coupled. This is to obtain a consistently accurate regulation of the output and to reduce the leakage inductor and hence the turning off spikes (refer to Figure 1). That is why a low air-gap ferrite must be used. Practically, a ferrite whose (ni) is lower than 200 A.turns, seems to be a good choice. So, N must be lower than 1. (The appendix gives details of OREGA transformers; the SMT4 suits our application). N: turn ratio (refer to II-5) (VT)max: maximum voltage the power switch must face (VD)max: maximum voltage the 120 V output diode must face Motorola Applications Data AN1669 TURNING OFF SPIKES VT VIN + NVO VIN TIME ON-TIME OFF-TIME DEAD-TIME Figure 1. Voltage Spikes Due to the Leakage Inductor III-2 -- MC44603 pins use: (refer to the application schematics) 3 -- Foldback (pin 5): 1 -- Vcc (PIN 1): The pin Vcc must be connected to a transformer auxilliary winding. This extra winding turns number can be taken equal to 5 in order to obtain a Vcc nearly equal to 15 V. 2 -- Vc and OUTPUT (pins 2 and 3): Part of Vcc must be applied to this pin thanks to a resistor divider. This voltage value must be slightly higher than 1 V in normal use, so that this value drops below this threshold value as soon as an overload occurs. 4 -- Overvoltage protection (pin 6): Vc is the output high state of the circuit. This pin offers the possibility of setting the output source current at a different level than the sink current but it is no use in our case. In fact, a resistor of 33.2 W must be connected between the output and the MOSFET gate to make the switchings smoother. A resistor of about 1 kW can be connected between the gate and the ground (or the current sense external resistor) to avoid any inadvertent MOSFET switching on due to noise (see Figure 2). This pin can remain free and then, the Vcc threshold level is fixed equal to nearly 17 V. On the other hand, to make detection quicker and more accurate, an external resistor divider can be used with a diode and an integration capacitor (refer to the proposed application). The resistor divider is not directly connected to the Vcc because Vcc has a high time constant (refer to the application schematic -- Figure 4). 5 -- Current sense (pin 7): 1 2 VCC The current sense resistor must be designed in order to limit the current below the maximum peak calculated in section II in order to limit the power that the converter is able to draw from the mains; in a fixed frequency mode, Pin = 1/2 x L x Ipk2 x fosc. VC output 3 33.2W Now, 1 kW Rs (Ipk)max = 5.4A In addition, the (Vcs) clamp level is nearly 1 V (refer to the data sheet). So, (Rs) the current sense resistor, must be equal to (1 V / 5.4A), that is nearly: 0.18 . Figure 2. Motorola Applications Data This value can be obtained using a 1 W, 0.2 resistor and a resistor divider (442 , 3.16 k) (refer to Figure 3). 5 AN1669 Even if capacitors have discrete values, the choice of Rref allows you to fix precisely the oscillator frequency (however, Rref also fixes the internal current source (Iref), which must be lower than 500 A and higher than 100 A). 7 fosc 442W 3.16kW 0.2W Rs Figure 3. Finally, as the fixed frequency mode is obtained for any working point, the peak current limitation results in an accurate input power limitation (135 W in this application note). 6 -- Oscillator (pins 10 and 16): The oscillator frequency is determined by the couple (CT, Rref) (refer to the data sheet). 6 + 40 kHz + 10 k + 1nF Rref C T 7 -- Stand-by mode (pins 12 and 15): In the MC44603, it is possible to reduce the working frequency when little power is being drawn from the mains (stand-by mode). This stand-by frequency is fixed by connecting a resistor RFstby to pin 15, while the power level at which the stand-by mode must be applied is determined by connecting another resistor RPstby to pin 12 (this power level is labelled PthL in the data sheet). In the data sheet, the equations needed to calculate RFstby and RPstby are indicated. Using them, to obtain a power level equal to 10 W and a stand-by frequency equal to 20 kHz, the calculated RFstby and RPstby values are: RPstby = 8.45 k and RFstby = 22.1 k Motorola Applications Data AN1669 80 VAC TO 140 VAC RFI FILTER C3 1 nF / 1 kV R1 1W/5W R3 4.7 kW C4-C7 1 nF / 500 V D1-D4 MR504 C32 C1 220 mF 250 V 120 V / 0.5 A 47 kW D8 MR856 D5 1N4934 C2 220 mF 25 V R2 22 kW / 2 W 9 1 nF C10 1 mF 10 7 11 6 12 R15 8.45 kW 13 C11 1 nF MC44603P C9 C16 100 pF L1 1 mH R7 15 kW 28 V / 1 A D9 MR852 R5 1.21 kW R16 10 kW 3 R6 200 W 15 2 16 1 R10 33.2 W R9 442 W C26 R18 22.1 kW R19 10 kW C13 100 nF 3.16 kW 220 pF D10 MR852 R26 1 kW C25 1000 mF R14 0.2 W 8V/1A D11 MR852 R24 270 W MOC8101 R21 10 kW C12 10 nF C24 0.1 mF 220 pF C21 1000 mF R25 1 kW C28 0.1 mF 15 V / 1 A C23 R17 10 kW C27 1000 mF LP C14 4.7 nF MTP10N40E 14 C31 0.1 mF 220 pF Laux R8 1 kW 4 C29 1N4937 D6 1N4148 C15 1 nF 5 C30 100 mF 47 nF R4 27 kW 8 220 pF C19 100 nF C22 0.1 mF R23 117.5 kW D14 1N4733 C20 33 nF TL431 R22 2.5 kW Figure 4. 110 W Output Off-Line Flyback Converter with MOSFET Switch, 80 V 140 V Mains Voltage Motorola Applications Data 7 AN1669 Table 1. 110 W Fly-Back Converter, 80 Vrms-140 Vrms Mains Range, MC44603 and MTP10N40E Test Conditions Line Regulation Results Vin = 90 Vac to 140 Vac Fmains = 50 HZ Fmains = 50 HZ 120 V 28 V Iout = 1A 15 V Iout = 1A 8V Iout = 1A Load Regulation D=0V D=0V D=0V D=0V Iout = 0.5A Vin = 110 Vac 120 V Cross Regulation D = 0.05 V Iout = 0.3A to 0.5A Vin = 110 Vac Iout (120 V) = 0.5A Iout (28 V) = 0A to 1A 120 V D=0V Iout (15 V) = 1A Iout (8 V ) = 1A Efficiency Vin = 110 Vac, Po = 110 W 84.5% Vin = 110 Vac, Pout = 0 W 1.2 W Standby Mode P input Switch. freq. 20 KHz fully stable Output short circuit Safe on all outputs Start-up Pin 110 W Vac = 80 V MOSFET application: information about the transformer Lp5 Lp5 110 W Lp (turns) 30 Laux (turns) 5 L1 (turns) 40 L2 (turns) 10 L3 (turns) 5 L4 (turns) 3 Al (nH/turns2) 274 Core E-4215A Material B2 Former specific Thomson design Wire size (mm2) 0.315 all windings Flyback transformer construction For cost reduction and simplicity, all windings have the same size. For optimal Lp/Laux coupling, Laux is wound on the second section of Lp. Former The normalized primary/secondary isolation is obtained using the multi-slotted former depicted on the figure. This former uses designs patented by LCC Thomson. L14 L14 Lp4 Lp4 L13 L12 L13 Lp3 Lp3 L42 L42 Lp2 L11 L12 Lp2 L41 L41 Lp1 Lp1 L11 SMT4 Lp Primary Winding (Lp1//Lp3//Lp5) + (Lp2//Lp4) Laux Auxilliary Winding L1 High Voltage Secondary Winding L11//L12//(L13 + L14) L2 Secondary Winding (28 V) (2 X 10 turns) L3 Secondary Winding (15 V) (2 X 5 turns) L4 Secondary Winding (8 V) L41//L42 8 Motorola Applications Data AN1669 and the optimal working frequency is: IV -- APPLICATION 2: 220 V INPUT VOLTAGE L x fosc 24.3 IV-1 -- Choice of the transformer: fosc 55 kHz So, the following values can be chosen: One way to use the above design equations, consists of drawing up a table showing how the main SMPS parameters vary with the value of the turn ratio. To calculate these values, it is necessary to know the input power level. This value is taken equal to (135 W) in our application (135 W corresponds to an efficiency equal to about 80%. The application results will show that this assumption ensures a desirable margin with the nominal input voltage). On the other hand, the parameters calculation shows that (L x fosc)max is the (L x fosc) value that results in the lowest (Ipk)max and (pon)max ones (refer to Ipk or pon expressions). This (L x fosc) value is the maximum one that guarantees a fixed frequency working for any working point (refer to section II-1). The SMPS parameters given in the following table are calculated using this threshold value. L = 438 H fosc = 50 kHz and then Ipk = 3.5A BIPOLAR transistor case: As the gain of a Bipolar transistor decreases when the collector current level rises, the SMPS peak current must be as low as possible. That is why N must be chosen as high as possible. Now, if classical BIPOLAR transistors are able to face 1000 V or 1200 V, their VCEO is generally low. The transistor used in the application, the MJE18206, has a VCES equal to 1200 V and a VCEO equal to 600 V. Since there are damped oscillations (converging to Vin) during the dead- time (refer to Figure 1), the transistor may be turned on while its VCE voltage is higher than Vin (the maximum Vin value being nearly equal to 400 V). That is why, even if a resistor is connected between the base and the emitter of the transistor (refer to section IV-2), the (VT)max (that is, (Vin+NVo)max) must be chosen lower than 600 V, to ensure system reliability. In addition to this, a second choice criterion is (ni)max, since transformer saturation must be avoided. (N = 1.6) seems to be a good choice that enables the use of a ferrite (AL = 250nH/turns2; ni = 180)) Consequently, Choice criteria and definition of the transformer: As shown by the following table, the higher the turn ratio (N) is, the lower the peak current is. Now, the (ni)max is proportional to N and the voltage the transistor must face, increases when N rises. That is why an optimal N value must be chosen. MOSFET case: To perform a low cost SMPS, it is required to use a MOSFET 600 V. It is necessary to have a safety voltage margin, to avoid the need to incorporate a lossy and costly clamping network that would cut the voltage spikes due to the leakage inductor at the power switch turning off (refer to Figure 1 in section III-1). Practically, about 550 V is acceptable. Consequently, (N = 1.2) seems to be a maximum value. Now, in order to obtain a well coupled transformer with a low leakage inductor value, it is desirable to use a ferrite with a low air-gap. So, in order to be able to use a ferrite (ni = 140, AL = 274 nH/ turns2), (N = 1) seems to be a preferable value. L + AL x (N x 40)2 L 1mH So, the optimal working frequency is: L x fosc 43.7 fosc 43 kHz Finally, the following value can be taken: L = 1mH fosc = 43 kHz (Ipk)max = 2.5A Consequently, Lp + AL x (N x 40)2 Lp 438 H (VT)max (V) (VD)max (V) MOSFET on losses/Rdson (W/W) BIPOLAR on losses/VCE (W/A) (ni)max N (L.fosc)max (Ipk)max (A) 0.75 16.2 4.1 490 650 1.5 0.54 122 1.00 24.3 3.3 520 520 1.2 0.54 133 1.20 30.9 3.0 540 450 1.1 0.54 144 1.40 37.4 2.7 570 400 1.0 0.54 150 1.60 43.7 2.5 590 370 0.9 0.54 159 1.80 49.7 2.3 620 340 0.8 0.54 168 55.5 2.2 640 320 0.8 0.54 176 2.00 NOTE: N: turn ratio (refer to II-5) (VT)max: maximum voltage the power switch must face (VD)max: maximum voltage the 120 V output diode must face Motorola Applications Data 9 AN1669 r1 = 22 W r2 = 4.7 W IV-2 -- MC44603 pins use: (refer to the application schematics) As (Vcc 15 V), the obtained base currents are: 1 -- Vcc (PIN 1): IB1 410 mA IB2 850 mA The pin Vcc must be connected to a transformer auxilliary winding. This extra winding turns number can be chosen equal to 5, in order to obtain a Vcc nearly equal to 15 V. These base currents enable a correct transistor drive. 2 -- Vc and OUTPUT (pins 2 and 3): 3 -- Foldback (pin 5): Vc is the output high state of the circuit. This pin offers the possibility of setting the output source current at a different level than the sink current. A portion of Vcc must be applied to this pin thanks to a resistor divider. This voltage value must be slightly higher than 1 V in normal working so that this value drops below this threshold value as soon as an overload occurs. -- MOSFET case: A resistor of 10 must be connected between the output and the MOSFET gate to make the switchings smoother. A resistor of about 1 k can be connected between the gate and the ground (or the current sense external resistor) to avoid any inadvertent MOSFET switching on due to noise. 4 -- Overvoltage protection (pin 6): For the on-time, a bipolar transistor requires a base current labelled IB1, that must be higher than: (Ic)max / min This pin can remain free and then, the Vcc threshold level is fixed equal to nearly 17 V. On the other hand, to make detection quicker and more accurate, an external resistor divider can be used with a diode and an integration capacitor. In the proposed application, this resistor divider is not directly connected to the Vcc because Vcc has a high time constant (refer to the application schematics). where 5 -- Current sense (pin 7): -- BIPOLAR transistor case: (Ic)max is the maximum collector current (that is Ipkmax if the current sense resistor is well designed -- refer to section IV-5), and min is the minimum guaranteed transistor gain for (Ic = (Ic)max) Now, with the MJE18206: (min 7) for (Ic)max = 2.5A So, (IB1 = 400mA ) is a good value that ensures a safety margin. On the other hand, the turn off base current peak must be nearly equal to (2 x IB1). The couple (Dz,Cz) is used to build a voltage source Vz (during the on-time), able to produce IB2. So, IB1 = (Vcc - Vz - Vbe) / (r1 + r2) IB2 = (Vz + Vbe) / r2 Consequently, using Vz = 3.3 V Cz = 1 F 1 2 The current sense resistor must be designed in order to limit the current down to the maximum peak calculated in section II in order to limit the power the converter is able to draw from the mains (in a fixed frequency mode, Pin = 1/ 2 x L x Ipk2 x fosc). -- MOSFET case: (Ipk)max = 3.5A Now, the (Vcs) clamp level is nearly 1 V (refer to the data sheet). So, (Rs) the current sense resistor, must be equal to (1 V / 3.5A), that is nearly: 0.28W (2 x 0.56 in parallel). -- BIPOLAR case: (Ipk)max = 2.5A So, (Rs) must be equal to (1 V / 2.5A), that is: 0.4 W (3 x 1.2 W in parallel). Finally, as the fixed frequency mode is ensured for any working point, the peak current limitation results in an accurate input power limitation (135 W in this application). VCC 1 VC 2 output 3 10W VCC VC r1 22W Dz (3.3V) output 3 1 kW r2 4.7W Rs MOSFET drive Cz 1 F 47W Rs BIPOLAR transistor drive Figure 5. 10 Motorola Applications Data AN1669 6 -- Oscillator (pin 10 & 16): The oscillator frequency is determined by the couple (CT, Rref) (refer to the data sheet). As capacitors have discrete values, the choice of Rref allows you to fix precisely the oscillator frequency (however, Rref also fixes the internal current source (Iref), which must be lower than 500 A and higher than 100 A). MOSFET case: fosc = 50 kHz Rref = 10 k CT = 820pF BIPOLAR case: fosc = 43 kHz Rref = 10 k CT = 1nF This stand-by frequency is fixed by connecting a resistor RFstby to pin 15, while the power level at which the stand-by mode must be applied is determined by connecting another resistor RPstby to pin 12 (this power level is labelled PthL in the data sheet). In the data sheet, the equations needed to calculate RFstby and RPstby are indicated. Using them, to obtain a power level equal to 15 W and a stand-by frequency equal to 20 kHz, the calculated RFstby and RPstby values are: MOSFET case: RPstby = 10 k RFstby = 27 k BIPOLAR case: RPstby = 10 k RFstby = 22 k 7 -- Stand-by mode (pins 12 and 15): In the MC44603, it is possible to reduce the working frequency when little power is being drawn from the mains (stand-by mode). Motorola Applications Data 11 AN1669 180 VAC TO 280 VAC C3 1 nF / 1 KV RFI FILTER R1 1W/5W R3 4.7 kW C4-C7 1 nF / 1000 V D1-D4 1N4007 C32 C1 100 mF R20 22 kW 5W D5 1N4934 C2 220 mF R2 68 kW / 2 W 9 C10 820 pF 1 mF 10 7 11 6 12 R15 10 kW 13 C11 1 nF MC44603P C9 R9 C16 100 pF 1 kW L1 1 mH R7 180 kW R8 15 kW 4 D8 MR856 C30 100 mF D7 MR856 C29 28 V / 1 A D9 MR852 R5 1.2 kW R16 10 kW 3 15 2 16 1 R10 10 W R6 180 W C26 R18 27 kW R19 10 kW C13 100 nF 220 pF D10 MR852 R26 1 kW C25 1000 mF R14 2 X 0.56 W// 8V/1A D11 MR852 R24 270 W MOC8101 R21 10 kW C12 6.8 nF C24 0.1 mF 220 pF C21 1000 mF R25 1 kW C28 0.1 mF 15 V / 1 A C23 R17 10 kW C27 1000 mF LP C14 4.7 nF MTP6N60E 14 C31 0.1 mF 220 pF Laux D6 1N4148 C15 1 nF 5 120 V / 0.5 A C17 47 nF R4 27 kW 8 220 pF C19 100 nF C22 0.1 mF R23 117.5 kW D14 1N4733 C20 33 nF TL431 R22 2.5 kW Figure 6. 110 W Output Off-Line Flyback Converter with MOSFET Switch. 180 V-280 V MAINS RANGE 12 Motorola Applications Data AN1669 Table 2. 110 W Fly-Back Converter, 180 V-280 V Mains Range, MC44603 and MTP6N60E Test Conditions Line Regulation Results Vin = 180 Vac to 280 Vac Fmains = 50 HZ 120 V 28 V Iout = 1A 15 V Iout = 1A 8V Iout = 1A Load Regulation D=0V D=0V D=0V D=0V Iout = 0.5A Vin = 220 Vac 155 V Cross Regulation D = 0.05 V Iout = 0.3A to 0.5A Vin = 220 Vac Iout (120 V) = 0.5A Iout (28 V) = 0A to 1A 120 V D=0V Iout (15 V) = 1A Iout (8 V ) = 1A Efficiency Vin = 220 Vac, Po = 110 W 84% Vin = 220 Vac, Pout = 0 W 3W Standby Mode P input Switch. freq. 20 KHz fully stable Output short circuit Safe on all outputs Start-up Pin 110 W Vac = 160 V MOSFET application: information about the transformer Lp5 Lp5 110 W Lp (turns) 40 Laux (turns) 5 L1 (turns) 40 L2 (turns) 10 L3 (turns) 5 L4 (turns) 3 Al (nH/turns2) 274 Core E-4215A Material B2 Former specific Thomson design Wire size (mm2) 0.315 all windings Flyback transformer construction For cost reduction and simplicity, all windings have the same size. For optimal Lp/Laux coupling, Laux is wound on the second section of Lp. Former The normalized primary/secondary isolation is obtained using the multi-slotted former depicted on the figure. This former uses designs patented by LCC Thomson. L14 L14 Lp4 Lp4 L13 L12 L13 Lp3 Lp3 L42 L42 Lp2 L11 L12 Lp2 L41 L41 Lp1 Lp1 L11 SMT4 Lp Primary Winding (Lp1//Lp3//Lp5) + (Lp2//Lp4) Laux Auxilliary Winding L1 High Voltage Secondary Winding L11//L12//(L13 + L14) L2 Secondary Winding (28 V) (2 X 10 turns) L3 Secondary Winding (15 V) (2 X 5 turns) L4 Secondary Winding (8 V) L41//L42 Motorola Applications Data 13 AN1669 180 VAC TO 280 VAC C3 1 nF / 1 KV RFI FILTER R1 1W/5W R3 4.7 kW C4-C7 1 nF / 1000 V D1-D4 1N4007 C32 120 V / 0.5 A D5 1N493 4 C2 220 mF R2 68 kW / 2 W 9 1 nF C10 1 mF 10 7 11 6 12 R15 10 kW 13 C11 1 nF 14 R16 10 kW 15 16 1 L1 1 mH 28 V / 1 A D9 MR852 R5 1.2 kW R18 22 kW R19 10 kW R8 15 kW R6 180 W D13 1N4728 C26 C34 1 mF 47 W R14 3 X 1.2 W// 220 pF D10 MR852 C25 1000 mF D12 MR856 R13 1 kW 8V/1A D11 MR852 R24 270 W MOC8101 R21 10 kW C12 6.8 nF C24 0.1 mF 220 pF C21 1000 mF R25 1 kW C28 0.1 mF 15 V / 1 A C18 2.2 nF MJF18206 R26 4.7 W C13 100 nF C27 1000 mF LP C14 4.7 nF C23 R17 10 kW C31 0.1 mF 220 pF Laux R7 180 kW R11 22 W C29 D6 1N4148 C15 1 nF 4 2 C30 100 mF VCC R9 C16 100 pF 1 kW 5 3 D8 MR856 R4 27 kW 8 MC44603P C9 220 pF C1 100 mF C19 100 nF C22 0.1 mF R23 117.5 kW D14 1N4733 C20 33 nF TL431 R22 2.5 kW Figure 7. 110 W Output Off-Line Flyback Converter with Bipolar Switch. 180 V-280 V MAINS RANGE 14 Motorola Applications Data AN1669 Table 3. 110 W Fly-Back Converter, 180 V-280 V Mains Range, MC44603 and MJF18206 Test Conditions Line Regulation Results Vin = 180 Vac to 280 Vac Fmains = 50 HZ 120 V 28 V Iout = 1A 15 V Iout = 1A 8V Iout = 1A Load Regulation D=0V D=0V D=0V D=0V Iout = 0.5A Vin = 220 Vac 120 V Cross Regulation D = 0.05 V Iout = 0.2A to 0.5A Vin = 220 Vac Iout (120 V) = 0.5A Iout (28 V) = 0A to 1A 120 V D=0V Iout (15 V) = 1A Iout (8 V ) = 1A Efficiency Vin = 220 Vac, Po = 110 W 85% Vin = 220 Vac, Pout = 0 W 3W Standby Mode P input Switch. freq. 20 KHz fully stable Output short circuit Safe on all outputs Start-up Pin 110 W Vac = 160 V BIPOLAR application: information about the transformer Lp5 Lp5 110 W Lp (turns) 64 Laux (turns) 5 L1 (turns) 40 L2 (turns) 10 L3 (turns) 5 L4 (turns) 3 Al (nH/turns2) 250 Core E-4215A Material B2 Former specific Thomson design Wire size (mm2) 0.315 all windings Flyback transformer construction For cost reduction and simplicity, all windings have the same size. For optimal Lp/Laux coupling, Laux is wound on the second section of Lp. Former The normalized primary/secondary isolation is obtained using the multi-slotted former depicted on the figure. This former uses designs patented by LCC Thomson. L14 L14 Lp4 Lp4 L13 L12 L13 Lp3 Lp3 L42 L42 Lp2 L11 L12 Lp2 L41 L41 Lp1 Lp1 L11 SMT4 Lp Primary Winding (Lp1//Lp3//Lp5) + (Lp2//Lp4) Laux Auxilliary Winding L1 High Voltage Secondary Winding L11//L12//(L13 + L14) L2 Secondary Winding (28 V) (2 X 10 turns) L3 Secondary Winding (15 V) (2 X 5 turns) L4 Secondary Winding (8 V) L41//L42 Motorola Applications Data 15 AN1669 V -- CONCLUSION These applications show a significant advantage of the fixed frequency mode: it enables us to precisely limit the maximum power that may be drawn by the converter from the mains (135 W in our case). Note that the stand by losses are lower in the 110 V application because in this case, only a low loss, costly clamping network is used to protect the MOSFET (no snubber). Indeed, the snubber and clamping arrangements dissipate some energy (that is not insignificant) at each switching. That is why the reduction of the switching frequency is a very effective means to decrease the stand-by losses (the snubber and clamping arrangement cannot be removed in most cases). This application note does not pay much attention to the MC44603's protection features. Two features are especially noteworthy: -- the foldback that protects the converter when there is an overload -- the effective demagnetization section that ensures a discontinuous mode Notes: -- the MOSFET on time losses are high in the 110 V application. The use of a MOSFET having a lower Rdson (or two MOSFET in parallel) would improve the efficiency -- because of these losses, the input range of the proposed solution is actually: 90 V-140 V In order to minimize the length of this application note, it does not consider a universal mains range application. Such a SMPS could be designed using the methods described here. APPENDIX OREGA TRANSFORMERS Type SMT1 SMT3 SMT4 SMT47 AL (ni) @100C Ferrite Wire (mm) Nmax 448 40 5H20 0.25 56 260 80 5H20 0.25 56 240 85 5H20 0.224 68 220 90 5H20 0.224 68 180 125 5H20 0.224 68 350 80 B1 0.315 75 250 110 B1 0.315 75 250 130 B3 0.315 75 190 160 B1 0.315 75 178 180 B1 0.315 75 336 110 B1 0.28 68 320 135 B3 0.315 52 274 140 B1 0.40 36 250 180 B3 0.40 36 238 200 B3 0.40 36 215 190 B1 0.40 36 192 210 B1 0.40 36 192 245 B3 0.40 36 560 100 B3 0.315 76 428 140 B1 0.40 46 428 150 B3 0.50 26 372 190 B3 7 X 0.2 20 315 220 B3 7 X 0.2 20 262 270 B1 7 X 0.2 20 234 310 B3 7 X 0.2 20 The last column indicates the maximum number of turns per slot (refer to page 14) that will fall within the insulation norms, when using wires whose size is indicated in the "wire" column. 16 Motorola Applications Data AN1669 NOTES Motorola Applications Data 17 AN1669 NOTES 18 Motorola Applications Data AN1669 All products are sold on Motorola's Terms & Conditions of Supply. 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