1
Motorola Applications Data
AN1669
MC44603 in a 110 W Output SMPS Application
(80-140 Vrms and 180-280 Vrms Mains Voltages)
by Joël Turchi
Power Management Products Operation
Application Laboratory, Motorola, Toulouse, France
The purpose of this application note is to present a way of
designing the MC44603 in a fly–back dedicated to a 110 W
output power supply in two distinct cases: 110 Vrms mains
and 220 Vrms mains.
For this range of power, the discontinuous mode must be
chosen as it limits the stress on the power switch and the out-
put diodes. This kind of working can be guaranteed, thanks to
the demagnetization arrangement of the MC44603.
This application note considers both high and low mains
voltages.
— In the high voltage a.c. line case, it deals with both
MOSFET and BIPOLAR transistor use.
— In the low mains voltage case, only the MOSFET solu-
tion is considered as the inductor peak current is high.
The MC44603 is a high flexibility SMPS controller. It offers
a wide variety of protection (accurate maximum current limita-
tion, Vcc overvoltage detection, fold–back, demagnetization,
soft start).
In addition, distinct working modes can be used with this
circuit:
—a fixed frequency working mode
Here, the frequency is given by the circuit oscillator.
—a variable frequency working mode
This kind of working can be obtained when the chosen
oscillator period is shorter than the cycle time for trans-
former magnetization and complete demagnetization.
—a stand–by mode
This reduces the working frequency during this phase.
This method reduces the stand–by losses that are
mainly proportional to the switching frequency, and
avoids the unstable very high frequency working that
is generally associated with traditional free frequency
systems.
In the MC44603 design proposed here, the variable
frequency mode is not used. Indeed, as shown later , the fixed
frequency mode enables a more accurate control of the power
that is drawn from the mains.
I — THE APPLICATIONS
OUTPUT: 110 W
In the following calculations, the maximum input power will
be considered equal to 135 W (η
[
80%).
Application 1: 110 V mains 80 Vrms ↔140 Vrms
Application 2: European mains 180 Vrms ↔ 280 Vrms
The outputs: 120 V 0.5 A
28 V 1.0 A
15 V 1.0 A
8 V 1.0 A
Consequences for the Transformer
This output voltage values are obtained using four second-
ary windings of the transformer. Now, as a diode is located
between the output capacitor and the winding, this diode volt-
age must be taken into account in the choice of the number of
turns of each of the windings.
So, if the diode voltage is considered equal to 1 V, the
desired voltages on the windings are nearly:
121 V
29 V
16 V
9 V
To obtain the output voltage values of the specification with
a good accuracy , we need at least 3 turns for the 9 V and then:
9 V 3 turns
16 V 5 turns
29 V 10 turns
121 V 40 turns
This solution seems to be correct, since with three turns, the
9 V should be well coupled; also, 40 turns (for the 120 V) is not
too large a value (no ferrite saturation).
In our application, in order to obtain a very well regulated
high voltage output, an opto coupler is used. By this means,
the regulation uses feedback drawn from the 120 V output.
II — GENERAL DESIGN EQUATIONS
Calculation of the Main SMPS Parameters
II–1 — Fixed frequency working mode:
To obtain a fixed frequency discontinuous working mode
(the discontinuous mode is guaranteed by the demagnetiza-
tion section), the transformer primary inductor magnetization
and demagnetization cycle must be shorter than the oscillator
period (that is the chosen working period) for any working
point.
This condition can be expressed by the following inequality:
(ineq 1)
Ton
)
Toff
v
Tosc
Order this document
by AN1669/D
MOTOROLA
SEMICONDUCTOR APPLICATION NOTE
Motorola, Inc. 1998
AN1669
2 Motorola Applications Data
where: Ton is the on–time, Toff is the off–time
Tosc is the MC44603 oscillator period
The current increases linearly during the on–time and
decreases linearly in the inductor for the off–time.
(eqn 1)
So, Ipk
+
Vin
L
Ton
(eqn 2)
and, Ipk
+
NVo
L
Toff
where: Ipk is the primary inductor peak current
Vin is the rectified a.c. line voltage
(fly–back input voltage)
L is the primary inductor value
NVo is the output voltage considered in the
primary side
(N: turn ratio)
(eqn 3)
So, Ton
)
Toff
+
L
Ipk
Vin
)
NVo
Vin
NVo
In addition, the energy drawn through the transformer
during one cycle is:
(eqn 4)
E
+
1
2
L
Ipk2
(eqn 5)
So, Pin (input power) is:
Pin
+
1
2
L
Ipk2
fosc
where fosc is the MC44603 oscillator frequency
Using equations 3 and 5 and inequality 1,
condition required to work in a fixed frequency mode:
(ineq 2)
L
fosc
v
1
2
Pin
ǒ
Vin
NVo
Vin
)
NVo
Ǔ
2
II–2 — Peak inductor current expression
Pin is given by the following expression:
(eqn 6)
Pin
+
Vin
Iin
where Iin is the input current
(eqn 7)
and Iin
+
1
2
Ipk
d
where d is the duty cycle
(eqn 8)
Now, Ton
+
d
Tosc
Using equations 1, 7 and 8:
(eqn 9)
Iin
+
1
2
L
Ipk2
fosc
Vin
Using equations 6 and 9:
(eqn 10)
Ipk
+
2
Pin
L
fosc
Ǹ
Thus, Peak inductor current:
(eqn 11)
(Ipk)max
+
2
(Pin)max
L
fosc
Ǹ
II–3 — Power switch on–time losses
II–3–1 — MOSFET:
Rdson being the MOSFET on–time resistor, the on–time
losses can be calculated from:
(eqn 12)
Pon
+ǒ
1
Tosc
Ǔ ŕ
Ton
0
Rdson
IT2
dt
where IT is the MOSFET current.
(eqn 13)
So, as: IT
+
Vin
t
L
(eqn 14)
Pon
+
1
3
Rdson
Vin2
Ton3
(Tosc
L)
2
Consequently, using equations 1, 5 and 14, the following
expression can be written:
(eqn 15)
Pon
+
2
2
Ǹ
3
Rdson
Pin
Vin
Pin
L
fosc
Ǹ
Consequently,
maximum MOSFET on–time losses
(eqn 16)
(Pon)max
+
2
2
Ǹ
3
Rdson
(Pin)max
(Vin)min
(Pin)max
L
fosc
Ǹ
Using the (Ipk)max value, the following equation could also
be written:
(Pon)max
+
1
3
Rdson
(Ipk)max2
dmax
where dmax is the maximum duty cycle.
II–3–2 — BIPOLAR TRANSISTOR
(eqn 17)
Pon
+
1
Tosc
ŕ
Ton
0
VCE
IT
dt
(eqn 18)
So: Pon
+
V
C
E
ǀ
IT
ǁ
(eqn 19)
Thus: Pon
+
V
C
E
Iin
where Iin is the input current
(eqn 20)
So: Pon
+
VCE
Pin
Vin
Consequently,
maximum BIPOLAR transistor on–time losses
(eqn 21)
(Pon)max
+
VCE
(Pin)max
(Vin)min
II–4 — Maximum duty cycle
(eqn 22)
d
+
Ton
Tosc
The duty cycle, d, is equal to:
AN1669
3
Motorola Applications Data
(eqn 23)
Now, Ton
+
L
Ipk
Vin
(eqn 24)
so, d
+
L
fosc
Ipk
Vin
Thus, using equation 10:
(eqn 25)
d
+
2
Pin
L
fosc
Vin2
Ǹ
So, maximum duty cycle:
(eqn 26)
dmax
+
2
(Pin)max
L
fosc
(Vin)min2
Ǹ
II–5 — Other design parameters
The other parameters that must be taken into account are:
(VT)max: the maximum voltage the power switch must
face:
(eqn 27)
(VT)max
+
(2
Ǹ
(Vin)max)
)
(N
120) (V)
where: (Vin)max is the maximum rms a.c. line voltage
(140 V or 280 V according to the line)
120 is the maximum output voltage
N is the transformer turn ratio between the
120 V output winding and the primary inductor
(VD)max: the maximum voltage the 120 V output diode
must face:
(eqn 28)
(VD)max
+ǒ
2
Ǹ
(Vin)max
N
Ǔ)
120 (V)
(ni): the transformer ferrite saturation parameter
To avoid any risk of saturation in the transformer , the induc-
tor peak current must be lower than (ni/ np), where np is the
turns number of the primary inductor.
(eqn 29)
np
+
Nxn
120V
Now,
where n120V is the turns number of the 120 V winding
(eqn 30)
ni
+
Nxn
120V xIpk
So,
These parameters are the main elements that have to be
taken into account as they allow you to choose the power
switch, the diodes, the transformer and the working frequency.
As shown by the preceding calculations, the design
parameters depend on some elements like N or the value of
(fosc x L).
SUMMARY
Condition Required to Work in a Fixed Frequency Mode L
fosc
v
1
2
Pin
ǒ
Vin
NVo
Vin
)
NVo
Ǔ
2
Maximum Peak Inductor Current (Ipk)max
+
2
(Pin)max
L
fosc
Ǹ
Maximum Power Mosfet On–T ime Losses (Pon)max
+
1
3
Rdson
(Ipk)max2
dmax
Maximum Bipolar T ransistor On–Time Losses (Pon)max
+
VCE
(Pin)max
(Vin)min
Maximum Duty Cycle dmax
+
2
(Pin)max
L
fosc
(Vin)min2
Ǹ
AN1669
4 Motorola Applications Data
III — APPLICATION 1: 110 V INPUT
III–1 — Choice of the transformer
One way to use the above design equations, consists of
drawing up a table showing how the main SMPS parameters
vary with the value of the turn ratio.
To calculate these values, it is necessary to know the input
power level. This value is taken equal to (135W) in our
application (135W corresponds to an efficiency equal to about
80%. The application results will show that this assumption
ensures a desirable margin with the nominal input voltage).
On the other hand, the parameters calculation shows that
(L x fosc)max is the (L x fosc) value that results in the lowest
(Ipk)max and (Pon)max ones (refer to Ipk or Pon expressions).
This (L x fosc) value is the maximum one that guarantees a
fixed frequency working for any working point (refer to section
II–1). The SMPS parameters given in the following table are
calculated using this threshold value.
Choice criteria and definition of the transformer:
As shown by the following table, the higher the turn ratio (N)
is, the lower the peak current is. Now, the (ni)max is propor-
tional to N and the voltage the transistor must face, increases
when N rises. That is why an optimal N value must be chosen.
In fact, there are three main choice criteria:
— the peak current and the on–time losses.
N must be as large as possible in order to reduce the
peak current and the on–time losses
— the voltage the power MOSFET must face.
Indeed, this voltage must be as low as possible to
reduce its cost and in order to decrease the Rdson.
That is why a MOSFET 400 V should be used. It is nec-
essary to have a safety voltage margin, to avoid the
need to incorporate a lossy and costly clamping network
that would cut the voltage spikes due to the leakage
inductor (refer to Figure 1). Consequently, N must be
chosen lower than 1.25 (VTmax = 350 V).
— the transformer must be well coupled.
This is to obtain a consistently accurate regulation of
the output and to reduce the leakage inductor and
hence the turning off spikes (refer to Figure 1). That is
why a low air–gap ferrite must be used. Practically, a
ferrite whose (ni) is lower than 200 A.turns, seems to be
a good choice. So, N must be lower than 1. (The appen-
dix gives details of OREGA transformers; the SMT4
suits our application).
Finally, in order to use a ferrite (ni = 180, AL = 250 nH/
turns2), (N = 0.75) seems to be a suitable value.
Indeed, this value should result in a well coupled trans-
former with a low leakage inductor value. On the other hand,
if we do not take into account the turning off spikes, then the
theoretical highest value the power MOSFET must face, is
290 V. Consequently, with a 400 V power switch, only a low
loss clamping arrangement is required.
The chosen MOSFET, is the MTP10N40E (Rdson = 0.55
W
,
400 V).
Consequently,
Lp
+
ALx (N x 40)2Lp 225
≈µH
and the optimal working frequency is:
Lpx fosc 9.3 fosc 41.3 kHz
≤≤
So, the following values can be chosen:
Lp = 225 µH
fosc ≈ 40 kHz (Rref = 10 kΩ, CT = 1nF)
and then: Ipk ≈ 5.4A
N(L.fosc)max (Ipk)max
(A) (VT)max
(V) (VD)max
(V) MOSFET on losses/ Rdson
(W/
W
)(ni)max
0.50 5.6 6.9 260 520 5.7 139
0.75 9.3 5.4 290 390 4.3 162
0.90 11.0 5.0 300 340 4.1 180
1.00 12.5 4.6 320 320 3.7 184
1.25 14.9 4.3 350 280 3.5 215
1.50 17.3 4.0 380 250 3.2 240
2.00 21.9 3.5 440 220 2.8 281
NOTE: N: turn ratio (refer to II–5)
(VT)max: maximum voltage the power switch must face
(VD)max: maximum voltage the 120 V output diode must face
AN1669
5
Motorola Applications Data
Figure 1. Voltage Spikes Due to the Leakage Inductor
TURNING OFF
SPIKES
ON–TIME OFF–TIME DEAD–TIME
TIME
VT
VIN
VIN + NVO
III–2 — MC44603 pins use: (refer to the application
schematics)
1 — Vcc (PIN 1):
The pin Vcc must be connected to a transformer auxilliary
winding. This extra winding turns number can be taken equal
to 5 in order to obtain a Vcc nearly equal to 15 V.
2 — Vc and OUTPUT (pins 2 and 3):
Vc is the output high state of the circuit. This pin offers the
possibility of setting the output source current at a different
level than the sink current but it is no use in our case.
In fact, a resistor of 33.2
W
must be connected between the
output and the MOSFET gate to make the switchings
smoother . A resistor of about 1 k
W
can be connected between
the gate and the ground (or the current sense external resis-
tor) to avoid any inadvertent MOSFET switching on due to
noise (see Figure 2).
Figure 2.
VCC
VC
output
33.2
W
1 k
W
Rs
1
2
3
3 — Foldback (pin 5):
Part of Vcc must be applied to this pin thanks to a resistor
divider. This voltage value must be slightly higher than 1 V in
normal use, so that this value drops below this threshold value
as soon as an overload occurs.
4 — Overvoltage protection (pin 6):
This pin can remain free and then, the Vcc threshold level
is fixed equal to nearly 17 V.
On the other hand, to make detection quicker and more
accurate, an external resistor divider can be used with a diode
and an integration capacitor (refer to the proposed applica-
tion). The resistor divider is not directly connected to the Vcc
because Vcc has a high time constant (refer to the application
schematic — Figure 4).
5 — Current sense (pin 7):
The current sense resistor must be designed in order to limit
the current below the maximum peak calculated in section II
in order to limit the power that the converter is able to draw
from the mains; in a fixed frequency mode, Pin = 1/2 x L x Ipk2
x fosc.
Now, (Ipk)max = 5.4A
In addition, the (Vcs) clamp level is nearly 1 V (refer to the
data sheet).
So, (Rs) the current sense resistor, must be equal to
(1 V / 5.4A), that is nearly: 0.18 Ω.
This value can be obtained using a 1W, 0.2 Ω resistor and
a resistor divider (442 Ω, 3.16 kΩ) (refer to Figure 3).
AN1669
6 Motorola Applications Data
Figure 3.
442
W
0.2
W
Rs
7
3.16k
W
Finally, as the fixed frequency mode is obtained for any
working point, the peak current limitation results in an accurate
input power limitation (135W in this application note).
6 — Oscillator (pins 10 and 16):
The oscillator frequency is determined by the couple
(CT, Rref) (refer to the data sheet).
Even if capacitors have discrete values, the choice of Rref
allows you to fix precisely the oscillator frequency (however,
Rref also fixes the internal current source (Iref), which must be
lower than 500 µA and higher than 100 µA).
fosc
+
40 kHz Rref
+
10kΩ
CT
+
1nF
7 — Stand–by mode (pins 12 and 15):
In the MC44603, it is possible to reduce the working fre-
quency when little power is being drawn from the mains
(stand–by mode).
This stand–by frequency is fixed by connecting a resistor
RFstby to pin 15, while the power level at which the stand–by
mode must be applied is determined by connecting another
resistor RPstby to pin 12 (this power level is labelled PthL in the
data sheet).
In the data sheet, the equations needed to calculate RFstby
and RPstby are indicated.
Using them, to obtain a power level equal to 10W and a
stand–by frequency equal to 20 kHz, the calculated RFstby
and RPstby values are:
RPstby = 8.45 kΩ and RFstby = 22.1 kΩ
AN1669
7
Motorola Applications Data
Figure 4. 110W Output Off–Line Flyback Converter with MOSFET Switch, 80 V ↔ 140 V Mains Voltage
C30
100
m
FC31
0.1
m
F
D8
MR856
C32 220 pF
120 V / 0.5 A
C27
1000
m
FC28
0.1
m
F
D9
MR852
C29 220 pF
28 V / 1 A
C25
1000
m
FC24
0.1
m
F
D10
MR852
C26 220 pF
15 V / 1 A
C21
1000
m
FC22
0.1
m
F
D11
MR852
C23 220 pF
8 V / 1 A
R22
2.5 k
W
R21
10 k
W
C19
100 nF
R23
117.5 k
W
R24
270
W
TL431
C20
33 nF
D14
1N4733
MOC8101
RFI
FILTER R1
1
W
/ 5 W
R25
1 k
W
C12
10 nF
C1
220
m
F
250 V 47 k
W
R2
22 k
W
/ 2 W
C11
1 nF
R19
10 k
W
R18
22.1 k
W
C13
100 nF 3.16 k
W
C9 1 nF
C10 1
m
F
R15
8.45 k
W
R16
10 k
W
R17
10 k
W
16
15
14
13
12
11
10
9
1
2
3
4
5
6
7
8
C4–C7
1 nF / 500 V
47 nF
80
VAC
T
O
140
VAC
C3
1 nF / 1 kV
L1
1
m
H1N4937
D5
1N4934
R4
27 k
W
C2
220
m
F
25 V
D1–D4
MR504
R5
1.21 k
W
D6
1N4148
R6
200
W
LP
Laux
R3
4.7 k
W
MC44603P
R10
33.2
W
R7
15 k
W
C15
1 nF
C16
100 pF
R8
1 k
W
R14
0.2
W
R26
1 k
W
R9
442
W
C14
4.7 nF
MTP10N40E
AN1669
8 Motorola Applications Data
Table 1. 110W Fly–Back Converter , 80 Vrms–140 Vrms Mains Range, MC44603 and MTP10N40E
Test Conditions Results
Line Regulation Vin = 90 Vac to 140 Vac Fmains = 50 HZ
Fmains = 50 HZ
120 V Iout = 0.5A
D
= 0 V
28 V Iout = 1A
D
= 0 V
15 V Iout = 1A
D
= 0 V
8V Iout = 1A
D
= 0 V
Load Regulation Vin = 110 Vac
120 V Iout = 0.3A to 0.5A
D
= 0.05 V
Cross Regulation Vin = 110 Vac
Iout (120 V) = 0.5A
Iout (28 V) = 0A to 1A
120 V Iout (15 V) = 1A
D
= 0 V
Iout (8 V ) = 1A
Efficiency Vin = 110 Vac, Po = 110 W 84.5%
Standby Mode
P input Vin = 110 Vac, Pout = 0 W 1.2 W
Switch. freq. 20 KHz fully stable
Output short circuit Safe on all outputs
Start–up Pin 110 W Vac = 80 V
MOSFET application: information about the transformer
110W
Lp (turns) 30
Laux (turns) 5
L1 (turns) 40
L2 (turns) 10
L3 (turns) 5
L4 (turns) 3
Al (nH/turns2) 274
Core E–4215A
Material B2
Former specific Thomson design
Wire size (mm2)0.315 all windings
Flyback transformer construction
For cost reduction and simplicity, all windings have the
same size. For optimal Lp/Laux coupling, Laux is wound on
the second section of Lp.
Former
The normalized primary/secondary isolation is obtained
using the multi–slotted former depicted on the figure. This for-
mer uses designs patented by LCC Thomson.
SMT4
Lp Primary Winding
(Lp1//Lp3//Lp5) + (Lp2//Lp4)
Laux Auxilliary Winding
L1 High Voltage Secondary Winding
L11//L12//(L13 + L14)
L2 Secondary Winding (28 V)
(2 X 10 turns)
L3 Secondary Winding (15 V)
(2 X 5 turns)
L4 Secondary Winding (8 V)
L41//L42
Lp1
L11 L41
L12 L42
Lp2
Lp3
L13
Lp4
L14
Lp5
Lp1
L41
L42
Lp3
Lp4
Lp5
L11
L12
Lp2
L13
L14
AN1669
9
Motorola Applications Data
IV — APPLICATION 2: 220 V INPUT VOLTAGE
IV–1 — Choice of the transformer:
One way to use the above design equations, consists of
drawing up a table showing how the main SMPS parameters
vary with the value of the turn ratio.
To calculate these values, it is necessary to know the input
power level. This value is taken equal to (135W) in our
application (135W corresponds to an efficiency equal to about
80%. The application results will show that this assumption
ensures a desirable margin with the nominal input voltage).
On the other hand, the parameters calculation shows that
(L x fosc)max is the (L x fosc) value that results in the lowest
(Ipk)max and (pon)max ones (refer to Ipk or pon expressions).
This (L x fosc) value is the maximum one that guarantees a
fixed frequency working for any working point (refer to section
II–1). The SMPS parameters given in the following table are
calculated using this threshold value.
Choice criteria and definition of the transformer:
As shown by the following table, the higher the turn ratio (N)
is, the lower the peak current is. Now, the (ni)max is propor-
tional to N and the voltage the transistor must face, increases
when N rises. That is why an optimal N value must be chosen.
MOSFET case:
T o perform a low cost SMPS, it is required to use a MOSFET
600 V . It is necessary to have a safety voltage margin, to avoid
the need to incorporate a lossy and costly clamping network
that would cut the voltage spikes due to the leakage inductor
at the power switch turning off (refer to Figure 1 in section
III–1).
Practically, about 550 V is acceptable. Consequently,
(N = 1.2) seems to be a maximum value.
Now, in order to obtain a well coupled transformer with a low
leakage inductor value, it is desirable to use a ferrite with a low
air–gap.
So, in order to be able to use a ferrite (ni = 140, AL = 274 nH/
turns2), (N = 1) seems to be a preferable value.
Consequently,
Lp
+
ALx (N x 40)2Lp 438
≈µH
and the optimal working frequency is:
L x fosc 24.3 fosc 55 kHz
≤≤
So, the following values can be chosen:
L = 438 µH
fosc = 50 kHz
and then Ipk = 3.5A
BIPOLAR transistor case:
As the gain of a Bipolar transistor decreases when the col-
lector current level rises, the SMPS peak current must be as
low as possible. That is why N must be chosen as high as pos-
sible. Now, if classical BIPOLAR transistors are able to face
1000 V or 1200 V, their VCEO is generally low . The transistor
used in the application, the MJE18206, has a VCES equal to
1200 V and a VCEO equal to 600 V. Since there are damped
oscillations (converging to Vin) during the dead– time (refer to
Figure 1), the transistor may be turned on while its VCE voltage
is higher than Vin (the maximum V in value being nearly equal
to 400 V). That is why , even if a resistor is connected between
the base and the emitter of the transistor (refer to section
IV–2), the (VT)max (that is, (V in+NV o)max) must be chosen
lower than 600 V, to ensure system reliability.
In addition to this, a second choice criterion is (ni)max, since
transformer saturation must be avoided.
(N = 1.6) seems to be a good choice that enables the use
of a ferrite (AL = 250nH/turns2; ni = 180))
Consequently,
L
+
ALx (N x 40)2L 1mH
≈
So, the optimal working frequency is:
L x fosc 43.7 fosc 43 kHz
≤≤
Finally, the following value can be taken:
L = 1mH
fosc = 43 kHz
(Ipk)max = 2.5A
N (L.fosc)max (Ipk)max
(A) (VT)max
(V) (VD)max
(V)
MOSFET on
losses/Rdson
(W/
W
)
BIPOLAR on
losses/VCE
(W/A) (ni)max
0.75 16.2 4.1 490 650 1.5 0.54 122
1.00 24.3 3.3 520 520 1.2 0.54 133
1.20 30.9 3.0 540 450 1.1 0.54 144
1.40 37.4 2.7 570 400 1.0 0.54 150
1.60 43.7 2.5 590 370 0.9 0.54 159
1.80 49.7 2.3 620 340 0.8 0.54 168
2.00 55.5 2.2 640 320 0.8 0.54 176
NOTE: N: turn ratio (refer to II–5)
(VT)max: maximum voltage the power switch must face
(VD)max: maximum voltage the 120 V output diode must face
AN1669
10 Motorola Applications Data
IV–2 — MC44603 pins use: (refer to the application
schematics)
1 — Vcc (PIN 1):
The pin Vcc must be connected to a transformer auxilliary
winding. This extra winding turns number can be chosen equal
to 5, in order to obtain a Vcc nearly equal to 15 V.
2 — Vc and OUTPUT (pins 2 and 3):
Vc is the output high state of the circuit. This pin offers the
possibility of setting the output source current at a different
level than the sink current.
— MOSFET case:
A resistor of 10 Ω must be connected between the output and
the MOSFET gate to make the switchings smoother. A resistor
of about 1 kΩ can be connected between the gate and the
ground (or the current sense external resistor) to avoid any
inadvertent MOSFET switching on due to noise.
— BIPOLAR transistor case:
For the on–time, a bipolar transistor requires a base current
labelled IB1, that must be higher than:
(Ic)max / ßmin
where (Ic)max is the maximum collector current
(that is Ipkmax if the current sense resistor is well
designed — refer to section IV–5), and
ßmin is the minimum guaranteed transistor
gain for (Ic = (Ic)max)
Now, with the MJE18206: (ßmin ≈7) for (Ic)max = 2.5A
So, (IB1 = 400mA ) is a good value that ensures a safety
margin.
On the other hand, the turn off base current peak must be
nearly equal to (2 x IB1). The couple (Dz,Cz) is used to build
a voltage source Vz (during the on–time), able to produce IB2.
So, IB1 = (Vcc – Vz – Vbe) / (r1 + r2)
IB2 = (Vz + Vbe) / r2
Consequently, using Vz = 3.3 V
Cz = 1 µF
r1 = 22
W
r2 = 4.7
W
As (Vcc ≈ 15 V), the obtained base currents are:
IB1 ≈ 410 mA
IB2 ≈ 850 mA
These base currents enable a correct transistor drive.
3 — Foldback (pin 5):
A portion of Vcc must be applied to this pin thanks to a resis-
tor divider . This voltage value must be slightly higher than 1 V
in normal working so that this value drops below this threshold
value as soon as an overload occurs.
4 — Overvoltage protection (pin 6):
This pin can remain free and then, the Vcc threshold level
is fixed equal to nearly 17 V.
On the other hand, to make detection quicker and more
accurate, an external resistor divider can be used with a diode
and an integration capacitor . In the proposed application, this
resistor divider is not directly connected to the Vcc because
Vcc has a high time constant (refer to the application
schematics).
5 — Current sense (pin 7):
The current sense resistor must be designed in order to limit
the current down to the maximum peak calculated in section
II in order to limit the power the converter is able to draw from
the mains (in a fixed frequency mode, Pin = 1/ 2 x L x Ipk2 x
fosc).
— MOSFET case: (Ipk)max = 3.5A
Now, the (Vcs) clamp level is nearly 1 V (refer to the data
sheet).
So, (Rs) the current sense resistor, must be equal to
(1 V / 3.5A), that is nearly: 0.28
W
Ω (2 x 0.56 Ω in parallel).
— BIPOLAR case: (Ipk)max = 2.5A
So, (Rs) must be equal to (1 V / 2.5A), that is: 0.4
W
(3 x 1.2
W
in parallel).
Finally, as the fixed frequency mode is ensured for any
working point, the peak current limitation results in an accurate
input power limitation (135W in this application).
Figure 5.
VCC
VC
output
10
W
1 k
W
Rs
1
2
3
MOSFET drive BIPOLAR transistor drive
VCC
VC
output
r2
4.7
W
47
W
Rs
1
2
3
r1
22
W
Cz
1 µF
Dz (3.3V)
AN1669
11
Motorola Applications Data
6 — Oscillator (pin 10 & 16):
The oscillator frequency is determined by the couple
(CT, Rref) (refer to the data sheet).
As capacitors have discrete values, the choice of Rref
allows you to fix precisely the oscillator frequency (however,
Rref also fixes the internal current source (Iref), which must be
lower than 500 µA and higher than 100 µA).
MOSFET case:
fosc = 50 kHz Rref = 10 kΩCT= 820pF
BIPOLAR case:
fosc = 43 kHz Rref = 10 kΩCT= 1nF
7 — Stand–by mode (pins 12 and 15):
In the MC44603, it is possible to reduce the working fre-
quency when little power is being drawn from the mains
(stand–by mode).
This stand–by frequency is fixed by connecting a resistor
RFstby to pin 15, while the power level at which the stand–by
mode must be applied is determined by connecting another
resistor RPstby to pin 12 (this power level is labelled PthL in the
data sheet).
In the data sheet, the equations needed to calculate RFstby
and RPstby are indicated.
Using them, to obtain a power level equal to 15W and a
stand–by frequency equal to 20 kHz, the calculated RFstby and
RPstby values are:
MOSFET case: RPstby = 10 kΩRFstby = 27 kΩ
BIPOLAR case: RPstby = 10 kΩRFstby = 22 kΩ
AN1669
12 Motorola Applications Data
Figure 6. 110 W Output Off–Line Flyback Converter with MOSFET Switch. 180 V–280 V MAINS RANGE
C30
100
m
FC31
0.1
m
F
D8
MR856
C32 220 pF
120 V / 0.5 A
C27
1000
m
FC28
0.1
m
F
D9
MR852
C29 220 pF
28 V / 1 A
C25
1000
m
FC24
0.1
m
F
D10
MR852
C26 220 pF
15 V / 1 A
C21
1000
m
FC22
0.1
m
F
D11
MR852
C23 220 pF
8 V / 1 A
R22
2.5 k
W
R21
10 k
W
C19
100 nF
R23
117.5 k
W
R24
270
W
TL431
C20
33 nF
D14
1N4733
MOC8101
RFI
FILTER R1
1
W
/ 5 W
R25
1 k
W
C12
6.8 nF
C1
100
m
FR20
22 k
W
5 W
R2
68 k
W
/ 2 W
C11
1 nF
R19
10 k
W
R18
27 k
W
C13
100 nF
C9 820 pF
C10 1
m
F
R15
10 k
W
R16
10 k
W
R17
10 k
W
16
15
14
13
12
11
10
9
1
2
3
4
5
6
7
8
C4–C7
1 nF / 1000 V
C17
47 nF
180
VAC
T
O
280
VAC
C3
1 nF / 1 KV
L1
1
m
HD7
MR856
D5
1N4934
R4
27 k
W
C2
220
m
F
D1–D4
1N4007
R5
1.2 k
W
D6
1N4148
R6
180
W
LP
Laux
R3
4.7 k
W
MC44603P
R10
10
W
R7
180 k
W
C15
1 nF
C16
100 pF
R8
15 k
W
R14
2 X 0.56
W
//
R26
1 k
W
C14
4.7 nF
MTP6N60E
R9
1 k
W
AN1669
13
Motorola Applications Data
Table 2. 110W Fly–Back Converter , 180 V–280 V Mains Range, MC44603 and MTP6N60E
Test Conditions Results
Line Regulation Vin = 180 Vac to 280 Vac
Fmains = 50 HZ
120 V Iout = 0.5A
D
= 0 V
28 V Iout = 1A
D
= 0 V
15 V Iout = 1A
D
= 0 V
8V Iout = 1A
D
= 0 V
Load Regulation Vin = 220 Vac
155 V Iout = 0.3A to 0.5A
D
= 0.05 V
Cross Regulation Vin = 220 Vac
Iout (120 V) = 0.5A
Iout (28 V) = 0A to 1A
120 V Iout (15 V) = 1A
D
= 0 V
Iout (8 V ) = 1A
Efficiency Vin = 220 Vac, Po = 110 W 84%
Standby Mode
P input Vin = 220 Vac, Pout = 0 W 3 W
Switch. freq. 20 KHz fully stable
Output short circuit Safe on all outputs
Start–up Pin 110 W Vac = 160 V
MOSFET application: information about the transformer
110W
Lp (turns) 40
Laux (turns) 5
L1 (turns) 40
L2 (turns) 10
L3 (turns) 5
L4 (turns) 3
Al (nH/turns2) 274
Core E–4215A
Material B2
Former specific Thomson design
Wire size (mm2)0.315 all windings
Flyback transformer construction
For cost reduction and simplicity, all windings have the
same size. For optimal Lp/Laux coupling, Laux is wound on
the second section of Lp.
Former
The normalized primary/secondary isolation is obtained
using the multi–slotted former depicted on the figure. This for-
mer uses designs patented by LCC Thomson.
SMT4
Lp Primary Winding
(Lp1//Lp3//Lp5) + (Lp2//Lp4)
Laux Auxilliary Winding
L1 High Voltage Secondary Winding
L11//L12//(L13 + L14)
L2 Secondary Winding (28 V)
(2 X 10 turns)
L3 Secondary Winding (15 V)
(2 X 5 turns)
L4 Secondary Winding (8 V)
L41//L42
Lp1
L11 L41
L12 L42
Lp2
Lp3
L13
Lp4
L14
Lp5
Lp1
L41
L42
Lp3
Lp4
Lp5
L11
L12
Lp2
L13
L14
AN1669
14 Motorola Applications Data
Figure 7. 110 W Output Off–Line Flyback Converter with Bipolar Switch. 180 V–280 V MAINS RANGE
C30
100
m
FC31
0.1
m
F
D8
MR856
C32 220 pF
120 V / 0.5 A
C27
1000
m
FC28
0.1
m
F
D9
MR852
C29 220 pF
28 V / 1 A
C25
1000
m
FC24
0.1
m
F
D10
MR852
C26 220 pF
15 V / 1 A
C21
1000
m
FC22
0.1
m
F
D11
MR852
C23 220 pF
8 V / 1 A
R22
2.5 k
W
R21
10 k
W
C19
100 nF
R23
117.5 k
W
R24
270
W
TL431
C20
33 nF
D14
1N4733
MOC8101
RFI
FILTER R1
1
W
/ 5 W
R25
1 k
W
C12
6.8 nF
C1
100
m
F
R2
68 k
W
/ 2 W
C11
1 nF
R19
10 k
W
R18
22 k
W
C13
100 nF
C9 1 nF
C10 1
m
F
R15
10 k
W
R16
10 k
W
R17
10 k
W
16
15
14
13
12
11
10
9
1
2
3
4
5
6
7
8
C4–C7
1 nF / 1000 V
180
VAC
T
O
280
VAC
C3
1 nF / 1 KV
L1
1
m
H
D5
1N493
4
R4
27 k
W
C2
220
m
F
D1–D4
1N4007
R5
1.2 k
W
D6
1N4148
R6
180
W
LP
Laux
R3
4.7 k
W
MC44603P
R26
R7
180 k
W
C15
1 nF
C16
100 pF
R8
15 k
W
R14
3 X 1.2
W
//
47
W
C14
4.7 nF
MJF18206
R9
1 k
W
R13
1 k
W
4.7
W
C34 1
m
FD12
MR856
C18
2.2 nF
R11
22
W
VCC
D13
1N4728
AN1669
15
Motorola Applications Data
Table 3. 110W Fly–Back Converter , 180 V–280 V Mains Range, MC44603 and MJF18206
Test Conditions Results
Line Regulation Vin = 180 Vac to 280 Vac
Fmains = 50 HZ
120 V Iout = 0.5A
D
= 0 V
28 V Iout = 1A
D
= 0 V
15 V Iout = 1A
D
= 0 V
8V Iout = 1A
D
= 0 V
Load Regulation Vin = 220 Vac
120 V Iout = 0.2A to 0.5A
D
= 0.05 V
Cross Regulation Vin = 220 Vac
Iout (120 V) = 0.5A
Iout (28 V) = 0A to 1A
120 V Iout (15 V) = 1A
D
= 0 V
Iout (8 V ) = 1A
Efficiency Vin = 220 Vac, Po = 110 W 85%
Standby Mode
P input Vin = 220 Vac, Pout = 0 W 3W
Switch. freq. 20 KHz fully stable
Output short circuit Safe on all outputs
Start–up Pin 110 W Vac = 160 V
BIPOLAR application: information about the transformer
110W
Lp (turns) 64
Laux (turns) 5
L1 (turns) 40
L2 (turns) 10
L3 (turns) 5
L4 (turns) 3
Al (nH/turns2) 250
Core E–4215A
Material B2
Former specific Thomson design
Wire size (mm2)0.315 all windings
Flyback transformer construction
For cost reduction and simplicity, all windings have the
same size. For optimal Lp/Laux coupling, Laux is wound on
the second section of Lp.
Former
The normalized primary/secondary isolation is obtained
using the multi–slotted former depicted on the figure. This for-
mer uses designs patented by LCC Thomson.
SMT4
Lp Primary Winding
(Lp1//Lp3//Lp5) + (Lp2//Lp4)
Laux Auxilliary Winding
L1 High Voltage Secondary Winding
L11//L12//(L13 + L14)
L2 Secondary Winding (28 V)
(2 X 10 turns)
L3 Secondary Winding (15 V)
(2 X 5 turns)
L4 Secondary Winding (8 V)
L41//L42
Lp1
L11 L41
L12 L42
Lp2
Lp3
L13
Lp4
L14
Lp5
Lp1
L41
L42
Lp3
Lp4
Lp5
L11
L12
Lp2
L13
L14
AN1669
16 Motorola Applications Data
V — CONCLUSION
These applications show a significant advantage of the
fixed frequency mode: it enables us to precisely limit the maxi-
mum power that may be drawn by the converter from the
mains (135 W in our case).
Note that the stand by losses are lower in the 1 10 V applica-
tion because in this case, only a low loss, costly clamping net-
work is used to protect the MOSFET (no snubber). Indeed, the
snubber and clamping arrangements dissipate some energy
(that is not insignificant) at each switching. That is why the
reduction of the switching frequency is a very effective means
to decrease the stand–by losses (the snubber and clamping
arrangement cannot be removed in most cases).
This application note does not pay much attention to the
MC44603’s protection features. Two features are especially
noteworthy:
— the foldback that protects the converter when there is
an overload
— the effective demagnetization section that ensures a
discontinuous mode
Notes:
— the MOSFET on time losses are high in the 110 V
application. The use of a MOSFET having a lower
Rdson (or two MOSFET in parallel) would improve the
efficiency
— because of these losses, the input range of the pro-
posed solution is actually: 90 V–140 V
In order to minimize the length of this application note, it
does not consider a universal mains range application. Such
a SMPS could be designed using the methods described here.
APPENDIX
OREGA TRANSFORMERS
Type AL(ni) @100°C Ferrite Wire ∅ (mm) Nmax
SMT1 448 40 5H20 0.25 56
260 80 5H20 0.25 56
240 85 5H20 0.224 68
220 90 5H20 0.224 68
180 125 5H20 0.224 68
SMT3 350 80 B1 0.315 75
250 110 B1 0.315 75
250 130 B3 0.315 75
190 160 B1 0.315 75
178 180 B1 0.315 75
SMT4 336 110 B1 0.28 68
320 135 B3 0.315 52
274 140 B1 0.40 36
250 180 B3 0.40 36
238 200 B3 0.40 36
215 190 B1 0.40 36
192 210 B1 0.40 36
192 245 B3 0.40 36
SMT47 560 100 B3 0.315 76
428 140 B1 0.40 46
428 150 B3 0.50 26
372 190 B3 7 X 0.2 20
315 220 B3 7 X 0.2 20
262 270 B1 7 X 0.2 20
234 310 B3 7 X 0.2 20
The last column indicates the maximum number of turns per slot (refer to page 14) that will fall within the insulation norms, when using wires
whose size is indicated in the ”wire” column.
AN1669
17
Motorola Applications Data
NOTES
AN1669
18 Motorola Applications Data
NOTES
AN1669
19
Motorola Applications Data
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updated, revised or withdrawn.
AN1669
20 Motorola Applications Data
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AN1669/D
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