REV. C
a
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700www.analog.com
Fax: 781/326-8703 © Analog Devices, Inc., 2002
AD828
Dual, Low Power
Video Op Amp
FEATURES
Excellent Video Performance
Differential Gain and Phase Error of 0.01% and 0.05
High Speed
130 MHz 3 dB Bandwidth (G = +2)
450 V/s Slew Rate
80 ns Settling Time to 0.01%
Low Power
15 mA Max Power Supply Current
High Output Drive Capability
50 mA Minimum Output Current per Amplifier
Ideal for Driving Back Terminated Cables
Flexible Power Supply
Specified for +5 V, 5 V, and 15 V Operation
3.2 V Min Output Swing into a 150 Load
(VS = 5 V)
Excellent DC Performance
2.0 mV Input Offset Voltage
Available in 8-Lead SOIC and 8-Lead Plastic Mini-DIP
FUNCTIONAL BLOCK DIAGRAM
1
2
3
4
8
7
6
5
AD828
V+
OUT2
–IN2
+IN2
OUT1
–IN1
+IN1
V–
GENERAL DESCRIPTION
The AD828 is a low cost, dual video op amp optimized for use
in video applications that require gains of +2 or greater and
high output drive capability, such as cable driving. Due to its
low power and single-supply functionality, along with excellent
differential gain and phase errors, the AD828 is ideal for power-
sensitive applications such as video cameras and professional
video equipment.
With video specs like 0.1 dB flatness to 40 MHz and low
differential gain and phase errors of 0.01% and 0.05°, along
with 50 mA of output current per amplifier, the AD828 is an
excellent choice for any video application. The 130 MHz gain
bandwidth and 450 V/µs slew rate make the AD828 useful in
many high speed applications, including video monitors, CATV,
color copiers, image scanners, and fax machines.
1/2
AD828
0.1F
0.1F
+V
–V
R
BT
7575
R
T
75
1k
R
T
75
1k
V
IN
Figure 1. Video Line Driver
The AD828 is fully specified for operation with a single 5 V
power supply and with dual supplies from ±5 V to ±15 V. This
power supply flexibility, coupled with a very low supply current
of 15 mA and excellent ac characteristics under all power supply
conditions, make the AD828 the ideal choice for many demand-
ing yet power-sensitive applications.
The AD828 is a voltage feedback op amp that excels as a gain
stage (gains > +2) or active filter in high speed and video systems
and achieves a settling time of 45 ns to 0.1%, with a low input
offset voltage of 2 mV max.
The AD828 is available in low cost, small 8-lead plastic mini-DIP
and SOIC packages.
0.04 15
0.07
0.05
0.06
510
0.03
0.01
0.02
SUPPLY VOLTAGE – V
DIFFERENTIAL PHASE – Degrees
DIFFERENTIAL GAIN – Percent
DIFF GAIN
DIFF PHASE
Figure 2. Differential Phase vs. Supply Voltage
REV. C–2–
AD828–SPECIFICATIONS
(@ TA = 25C, unless otherwise noted.)
Parameter Conditions V
S
Min Typ Max Unit
DYNAMIC PERFORMANCE
–3 dB Bandwidth Gain = +2 ±5 V 60 85 MHz
±15 V 100 130 MHz
0, +5 V 30 45 MHz
Gain = –1 ±5 V 35 55 MHz
±15 V 60 90 MHz
0, +5 V 20 35 MHz
Bandwidth for 0.1 dB Flatness Gain = +2 ±5 V 30 43 MHz
C
C
= 1 pF ±15 V 30 40 MHz
0, +5 V 10 18 MHz
Gain = –1 ±5 V 15 25 MHz
C
C
= 1 pF ±15 V 30 50 MHz
0, +5 V 10 19 MHz
Full Power Bandwidth
*
V
OUT
= 5 V p-p
R
LOAD
= 500 Ω±5 V 22.3 MHz
V
OUT
= 20 V p-p
R
LOAD
= 1 kΩ±15 V 7.2 MHz
Slew Rate R
LOAD
= 1 kΩ±5 V 300 350 V/µs
Gain = –1 ±15 V 400 450 V/µs
0, +5 V 200 250 V/µs
Settling Time to 0.1% –2.5 V to +2.5 V ±5 V 45 ns
0 V–10 V Step, A
V
= –1 ±15 V 45 ns
Settling Time to 0.01% –2.5 V to +2.5 V ±5 V 80 ns
0 V–10 V Step, A
V
= –1 ±15 V 80 ns
NOISE/HARMONIC PERFORMANCE
Total Harmonic Distortion F
C
= 1 MHz ±15 V –78 dB
Input Voltage Noise f = 10 kHz ±5 V, ±15 V 10 nV/√Hz
Input Current Noise f = 10 kHz ±5 V, ±15 V 1.5 pA/√Hz
Differential Gain Error NTSC ±15 V 0.01 0.02 %
(R
L
= 150 Ω)Gain = +2 ±5 V 0.02 0.03 %
0, +5 V 0.08 %
Differential Phase Error NTSC ±15 V 0.05 0.09 Degrees
(R
L
= 150 Ω)Gain = +2 ±5 V 0.07 0.1 Degrees
0, +5 V 0.1 Degrees
DC PERFORMANCE
Input Offset Voltage ±5 V, ±15 V 0.5 2 mV
T
MIN
to T
MAX
3mV
Offset Drift 10 µV/°C
Input Bias Current ±5 V, ±15 V 3.3 6.6 µA
T
MIN
10 µA
T
MAX
4.4 µA
Input Offset Current ±5 V, ±15 V 25 300 nA
T
MIN
to T
MAX
500 nA
Offset Current Drift 0.3 nA/°C
Open-Loop Gain V
OUT
= ±2.5 V ±5 V
R
LOAD
= 500 Ω35 V/mV
T
MIN
to T
MAX
2V/mV
R
LOAD
= 150 Ω24 V/mV
V
OUT
= ±10 V ±15 V
R
LOAD
= 1 kΩ5.5 9 V/mV
T
MIN
to T
MAX
2.5 V/mV
V
OUT
= ±7.5 V ±15 V
R
LOAD
= 150 Ω (50 mA Output) 3 5 V/mV
INPUT CHARACTERISTICS
Input Resistance 300 kΩ
Input Capacitance 1.5 pF
Input Common-Mode Voltage Range ±5 V +3.8 +4.3 V
–2.7 –3.4 V
±15 V +13 +14.3 V
–12 –13.4 V
0, +5 V +3.8 +4.3 V
+1.2 +0.9 V
Common-Mode Rejection Ratio V
CM
= +2.5 V, T
MIN
to T
MAX
±5 V 82 100 dB
V
CM
= ±12 V ±15 V 86 120 dB
T
MIN
to T
MAX
±15 V 84 100 dB
Parameter Conditions V
S
Min Typ Max Unit
OUTPUT CHARACTERISTICS
Output Voltage Swing R
LOAD
= 500 Ω±5 V 3.3 3.8 ±V
R
LOAD
= 150 Ω±5 V 3.2 3.6 ±V
R
LOAD
= 1 kΩ±15 V 13.3 13.7 ±V
R
LOAD
= 500 Ω±15 V 12.8 13.4 ±V
1.5
R
LOAD
= 500 Ω0, +5 V 3.5 ±V
Output Current ±15 V 50 mA
±5 V 40 mA
0, +5 V 30 mA
Short Circuit Current ±15 V 90 mA
Output Resistance Open-Loop 8 Ω
MATCHING CHARACTERISTICS
Dynamic
Crosstalk f = 5 MHz ±15 V –80 dB
Gain Flatness Match G = +1, f = 40 MHz ±15 V 0.2 dB
Skew Rate Match G = –1 ±15 V 10 V/µs
DC
Input Offset Voltage Match T
MIN
to T
MAX
±5 V, ±15 V 0.5 2 mV
Input Bias Current Match T
MIN
to T
MAX
±5 V, ±15 V 0.06 0.8 µA
Open-Loop Gain Match V
O
= ±10 V, R
L
= 1 kΩ, T
MIN
to T
MAX
±15 V 0.01 0.15 mV/V
Common-Mode Rejection Ratio Match V
CM
= ±12 V, T
MIN
to T
MAX
±15 V 80 100 dB
Power Supply Rejection Ratio Match ±5 V to ±15 V, T
MIN
to T
MAX
80 100 dB
POWER SUPPLY
Operating Range Dual Supply ±2.5 ±18 V
Single Supply +5 +36 V
Quiescent Current ±5 V 14.0 15 mA
T
MIN
to T
MAX
±5 V 14.0 15 mA
T
MIN
to T
MAX
±5 V 15 mA
Power Supply Rejection Ratio V
S
= ±5 V to ±15 V, T
MIN
to T
MAX
80 90 dB
*Full power bandwidth = slew rate/2 π V
PEAK
.
Specifications subject to change without notice.
ABSOLUTE MAXIMUM RATINGS
1
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±18 V
Internal Power Dissipation
2
Plastic DIP (N) . . . . . . . . . . . . . . . . . . See Derating Curves
Small Outline (R) . . . . . . . . . . . . . . . . . See Derating Curves
Input Voltage (Common Mode) . . . . . . . . . . . . . . . . . . . . ±V
S
Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . ±6 V
Output Short Circuit Duration . . . . . . . . See Derating Curves
Storage Temperature Range (N, R) . . . . . . . . –65°C to +125°C
Operating Temperature Range . . . . . . . . . . . . –40°C to +85°C
Lead Temperature Range (Soldering 10 sec) . . . . . . . . +300°C
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
Specification is for device in free air:
8-Lead Plastic DIP Package: θ
JA
= 100°C/W
8-Lead SOIC Package: θ
JA
= 155°C/W
2.0
0
–50 90
1.5
0.5
–30
1.0
50 703010–10 80–40 40 60200–20
AMBIENT TEMPERATURE – C
MAXIMUM POWER DISSIPATION – Watts
8-LEAD MINI-DIP PACKAGE
8-LEAD SOIC PACKAGE
TJ = 150C
Figure 3. Maximum Power Dissipation vs.
Temperature for Different Package Types
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the AD828 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
REV. C
AD828
–3–
ORDERING GUIDE
Temperature Package Package
Model Range Description Option
AD828AN –40°C to +85°C8-Lead Plastic DIP N-8
AD828AR –40°C to +85°C8-Lead Plastic SOIC SO-8
AD828AR-REEL7 –40°C to +85°C7" Tape and Reel SO-8
AD828AR-REEL –40°C to +85°C13" Tape and Reel SO-8
REV. C–4–
AD828
20
0
020
15
5
5
10
10 15
INPUT COMMON-MODE RANGE – V
SUPPLY VOLTAGE – V
–VCM
+VCM
TPC 1. Common-Mode Voltage Range vs. Supply
Voltage
20
0020
15
5
5
10
10 15
SUPPLY VOLTAGE – V
OUTPUT VOLTAGE SWING – V
RL = 150
RL = 500
TPC 2. Output Voltage Swing vs. Supply Voltage
30
0
10k
15
5
100
10
10
20
1k
25
OUTPUT VOLTAGE SWING – V p-p
LOAD RESISTANCE –
Vs = 15V
Vs = 5V
TPC 3. Output Voltage Swing vs. Load Resistance
–40C
7.7
5.7
020
7.2
6.2
5
6.7
10 15
SUPPLY VOLTAGE – V
QUIESCENT SUPPLY CURRENT PER AMP – mA
+25C
+85C
TPC 4. Quiescent Supply Current per Amp vs. Supply
Voltage for Various Temperatures
SLEW RATE – V/s
20501510
SUPPLY VOLTAGE – V
300
400
450
500
350
TPC 5. Slew Rate vs. Supply Voltage
FREQUENCY – Hz
100
1
0.01
1k 100M10k
CLOSED-LOOP OUTPUT IMPEDANCE –
100k 1M 10M
10
0.1
TPC 6. Closed-Loop Output Impedance vs. Frequency
—Typical Performance Characteristics
REV. C
AD828
–5–
7
1
140
4
2
–40
3
–60
6
5
120806040 100200–20
TEMPERATURE – C
INPUT BIAS CURRENT – A
TPC 7. Input Bias Current vs. Temperature
130
30
140
90
50
–40
70
–60
110
120100806040200–20
TEMPERATURE – C
SHORT CIRCUIT CURRENT – mA
SOURCE CURRENT
SINK CURRENT
TPC 8. Short Circuit Current vs. Temperature
80
40
–60 140
70
50
–40
60
100 120806040200–20
TEMPERATURE – C
PHASE MARGIN – Degrees
PHASE MARGIN
40
70
50
60
–3dB BANDWIDTH – MHz
80
GAIN BANDWIDTH
TPC 9. –3 dB Bandwidth and Phase Margin vs.
Temperature, Gain = +2
100
–20
1G
40
0
10k
20
1k
80
60
100M10M1M100k
FREQUENCY – Hz
100
40
0
20
80
60
PHASE MARGIN – Degrees
OPEN-LOOP GAIN – dB
15V SUPPLIES
5V SUPPLIES
PHASE 5V OR
15V SUPPLIES
RL = 1k
TPC 10. Open-Loop Gain and Phase Margin vs.
Frequency
6
3
100 1k 10k
4
5
7
8
LOAD RESISTANCE –
OPEN-LOOP GAIN – V/mV
15V
5V
9
TPC 11. Open-Loop Gain vs. Load Resistance
100
10
100M
30
20
1k100
40
50
60
70
80
90
10M1M100k10k
FREQUENCY – Hz
PSRR – dB
+SUPPLY
–SUPPLY
TPC 12. Power Supply Rejection vs. Frequency
REV. C–6–
AD828
140
60
1k 10M
120
80
10k
100
100k 1M
FREQUENCY – Hz
CMR – dB
TPC 13. Common-Mode Rejection vs. Frequency
30
10
0
100k 1M 100M10M
20
FREQUENCY – Hz
OUTPUT VOLTAGE – V p-p
RL = 1k
RL = 150
TPC 14. Large Signal Frequency Response
10
160200
2
2
0
4
6
8
140120100806040
SETTLING TIME ns
OUTPUT SWING FROM 0 TO V
0.1%1%
1% 0.01%
0.01%
0.1%
4
6
8
10
TPC 15. Output Swing and Error vs. Settling Time
–40
–100
10M
–70
–90
1k
–80
100
–50
–60
1M100k10k
FREQUENCY – Hz
HARMONIC DISTORTION – dB
V
IN
= 1V p-p
GAIN = +2
2
ND
HARMONIC
3
RD
HARMONIC
TPC 16. Harmonic Distortion vs. Frequency
50
010M
30
10
10
20
0
40
1M100k10k1k100
FREQUENCY – Hz
INPUT VOLTAGE NOISE – nV/ Hz
TPC 17. Input Voltage Noise Spectral Density vs.
Frequency
650
250
–60 140
550
350
–40
450
100 120806040200–20
TEMPERATURE – C
SLEW RATE – V/s
TPC 18. Slew Rate vs. Temperature
REV. C
AD828
–7–
FREQUENCY – Hz
GAIN – dB
10
0
–10
100k 1M 100M10M
–2
–4
–6
–8
2
4
6
8
VOUT
VIN
1k
150
AD828
1k
1pF
VS
15V
5V
+5V
0.1dB
FLATNESS
40MHz
43MHz
18MHz
VS = 5V
VS = +5V
VS = 15V
TPC 19. Closed-Loop Gain vs. Frequency
SUPPLY VOLTAGE – V
0.03
0.01
0.02
DIFFERENTIAL PHASE – Degrees
DIFFERENTIAL GAIN – Percent
0.04
15
0.07
0.05
0.06
510
DIFF GAIN
DIFF PHASE
TPC 20. Differential Gain and Phase vs. Supply Voltage
–30
–70
–110
100k 100M10M1M10k
–90
–50
–60
–80
–100
–40
FREQUENCY – Hz
CROSSTALK – dB
RL = 150
RL = 1k
TPC 21. Crosstalk vs. Frequency
FREQUENCY – Hz
GAIN – dB
5
0
–5
100k 1M 100M10M
–1
–2
–3
–4
1
2
3
4
VS = 5V
VS = +5V
VS = 15V
VOUT
VIN
1k
150
AD828
1k
1pF
VS
15V
5V
+5V
0.1dB
FLATNESS
50MHz
25MHz
19MHz
TPC 22. Closed-Loop Gain vs. Frequency, G = –1
FREQUENCY – Hz
GAIN – dB
1.0
0
–1.0
100k 1M 100M10M
–0.2
–0.4
–0.6
–0.8
0.2
0.4
0.6
0.8
VS = 5V
VS = 5V
VS = 15V
TPC 23. Gain Flatness Matching vs. Supply, G = +2
USE GROUND PLANE
PINOUT SHOWN IS FOR MINI-DIP PACKAGE
0.1F
VIN
RL
1/2
AD828
1FVOUT
5
6
7
4
0.1F
1F
1/2
AD828
5V
1
8
3
2
RL
5V
TPC 24. Crosstalk Test Circuit
REV. C–8–
AD828
8
3
2
+V
S
1TEKTRONIX
P6201 FET
PROBE
HP PULSE (LS)
OR FUNCTION (SS)
GENERATOR
1/2
AD828
1k
50
1k
3.3F
0.01F
R
L
V
OUT
3.3F
–V
S
V
IN
TEKTRONIX
7A24
PREAMP
0.01F
4
C
F
TPC 25. Inverting Amplifier Connection
10
90
0%
100
50ns
2V
2V
TPC 26. Inverter Large Signal Pulse Response
5 V
S
,
C
F
= 1 pF, R
L
= 1 k
Ω
10
90
0%
100
10ns
200mV
200mV
TPC 27. Inverter Small Signal Pulse Response
5 V
S
,
C
F
= 1 pF, R
L
= 150
Ω
10
90
0%
100
50ns
5V
5V
TPC 28. Inverter Large Signal Pulse Response
15 V
S
,
C
F
= 1 pF, R
L
= 1 k
Ω
10
90
0%
100
10ns
200mV
200mV
TPC 29. Inverter Small Signal Pulse Response
15 V
S
,
C
F
= 1 pF, R
L
= 1500
Ω
10
90
0%
100
10ns
200mV
200mV
TPC 30. Inverter Small Signal Pulse Response
5 V
S
,
C
F
= 0 pF, R
L
= 150
Ω
REV. C
AD828
–9–
8
3
2
+V
S
1
HP PULSE (LS)
OR FUNCTION (SS)
GENERATOR
TEKTRONIX
P6201 FET
PROBE
1/2
AD828
R
IN
100
50
1k
3.3F
0.01F
R
L
V
OUT
3.3F
–V
S
V
IN
TEKTRONIX
7A24
PREAMP
0.01F
4
C
F
1k
TPC 31. Noninverting Amplifier Connection
10
90
0%
100
50ns
2V
1V
TPC 32. Noninverting Large Signal Pulse Response
5 V
S
, C
F
= 1 pF, R
L
= 1 k
Ω
10
90
0%
100
200mV
100mV 10ns
TPC 33. Noninverting Small Signal Pulse Response
5 V
S
, C
F
= 1 pF, R
L
= 150
Ω
10
90
0%
100
50ns
5V
5V
TPC 34. Noninverting Large Signal Pulse Response
15 V
S
, C
F
= 1 pF, R
L
= 1 k
Ω
10
90
0%
100
200mV
100mV 10ns
TPC 35. Noninverting Small Signal Pulse Response
15 V
S
, C
F
= 1 pF, R
L
= 150
Ω
10
90
0%
100
200mV
100mV 10ns
TPC 36. Noninverting Small Signal Pulse Response
5 V
S
, C
F
= 0 pF, R
L
= 150
Ω
REV. C–10–
AD828
THEORY OF OPERATION
The AD828 is a low cost, dual video operational amplifier
designed to excel in high performance, high output current
video applications.
The AD828 consists of a degenerated NPN differential pair
driving matched PNPs in a folded-cascade gain stage (Figure 4).
The output buffer stage employs emitter followers in a class AB
amplifier that delivers the necessary current to the load while
maintaining low levels of distortion.
The AD828 will drive terminated cables and capacitive loads of
10 pF or less. As the closed-loop gain is increased, the AD828
will drive heavier cap loads without oscillating.
–IN
+IN
OUTPUT
+V
S
–V
S
Figure 4. Simplified Schematic
INPUT CONSIDERATIONS
An input protection resistor (R
IN
in TPC 31) is required in circuits
where the input to the AD828 will be subjected to transient or
continuous overload voltages exceeding the ±6 V maximum dif-
ferential limit. This resistor provides protection for the input
transistors by limiting their maximum base current.
For high performance circuits, the “balancing” resistor should be
used to reduce the offset errors caused by bias current flowing
through the input and feedback resistors. The balancing resistor
equals the parallel combination of R
IN
and R
F
and thus provides
a matched impedance at each input terminal. The offset voltage
error will then be reduced by more than an order of magnitude.
APPLYING THE AD828
The AD828 is a breakthrough dual amp that delivers precision and
speed at low cost with low power consumption. The AD828 offers
excellent static and dynamic matching characteristics, combined
with the ability to drive heavy resistive loads.
As with all high frequency circuits, care should be taken to main-
tain overall device performance as well as their matching. The
following items are presented as general design considerations.
Circuit Board Layout
Input and output runs should be laid out so as to physically
isolate them from remaining runs. In addition, the feedback
resistor of each amplifier should be placed away from the feed-
back resistor of the other amplifier, since this greatly reduces
interamp coupling.
Choosing Feedback and Gain Resistors
To prevent the stray capacitance present at each amplifier’s
summing junction from limiting its performance, the feedback
resistors should be ≤ 1 kΩ. Since the summing junction capaci-
tance may cause peaking, a small capacitor (1 pF to 5 pF) may
be paralleled with R
F
to neutralize this effect. Finally, sockets
should be avoided, because of their tendency to increase interlead
capacitance.
Power Supply Bypassing
Proper power supply decoupling is critical to preserve the
integrity of high frequency signals. In carefully laid out designs,
decoupling capacitors should be placed in close proximity to
the supply pins, while their lead lengths should be kept to a
minimum. These measures greatly reduce undesired inductive
effects on the amplifier’s response.
Though two 0.1 µF capacitors will typically be effective in
decoupling the supplies, several capacitors of different values
can be paralleled to cover a wider frequency range.
PARALLEL AMPS PROVIDE 100 mA TO LOAD
By taking advantage of the superior matching characteristics of the
AD828, enhanced performance can easily be achieved by employ-
ing the circuit in Figure 5. Here, two identical cells are paralleled
to obtain even higher load driving capability than that of a single
amplifier (100 mA min guaranteed). R1 and R2 are included to
limit current flow between amplifier outputs that would arise in
the presence of any residual mismatch.
2
+V
S
V
IN
V
OUT
3
8
1k
R2
5
–V
S
R
L
1/2
AD828
1/2
AD828
1F
0.1F
7
5
6
1
1F
0.1F
4
R1
5
1k
1k
1k
Figure 5. Parallel Amp Configuration
REV. C
AD828
–11–
3
2
1
1/2
AD828
AIN
1/2
AD828
510
2
3 BIN
RZ
100FT
RG59A/U
RZ = 75
1
1/2
AD828
BOUT
5
6
7
6
5
1/2
AD828 AOUT
7
510
510
536
510
510
536
510
RZ
Figure 6. Bidirectional Transmission CKT
Full-Duplex Transmission
Superior load handling capability (50 mA min/amp), high
bandwidth, wide supply voltage range, and excellent crosstalk
rejection makes the AD828 an ideal choice for even the most
demanding high speed transmission applications.
The schematic below shows a pair of AD828s configured to
drive 100 feet of coaxial cable in a full-duplex fashion.
Two different NTSC video signals are simultaneously applied at
A
IN
and B
IN
and are recovered at A
OUT
and B
OUT
, respectively.
This situation is illustrated in Figures 7 and 8. These pictures
clearly show that each input signal appears undisturbed at its out-
put, while the unwanted signal is eliminated at either receiver.
The transmitters operate as followers, while the receivers’ gain
is chosen to take full advantage of the AD828’s unparalleled
CMRR. In practice, this gain is adjusted slightly from its
theoretical value to compensate for cable nonidealities and losses.
R
Z
is chosen to match the characteristic impedance of the
cable employed.
Finally, although a coaxial cable was used, the same topology
applies unmodified to a variety of cables (such as twisted pairs
often used in telephony).
10
90
0%
100
500mV
500mV
10µs
A
IN
B
OUT
Figure 7. A Transmission/B Reception
10
90
0%
100
500mV
500mV
10µs
B
IN
A
OUT
Figure 8. B Transmission/A Reception
A High Performance Video Line Driver
The buffer circuit shown in Figure 9 will drive a back-terminated
75 Ω video line to standard video levels (1 V p-p) with 0.1 dB
gain flatness to 40 MHz with only 0.05° and 0.01% differential
phase and gain at the 3.58 MHz NTSC subcarrier frequency.
This level of performance, which meets the requirements for
high definition video displays and test equipment, is achieved
using only 7 mA quiescent current/amplifier.
2
3
1
1/2
AD828
8
0.1F
4
+15V
–15V
RBT
75
RT
75
VIN
1k
1.0F
0.1F1.0F
1k
75
RT
75
Figure 9. Video Line Driver
C00879–0–6/02(C)
PRINTED IN U.S.A.
–12–
AD828
REV. C
Revision History
Location Page
6/02–Data Sheet changed from REV. B to REV. C.
Renumbered Figures and TPCs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .Global
Changes to Figure 10 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
LOW DISTORTION LINE DRIVER
The AD828 can quickly be turned into a powerful, low distor-
tion line driver (see Figure 10). In this arrangement, the AD828
can comfortably drive a 75 Ω back-terminated cable with a
5 MHz, 2 V p-p input, while achieving the harmonic distortion
performance outlined in the following table.
Configuration 2nd Harmonic
1. No Load –78.5 dBm
2. 150 Ω R
L
Only –63.8 dBm
3. 150 Ω R
L
7.5 Ω R
C
–70.4 dBm
In this application, one half of the AD828 operates at a gain of +2.1
and supplies the current to the load, while the other provides the
overall system gain of +2. This is important for two reasons: the
first is to keep the bandwidth of both amplifiers the same, and
the second is to preserve the AD828’s ability to operate from low
supply voltage. R
C
varies with the load and must be chosen to
satisfy the following equation:
RC = MR
L
where M is defined by [(M + 1) G
S
= G
D
] and G
D
= Driver’s
Gain, G
S
= System Gain.
+V
S
1.1k
R
L
R
C
7.5
75
75
75
0.1F
1/2
AD828
1
8
1F
1k
–V
S
1k
V
IN
1/2
AD828
6
5
7
1k
0.1F
1F
4
3
2
Figure 10. Low Distortion Amplifier
OUTLINE DIMENSIONS
8-Lead Plastic Dual-in-Line Package [PDIP]
(N-8)
Dimensions shown in inches and (millimeters)
SEATING
PLANE
0.0598 (1.52)
0.0150 (0.38)
0.2098
(5.33)
MAX
0.0220 (0.56)
0.0142 (0.36)
0.1598 (4.06)
0.1154 (2.93)
0.0697 (1.77)
0.0453 (1.15)
0.1299
(3.30)
MIN
8
14
5
PIN 1
0.2799 (7.11)
0.2402 (6.10)
0.1000 (2.54)
BSC
0.4299 (10.92)
0.3480 (8.84)
0.1949 (4.95)
0.1154 (2.93)
0.0150 (0.38)
0.0079 (0.20)
0.3248 (8.25)
0.3000 (7.62)
8-Lead Standard Small Outline Package [SOIC]
(R-8)
Dimensions shown in millimeters and (inches)
0.25 (0.0098)
0.19 (0.0075)
1.27 (0.0500)
0.41 (0.0160)
0.50 (0.0196)
0.25 (0.0099) 45
8
0
1.75 (0.0688)
1.35 (0.0532)
SEATING
PLANE
0.25 (0.0098)
0.10 (0.0040)
85
41
5.00 (0.1968)
4.80 (0.1890)
PIN 1
0.1574 (4.00)
0.1497 (3.80)
1.27 (0.0500)
BSC
6.20 (0.2440)
5.80 (0.2284)
0.51 (0.0201)
0.33 (0.0130)
COPLANARITY
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
COMPLIANT TO JEDEC STANDARDS MS-012 AA
