LM2614BTL - National Semiconductor

LM2614

LM2614 400mA Sub-Miniature Adjustable DC-DC Converter Optimized for RF

Power Amplifiers

Literature Number: SNVS184A

LM2614

OBSOLETE

August 25, 2011

400mA Sub-Miniature Adjustable DC-DC Converter

Optimized for RF Power Amplifiers

General Description

The LM2614 DC-DC converter is optimized for powering RF

power amplifiers (PAs) from a single Lithium-Ion cell. It steps

down an input voltage of 2.8V to 5.5V to an output of 1.0V to

3.6V at up to 400mA (300mA for B grade). Output voltage is

set using an analog input to VCON in the application circuit.

The device offers three modes for mobile phones and similar

RF PA applications. Fixed-frequency PWM mode minimizes

RF interference. A SYNC input allows synchronizing the

switching frequency in a range of 500kHz to 1MHz. Low cur-

rent hysteretic PFM mode reduces quiescent current to

160µA (typ.). Shutdown mode turns the device off and re-

duces battery consumption to 0.02µA (typ.).

Current limit and thermal shutdown features protect the de-

vice and system during fault conditions.

The LM2614 is available in a 10 bump micro SMD package.

This packaging uses National's chip-scale micro SMD tech-

nology and offers the smallest possible size. A high switching

frequency (600kHz) allows use of tiny surface-mount compo-

nents.

The LM2614 can be dynamically controlled for output voltage

changes from 1.0V to 3.6V in <30µs. The device features ex-

ternal compensation to tailor the response to a wide range of

operating conditions.

Key Specifications

■Operates from a single LiION cell (2.8V to 5.5V)

■Adjustable output voltage (1.0V to 3.6V)

■±1% DC feedback voltage precision

■400mA maximum load capability(300mA for B grade)

■600µA typ PWM mode quiescent current

■0.02µA typ shutdown current

■600kHz PWM switching frequency

■SYNC input for PWM mode frequency synchronization

from 500kHz to 1MHz

■High efficiency (96% typ at 3.9VIN, 3.6VOUT and 200mA)

in PWM mode from internal synchronous rectification

■100% Maximum Duty Cycle for Lowest Dropout

Features

■Sub-miniature 10-bump thin micro SMD package

■Uses small ceramic capacitors

■5mV typ PWM mode output voltage ripple(COUT = 22µF)

■Internal soft start

■Current overload protection

■Thermal Shutdown

■External compensation

Applications

■Mobile Phones

■Hand-Held Radios

■RF PC Cards

■Battery Powered RF Devices

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LM2614 400mA Sub-Miniature Adjustable DC-DC Converter Optimized for RF Power Amplifiers

Typical Application Circuits

20036701

FIGURE 1. Typical Circuit for Powering RF Power Amplifiers

20036702

FIGURE 2. Typical Circuit for 2.5V Output Voltage

20036703

FIGURE 3. Typical Circuit for 1.5V Output Voltage

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LM2614

Connection Diagrams

10-Bump micro SMD Package

20036704

Top View 20036705

Bottom View

Ordering Information

Order Number Package Type NSC Package

Marking (*) Supplied As

LM2614ATL

10-bump Wafer Level Chip Scale

(micro SMD)

XYTT S50A 250 Tape and Reel

LM2614BTL XYTT S50B 250 Tape and Reel

LM2614ATLX XYTT S50A 3000 Tape and Reel

LM2614BTLX XYTT S50B 3000 Tape and Reel

(*) XY - denotes the date code marking (2 digit) in production

(*) TT - refers to die run/lot traceability for production

(*) S - product line designator

Package markings may change over the course of production.

Pin Descriptions

Pin Number Pin Name Function

A1 FB Feedback Analog Input.

B1 EANEG Inverting input of error amplifier

C1 EAOUT Output of error amplifier

D1 SYNC/MODE Synchronization Input. Use this digital input for frequency selection or modulation control. Set:

SYNC/MODE = high for low-noise 600kHz PWM mode

SYNC/MODE = low for low-current PFM mode

SYNC/MODE = a 500kHz–1MHz external clock for synchronization in PWM mode. (See

Synchronization and Operating Modes in the Device Information section.)

D2 EN Enable Input. Set this Schmitt trigger digital input high for normal operation. For shutdown, set

low. Set EN low during system power-up and other low supply voltage conditions. (See

Shutdown Mode in the Device Information section.)

D3 PGND Power Ground

C3 SW Switching Node connection to the internal PFET switch and NFET synchronous rectifier.

Connect to an inductor with a saturation current rating that exceeds the max Switch Peak

Current Limit of the LM2614.

B3 PVIN Power Supply Voltage Input to the internal PFET switch. Connect to the input filter capacitor.

A3 VDD Analog Supply Input. If board layout is not optimum, an optional 0.1µF ceramic capacitor is

suggested.

A2 SGND Analog and Control Ground

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LM2614

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

PVIN, VDD to SGND −0.2V to +6V

PGND to SGND, PVIN to VDD −0.2V to +0.2V

EN, EAOUT, EANEG, SYNC/MODE

to SGND −0.2V to +6V

FB, SW (GND −0.2V) to

(VDD +0.2V)

Storage Temperature Range −45°C to +150°C

Lead Temperature

(Soldering, 10 sec.) 260°C

Junction Temperature (Note 2) −25°C to +125°C

Minimum ESD Rating ±2 kV

(Human Body Model, C = 100 pF, R = 1.5 kΩ)

Thermal Resistance (θJA) (Note 3)140°C/W

Electrical Characteristics

Specifications with standard typeface are for TA = TJ = 25°C, and those in boldface type apply over the full Operating Temperature

Range of TA = TJ = −25°C to +85°C. Unless otherwise specified, PVIN = VDD = EN = SYNC/MODE = 3.6V.

Symbol Parameter Conditions Min Typ Max Units

VIN Input Voltage Range PVIN = VDD = VIN (Note 4)2.8 3.6 5.5 V

VFB Feedback Voltage 1.485 1.50 1.515 V

VHYST PFM Comparator Hysteresis

Voltage

PFM Mode (SYNC/MODE = 0V)

(Note 5) 24 mV

ISHDN Shutdown Supply Current VIN = 3.6V, EN = 0V 0.02 3µA

IQ1_PWM DC Bias Current into VDD SYNC/MODE = VIN

FB = 2V 600 725 µA

IQ2_PFM SYNC/MODE = 0V

FB = 2V 160 195 µA

RDSON (P) Pin-Pin Resistance for P FET 395 550 mΩ

RDSON (N) Pin-Pin Resistance for N FET 330 500 mΩ

RDSON (TC) FET Resistance Temperature

Coefficient

0.5 %/C

ILIM Switch Peak Current Limit (Note

LM2614ATL 510 690 850 mA

LM2614BTL 400 690 980

VIH Logic High Input, EN, SYNC/

MODE

0.95 1.3 V

VIL Logic Low Input, EN, SYNC/

MODE

0.4 0.80 V

FSYNC SYNC/MODE Clock Frequency

Range

(Note 7)500 1000 kHz

FOSC Internal Oscillator Frequency LM2614ATL, PWM Mode 468 600 732 kHz

LM2614BTL, PWM Mode 450 600 750

Tmin Minimum ON-Time of PFET

Switch in PWM Mode

200 ns

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

functional, but device specifications may not be guaranteed. For guaranteed specifications and associated test conditions, see the Min and Max limits and

Conditions in the Electrical Characteristics table. Typical (typ) specifications are mean or average values at 25°C and are not guaranteed.

Note 2: Thermal shutdown will occur if the junction temperature exceeds 150°C.

Note 3: Thermal resistance specified with 2 layer PCB (0.5/0.5 oz. cu).

Note 4: The LM2614 is designed for mobile phone applications where turn-on after system power-up is controlled by the system controller. Thus, it should be

kept in shutdown by holding the EN pin low until the input voltage exceeds 2.8V.

Note 5: The hysteresis voltage is the minimum voltage swing on the FB pin that causes the internal feedback and control circuitry to turn the internal PFET switch

on and then off during PFM mode. When resistor dividers are used like in the operating circuit of Figure 4, the hysteresis at the output will be the value of the

hysteresis at the feedback pin times the resistor divider ratio. In this case, 24mV (typ) x ((46.4k + 33.2k)/33.2k).

Note 6: Current limit is built-in, fixed, and not adjustable. If the current limit is reached while the voltage at the FB pin is pulled below 0.7V, the internal PFET

switch turns off for 2.5µs to allow the inductor current to diminish.

Note 7: SYNC driven with an external clock switching between VIN and GND. When an external clock is present at SYNC; the IC is forced to be in PWM mode

at the external clock frequency. The LM2614 synchronizes to the rising edge of the external clock.

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LM2614

Typical Performance Characteristics

LM2614ATL, Circuit of Figure 4, VIN = 3.6V, TA = 25°C, unless otherwise noted.

Quiescent Supply Current vs Supply Voltage

20036708

Shutdown Quiescent Current vs Temperature

(Circuit in Figure 3)

20036722

Output Voltage vs Supply Voltage

(VOUT = 1.0V, PWM MODE)

20036724

Output Voltage vs Supply Voltage

(VOUT = 1.5V, PWM MODE)

20036709

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LM2614

Output Voltage vs Output Current

(VOUT = 1.0V, PWM MODE)

20036710

Output Voltage vs Output Current

(VOUT = 1.5V, PWM MODE)

20036711

Output Voltage vs Output Current

(VOUT = 3.6V, PWM MODE)

20036732

Dropout Voltage vs Output Current

(VOUT = 3.6V, PWM MODE)

20036712

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LM2614

Switching Frequency vs Temperature

(Circuit in Figure 3, PWM MODE)

20036723

Feedback Bias Current vs Temperature

(Circuit in Figure 3)

20036731

Efficiency vs Output Current

(VOUT = 1.0V, PWM MODE)

20036713

Efficiency vs Output Current

(VOUT = 1.0V, PWM MODE, with Diode)

20036714

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LM2614

Efficiency vs Output Current

(VOUT = 1.5V, PWM MODE)

20036715

Efficiency vs Output Current

(VOUT = 1.5V, PWM MODE, with Diode)

20036716

Efficiency vs Output Current

(VOUT = 3.6V, PWM MODE)

20036717

Efficiency vs Output Current

(VOUT = 3.6V, PWM MODE, with Diode)

20036718

Efficiency vs Output Voltage

(PWM MODE, with Diode)

20036730

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LM2614

Device Information

The LM2614 is a simple, step-down DC-DC converter opti-

mized for powering RF power amplifiers (PAs) in mobile

phones, portable communicators, and similar battery pow-

ered RF devices. It is designed to allow the RF PA to operate

at maximum efficiency over a wide range of power levels from

a single LiION battery cell. It is based on a current-mode buck

architecture, with synchronous rectification in PWM mode for

high efficiency. It is designed for a maximum load capability

of 400mA (300mA for B grade) in PWM mode. Maximum load

range may vary from this depending on input voltage, output

voltage and the inductor chosen.

The device has all three of the pin-selectable operating

modes required for powering RF PAs in mobile phones and

other sophisticated portable devices with complex power

management needs. Fixed-frequency PWM operation offers

full output current capability at high efficiency while minimiz-

ing interference with sensitive IF and data acquisition circuits.

During standby operation, hysteretic PFM mode reduces qui-

escent current to 160µA typ. to maximize battery life. Shut-

down mode turns the device off and reduces battery

consumption to 0.02µA (typ).

DC PWM mode feedback voltage precision is ±1%. Efficiency

is typically 96% for a 200mA load with 3.6V output, 3.9V input.

The efficiency can be further increased by using a schottky

diode like MBRM120 as shown in Figure 4. PWM mode qui-

escent current is 600µA typ. The output voltage is dynamically

programmable from 1.0V to 3.6V by adjusting the voltage on

the VCON at the external feedback resistors. This ensures

longer battery life by being able to change the PA supply volt-

age dynamically depending on its transmitting power.

Additional features include soft-start, current overload pro-

tection, over voltage protection and thermal shutdown pro-

tection.

The LM2614 is constructed using a chip-scale 10-pin thin mi-

cro SMD package. This package offers the smallest possible

size, for space-critical applications such as cell phones,

where board area is an important design consideration. Use

of a high switching frequency (600kHz) reduces the size of

external components. Board area required for implementation

is only 0.58in2 (375mm2).

Use of a micro-SMD package requires special design con-

siderations for implementation. (See Micro SMD Package

Assembly and Use in the Application Information section.) Its

fine bump-pitch requires careful board design and precision

assembly equipment.

20036706

FIGURE 4. Typical Operating Circuit

CIRCUIT OPERATION

Referring to Figure 4, Figure 5, Figure 6 and Figure 7,

the LM2614 operates as follows. During the first part of each

switching cycle, the control block in the LM2614 turns on the

internal PFET switch. This allows current to flow from the input

through the inductor to the output filter capacitor and load. The

inductor limits the current to a ramp with a slope of (VIN–

VOUT)/L, by storing energy in a magnetic field. During the sec-

ond part of each cycle, the controller turns the PFET switch

off, blocking current flow from the input, and then turns the

NFET synchronous rectifier on. In response, the inductor's

magnetic field collapses, generating a voltage that forces cur-

rent from ground through the synchronous rectifier to the

output filter capacitor and load. As the stored energy is trans-

ferred back into the circuit and depleted, the inductor current

ramps down with a slope of VOUT/L. If the inductor current

reaches zero before the next cycle, the synchronous rectifier

is turned off to prevent current reversal. The output filter ca-

pacitor stores charge when the inductor current is high, and

releases it when low, smoothing the voltage across the load.

The output voltage is regulated by modulating the PFET

switch on time to control the average current sent to the load.

The effect is identical to sending a duty-cycle modulated rect-

angular wave formed by the switch and synchronous rectifier

at SW to a low-pass filter formed by the inductor and output

filter capacitor. The output voltage is equal to the average

voltage at the SW pin.

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LM2614

20036707

FIGURE 5. Simplified Functional Diagram

PWM OPERATION

While in PWM (Pulse Width Modulation) mode, the output

voltage is regulated by switching at a constant frequency and

then modulating the energy per cycle to control power to the

load. Energy per cycle is set by modulating the PFET switch

on-time pulse-width to control the peak inductor current. This

is done by comparing the signal from the current-sense am-

plifier with a slope compensated error signal from the voltage-

feedback error amplifier. At the beginning of each cycle, the

clock turns on the PFET switch, causing the inductor current

to ramp up. When the current sense signal ramps past the

error amplifier signal, the PWM comparator turns off the PFET

switch and turns on the NFET synchronous rectifier, ending

the first part of the cycle. If an increase in load pulls the output

voltage down, the error amplifier output increases, which al-

lows the inductor current to ramp higher before the compara-

tor turns off the PFET. This increases the average current

sent to the output and adjusts for the increase in the load.

Before going to the PWM comparator, the error signal is

summed with a slope compensation ramp from the oscillator

for stability of the current feedback loop. During the second

part of the cycle, a zero crossing detector turns off the NFET

synchronous rectifier if the inductor current ramps to zero.

The minimum on time of the PFET in PWM mode is about

200ns.

PWM Mode Switching Waveform

20036725

A: Inductor Current, 500mA/div

B: SW Pin, 2V/div

C: VOUT, 10mV/div, AC Coupled

FIGURE 6.

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LM2614

PFM Mode Switching Waveform

20036726

A: Inductor Current, 500mA/div

B: SW Pin, 2V/div

C: VOUT, 50mV/div, AC Coupled

FIGURE 7.

PFM OPERATION

Connecting the SYNC/MODE to SGND sets the LM2614 to

hysteretic PFM operation. While in PFM (Pulse Frequency

Modulation) mode, the output voltage is regulated by switch-

ing with a discrete energy per cycle and then modulating the

cycle rate, or frequency, to control power to the load. This is

done by using an error comparator to sense the output volt-

age. The device waits as the load discharges the output filter

capacitor, until the output voltage drops below the lower

threshold of the PFM error-comparator. Then the device ini-

tiates a cycle by turning on the PFET switch. This allows

current to flow from the input, through the inductor to the out-

put, charging the output filter capacitor. The PFET is turned

off when the output voltage rises above the regulation thresh-

old of the PFM error comparator. Thus, the output voltage

ripple in PFM mode is proportional to the hysteresis of the

error comparator.

In PFM mode, the device only switches as needed to service

the load. This lowers current consumption by reducing power

consumed during the switching action in the circuit, due to

transition losses in the internal MOSFETs, gate drive cur-

rents, eddy current losses in the inductor, etc. It also improves

light-load voltage regulation. During the second half of the

cycle, the intrinsic body diode of the NFET synchronous rec-

tifier conducts until the inductor current ramps to zero.

OPERATING MODE SELECTION

The LM2614 is designed for digital control of the operating

modes by the system controller. This prevents the spurious

switch over from low-noise PWM mode between transmission

intervals in mobile phone applications that can occur in other

products.

The SYNC/MODE digital input pin is used to select the oper-

ating mode. Setting SYNC/MODE high (above 1.3V) selects

600kHz current-mode PWM operation. PWM mode is opti-

mized for low-noise, high-power operation for use when the

load is active. Setting SYNC/MODE low (below 0.4V) selects

hysteretic voltage-mode PFM operation. PFM mode is opti-

mized for reducing power consumption and extending battery

life when the load is in a low-power standby mode. In PFM

mode, quiescent current into the VDD pin is 160µA typ. In con-

trast, PWM mode VDD-pin quiescent current is 600µA typ.

PWM operation is intended for use with loads of 50mA or

more, when low noise operation is desired. Below 100mA,

PFM operation can be used to allow precise regulation, and

reduced current consumption. However, it should be noted

that for PA applications the PFM mode need not be used as

output voltage slew rates are of more concern to the system

designer. The LM2614 has an over-voltage feature that pre-

vents the output voltage from rising too high, when the device

is left in PWM mode under low-load conditions. See Over-

voltage Protection, for more information.

Switch modes with the SYNC/MODE pin, using a signal with

a slew rate faster than 5V/100µs. Use a comparator, Schmitt

trigger or logic gate to drive the SYNC/MODE pin. Do not

leave the pin floating or allow it to linger between thresholds.

These measures will prevent output voltage errors in re-

sponse to an indeterminate logic state. The LM2614 switches

on each rising edge of SYNC. Ensure a minimum load to keep

the output voltage in regulation when switching modes fre-

quently.

FREQUENCY SYNCHRONIZATION

The SYNC/MODE input can also be used for frequency syn-

chronization. During synchronization, the LM2614 initiates

cycles on the rising edge of the clock. When synchronized to

an external clock, it operates in PWM mode. The device can

synchronize to a 50% duty-cycle clock over frequencies from

500kHz to 1MHz. If a different duty cycle is used other than

50% the range for acceptable duty cycles are 30% to 70%.

Use the following waveform and duty cycle guidelines when

applying an external clock to the SYNC/MODE pin. Clock un-

der/overshoot should be less than 100mV below GND or

above VDD. When applying noisy clock signals, especially

sharp edged signals from a long cable during evaluation, ter-

minate the cable at its characteristic impedance and add an

RC filter to the SYNC pin, if necessary, to soften the slew rate

and over/undershoot. Note that sharp edged signals from a

pulse or function generator can develop under/overshoot as

high as 10V at the end of an improperly terminated cable.

OVERVOLTAGE PROTECTION

The LM2614 has an over-voltage comparator that prevents

the output voltage from rising too high when the device is left

in PWM mode under low-load conditions. When the output

voltage rises by about 100mV (Figure 3) over its regulation

threshold, the OVP comparator inhibits PWM operation to

skip pulses until the output voltage returns to the regulation

threshold. When resistor dividers are used the OVP threshold

at the output will be the value of the threshold at the feedback

pin times the resistor divider ratio. In over voltage protection,

output voltage and ripple will increase.

SHUTDOWN MODE

Setting the EN digital input pin low (<0.4V) places the LM2614

in a 0.02µA (typ) shutdown mode. During shutdown, the PFET

switch, NFET synchronous rectifier, reference, control and

bias circuitry of the LM2614 are turned off. Setting EN high

enables normal operation. While turning on, soft start is acti-

vated.

EN should be set low to turn off the LM2614 during system

power-up and undervoltage conditions when the supply is

less than the 2.8V minimum operating voltage. The LM2614

is designed for compact portable applications, such as mobile

phones. In such applications, the system controller deter-

mines power supply sequencing. Although the LM2614 is

typically well behaved at low input voltages, this is not guar-

anteed.

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LM2614

INTERNAL SYNCHRONOUS RECTIFICATION

While in PWM mode, the LM2614 uses an internal NFET as

a synchronous rectifier to reduce rectifier forward voltage

drop and associated power loss. Synchronous rectification

provides a significant improvement in efficiency whenever the

output voltage is relatively low compared to the voltage drop

across an ordinary rectifier diode.

The internal NFET synchronous rectifier is turned on during

the inductor current down slope during the second part of

each cycle. The synchronous rectifier is turned off prior to the

next cycle, or when the inductor current ramps to zero at light

loads. The NFET is designed to conduct through its intrinsic

body diode during transient intervals before it turns on, elim-

inating the need for an external diode.

CURRENT LIMITING

A current limit feature allows the LM2614 to protect itself and

external components during overload conditions. In PWM

mode cycle-by-cycle current limit is normally used. If an ex-

cessive load pulls the voltage at the feedback pin down to

approximately 0.7V, then the device switches to a timed cur-

rent limit mode. In timed current limit mode the internal P-FET

switch is turned off after the current comparator trips and the

beginning of the next cycle is inhibited for 2.5µs to force the

instantaneous inductor current to ramp down to a safe value.

Timed current limit mode prevents the loss of current control

seen in some products when the voltage at the feedback pin

is pulled low in serious overload conditions.

DYNAMICALLY ADJUSTABLE OUTPUT VOLTAGE

The LM2614 can be used to provide dynamically adjustable

output voltage by using external feedback resistors. The out-

put can be varied from 1.0V to 3.6V in less than 30µs by using

an analog control signal (VCON) at the external feedback re-

sistors. This feature is useful in PA applications where peak

power is needed only when the handset is far away from the

base station or when data is being transmitted. In other in-

stances the transmitting power can be reduced and hence the

supply voltage to the PA can be reduced helping maintain

longer battery life. See Setting the Output Voltage in the Ap-

plication Information section for further details.

In dropout conditions the output voltage is VIN − IOUT (Rdc +

RDSON (P)) where Rdc is the series resistance of the inductor

and RDSON (P) is the on resistance of the PFET.

VCON Transient Response

(Circuit in Figure 4)

20036719

FIGURE 8.

VCON Transient Response in Dropout

(Circuit in Figure 4)

20036720

FIGURE 9.

Load Transient Response

(Circuit in Figure 3)

20036727

FIGURE 10.

Line Transient Response

(Circuit in Figure 3)

20036728

FIGURE 11.

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LM2614

SOFT-START

The LM2614 has soft start to reduce current inrush during

power-up and startup. This reduces stress on the LM2614

and external components. It also reduces startup transients

on the power source. Soft start is implemented by ramping up

the reference input to the error amplifier of the LM2614 to

gradually increase the output voltage.

THERMAL SHUTDOWN PROTECTION

The LM2614 has a thermal shutdown protection function to

protect itself from short-term misuse and overload conditions.

When the junction temperature exceeds 150°C the device

turns off the output stage and when the temperature drops

below 130°C it initiates a soft start cycle. Prolonged operation

in thermal shutdown conditions may damage the device and

is considered bad practice.

Application Information

SETTING THE OUTPUT VOLTAGE

The LM2614 can be used with external feedback resistors

and an analog signal to vary the output voltage. Select an

output voltage from 1.0V to 3.6V by setting the voltage on the

VCON as directed in Table 1.

TABLE 1. Output Voltage Selection

VCON (V) VOUT (V)

VCON = 0V VFB (1+R1/R2)

VCON > 0V VFB (1+R1/R2)−VCON (R1/R2)

Refer to Figure 12 for the relation between VOUT and VCON.

VOUT vs VCON

(Circuit in Figure 4)

20036721

FIGURE 12.

When the control voltage is between 1.85V and 0V, the output

voltage will vary in a monotonic fashion with respect to the

voltage on the control pin as per the equation in Table 1. Se-

lect the value of R2 to allow at least 100 times the feedback

pin bias current to flow through it.

EXTERNAL COMPENSATION

The LM2614 uses external components connected to the

EANEG and EAOUT pins to compensate the regulator (Fig-

ure 4). Typically, all that is required is a series connection of

one capacitor (C4) and one resistor (R3). A capacitor (C5) can

be connected across the EANEG and EAOUT pins to improve

the noise immunity of the loop. C5 reacts with R3 to give a

high frequency pole. C4 reacts with the high open loop gain

of the error amplifier and the resistance at the EANEG pin to

create the dominant pole for the system, while R3 and C4

react to create a zero in the frequency response. The pole

rolls off the loop gain, to give a bandwidth somewhere be-

tween 10kHz and 50kHz, this avoids a 100kHz parasitic pole

contributed by the current mode controller. Typical values in

the 220pF to 1nF (C4) range are recommended to create a

pole on the order of 10Hz or less.

The next dominant pole in the system is formed by the output

capacitance (C2) and the parallel combination of the load re-

sistance and the effective output resistance of the regulator.

This combined resistance (Ro) is dominated by the small sig-

nal output resistance, which is typically in the range of 3Ω to

15Ω. The exact value of this resistance, and therefore this

load pole depends on the steady state duty cycle and the in-

ternal ramp value. Ideally we want the zero formed by R3 and

C4 to cancel this load pole, such that R3=RoC2/C4. Due to

the large variation in Ro, this ideal case can only be achieved

at one operating condition. Therefore a compromise of about

5Ω for Ro should be used to determine a starting value for

R3. This value can then be optimized on the bench to give the

best transient response to load changes and changes in

VCON, under all conditions. Typical values are 10pF for C5

and 220pF to 470pF for C4, to ensure good response from

dropout conditions to VOUT(min).

INDUCTOR SELECTION

Use a 10µH inductor with saturation current rating higher than

the peak current rating of the device. The inductor's resis-

tance should be less than 0.3Ω for good efficiency. Table 2

lists suggested inductors and suppliers.

TABLE 2. Suggested Inductors and Their Suppliers

Part Number Vendor Phone FAX

DO1608C-103 Coilcraft 847-639-6400 847-639-1469

P1174.103T Pulse 858-674-8100 858-674-8262

ELL6RH100M Panasonic 714-373-7366 714-373-7323

CDRH5D18-100 Sumida 847-956-0666 847-956-0702

P0770.103T Pulse 858-674-8100 858-674-8262

13 www.national.com

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LM2614

For low-cost applications, an unshielded inductor is suggest-

ed. For noise critical applications, a toroidal or shielded in-

ductor should be used. A good practice is to lay out the board

with footprints accommodating both types for design flexibili-

ty. This allows substitution of a low-noise shielded inductor,

in the event that noise from low-cost unshielded models is

unacceptable.

The saturation current rating is the current level beyond which

an inductor loses its inductance. Different manufacturers

specify the saturation current rating differently. Some specify

saturation current point to be when inductor value falls 30%

from its original value, others specify 10%. It is always better

to look at the inductance versus current curve and make sure

the inductor value doesn’t fall below 30% at the peak current

rating of the LM2614. Beyond this rating, the inductor loses

its ability to limit current through the PWM switch to a ramp.

This can cause poor efficiency, regulation errors or stress to

DC-DC converters like the LM2614. Saturation occurs when

the magnetic flux density from current through the windings

of the inductor exceeds what the inductor’s core material can

support with a corresponding magnetic field.

CAPACITOR SELECTION

Use a 4.7µF or 10µF ceramic input capacitor. A 10µF ceramic

input capacitor is recommended if the PA represents a load

<14Ω. Use a 4.7µF ceramic output capacitor for getting faster

slew rates for output voltages from VOUT (min) to VOUT (max).

Use X7R or X5R types, do not use Y5V. The rise

time for the voltage from VOUT (min) to VOUT (max) depends

on the slew rate of the error amp, switch peak current limit

and the value of the output capacitor. The time for the output

to change from VOUT (max) to VOUT (min) depends on RLOAD

and COUT. Use of tantalum capacitors is not recommended.

Ceramic capacitors provide an optimal balance between

small size, cost, reliability and performance for cell phones

and similar applications. A 22µF ceramic output capacitor can

be used in applications requiring fixed output voltages and/or

increased tolerance to heavy load transients. A 10µF ceramic

output capacitor can be used in applications where the worst

case load transient step is less than 200mA. Table 3 lists

suggested capacitors and suppliers.

The input filter capacitor supplies current to the PFET switch

of the LM2614 in the first part of each cycle and reduces volt-

age ripple imposed on the input power source. The output

filter capacitor smoothes out current flow from the inductor to

the load, helps maintain a steady output voltage during tran-

sient load changes and reduces output voltage ripple. These

capacitors must be selected with sufficient capacitance and

sufficiently low ESR to perform these functions. Parallel com-

binations of smaller value ceramic capacitors can also be

used on the output as long as the combined value is at least

4.7µF for the application circuit in Figure 1.

The ESR, or equivalent series resistance, of the filter capac-

itors is a major factor in voltage ripple.

TABLE 3. Suggested Capacitors and Their Suppliers

Model Type Vendor Phone FAX

C1, C2 (Input or Output Filter Capacitor)

JMK212BJ475MG Ceramic Taiyo-Yuden 847-925-0888 847-925-0899

LMK316BJ475ML Ceramic Taiyo-Yuden 847-925-0888 847-925-0899

C2012X5R0J475K Ceramic TDK 847-803-6100 847-803-6296

JMK325BJ226MM Ceramic Taiyo-Yuden 847-925-0888 847-925-0899

JMK212BJ106MG Ceramic Taiyo-Yuden 847-925-0888 847-925-0899

micro SMD PACKAGE ASSEMBLY AND USE

Use of the micro SMD package requires specialized board

layout, precision mounting and careful reflow techniques, as

detailed in National Semiconductor Application Note

AN-1112. Refer to the section Surface Mount Technology

(SMT) Assembly Considerations. For best results in assem-

bly, alignment ordinals on the PC board should be used to

facilitate placement of the device.

The pad style used with micro SMD package must be the

NSMD (non-solder mask defined) type. This means that the

solder-mask opening is larger than the pad size. This pre-

vents a lip that otherwise forms if the solder-mask and pad

overlap, from holding the device off the surface of the board

and interfering with mounting. See Application Note AN-1112

for specific instructions how to do this.

The 10-Bump package used for the LM2614 has 300 micron

solder balls and requires 10.82mil pads for mounting on the

circuit board. The trace to each pad should enter the pad with

a 90° entry angle to prevent debris from being caught in deep

corners. Initially, the trace to each pad should be 6–7mil wide,

for a section approximately 6mil long, as a thermal relief. Then

each trace should neck up or down to its optimal width. The

important criterion is symmetry. This ensures the solder

bumps on the LM2614 reflow evenly and that the device sol-

ders level to the board. In particular, special attention must be

paid to the pads for bumps D3–B3. Because PGND and PVIN

are typically connected to large copper planes, inadequate

thermal reliefs can result in late or inadequate reflow of these

bumps.

The micro SMD package is optimized for the smallest possi-

ble size in applications with red or infrared opaque cases.

Because the micro SMD package lacks the plastic encapsu-

lation characteristic of larger devices, it is vulnerable to light.

Backside metalization and/or epoxy coating, along with front-

side shading by the printed circuit board, reduce this sensi-

tivity.

BOARD LAYOUT CONSIDERATIONS

PC board layout is an important part of DC-DC converter de-

sign. Poor board layout can disrupt the performance of a DC-

DC converter and surrounding circuitry by contributing to EMI,

ground bounce, and resistive voltage loss in the traces. These

can send erroneous signals to the DC-DC converter IC, re-

sulting in poor regulation or instability. Poor layout can also

result in reflow problems leading to poor solder joints between

the micro SMD package and board pads. Poor solder joints

can result in erratic or degraded performance.

Good layout for the LM2614 can be implemented by following

a few simple design rules.

1. Place the LM2614 on 10.82 mil (10.82/1000 in.) pads. As

a thermal relief, connect to each pad with a 7 mil wide,

approximately 7 mil long traces, and then incrementally

www.national.com 14

200367 Version 2 Revision 2 Print Date/Time: 2011/08/25 17:02:26

LM2614

increase each trace to its optimal width. The important

criterion is symmetry to ensure the solder bumps on the

LM2614 reflow evenly (see micro SMD Package

Assembly and Use).

2. Place the LM2614, inductor and filter capacitors close

together and make the traces short. The traces between

these components carry relatively high switching

currents and act as antennas. Following this rule reduces

radiated noise. Place the capacitors and inductor within

0.2 in. (5 mm) of the LM2614.

3. Arrange the components so that the switching current

loops curl in the same direction. During the first half of

each cycle, current flows from the input filter capacitor,

through the LM2614 and inductor to the output filter

capacitor and back through ground, forming a current

loop. In the second half of each cycle, current is pulled

up from ground, through the LM2614 by the inductor, to

the output filter capacitor and then back through ground,

forming a second current loop. Routing these loops so

the current curls in the same direction prevents magnetic

field reversal between the two half-cycles and reduces

radiated noise.

4. Connect the ground pins of the LM2614, and filter

capacitors together using generous component-side

copper fill as a pseudo-ground plane. Then, connect this

to the ground-plane (if one is used) with several vias. This

reduces ground-plane noise by preventing the switching

currents from circulating through the ground plane. It also

reduces ground bounce at the LM2614 by giving it a low-

impedance ground connection.

5. Use wide traces between the power components and for

power connections to the DC-DC converter circuit. This

reduces voltage errors caused by resistive losses across

the traces.

6. Route noise sensitive traces, such as the voltage

feedback path, away from noisy traces between the

power components. The voltage feedback trace must

remain close to the LM2614 circuit and should be routed

directly from VOUT at the output capacitor and should be

routed opposite to noise components. This reduces EMI

radiated onto the DC-DC converter's own voltage

feedback trace.

7. Place noise sensitive circuitry, such as radio IF blocks,

away from the DC-DC converter, CMOS digital blocks

and other noisy circuitry. Interference with noise-

sensitive circuitry in the system can be reduced through

distance.

In mobile phones, for example, a common practice is to place

the DC-DC converter on one corner of the board, arrange the

CMOS digital circuitry around it (since this also generates

noise), and then place sensitive preamplifiers and IF stages

on the diagonally opposing corner. Often, the sensitive cir-

cuitry is shielded with a metal pan and power to it is post-

regulated to reduce conducted noise, using low-dropout

linear regulators.

15 www.national.com

200367 Version 2 Revision 2 Print Date/Time: 2011/08/25 17:02:26

LM2614

Physical Dimensions inches (millimeters) unless otherwise noted

NOTES: UNLESS OTHERWISE SPECIFIED

1. EPOXY COATING

2. 63Sn/37Pb EUTECTIC BUMP

3. RECOMMEND NON-SOLDER MASK DEFINED LANDING PAD.

4. PIN A1 IS ESTABLISHED BY LOWER LEFT CORNER WITH RESPECT TO TEXT ORIENTATION.

5. XXX IN DRAWING NUMBER REPRESENTS PACKAGE SIZE VARIATION WHERE X1 IS PACKAGE WIDTH, X2 IS PACKAGE LENGTH AND X3 IS PACK-

AGE HEIGHT.

6. REFERENCE JEDEC REGISTRATION MO-211. VARIATION BD.

10-Bump micro SMD Package

NS Package Number TLP106WA

The dimensions for X1, X2 and X3 are as given:

X1 = 2.250 ±0.030 mm

X2 = 2.504 ±0.030 mm

X3 = 0.600 ±0.075 mm

www.national.com 16

200367 Version 2 Revision 2 Print Date/Time: 2011/08/25 17:02:26

LM2614

Notes

17 www.national.com

200367 Version 2 Revision 2 Print Date/Time: 2011/08/25 17:02:26

LM2614

Notes

LM2614 400mA Sub-Miniature Adjustable DC-DC Converter Optimized for RF Power Amplifiers

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