January 2012 Doc ID 022671 Rev 1 1/33
33
L6564T
10-pin transition-mode PFC controller
Features
■Guaranteed for extreme temperature range
(outdoor)
■Fast “bi-directional” input voltage feedforward
(1/V2 correction)
■Accurate adjustable output overvoltage
protection
■Protection against feedback loop
disconnection (latched shutdown)
■Inductor saturation protection
■AC brownout detection
■Low (≤ 100 µA) startup current
■6 mA max. operating bias current
■1% (@ TJ = 25 °C) internal reference voltage
■-600/+800 mA totem pole gate driver with
active pull-down during UVLO
■SSOP10 package
Applications
■PFC pre-regulators for:
– High-end AC-DC adapter/charger
– Desktop PC, server, web server
– IEC61000-3-2 or JEITA-MITI compliant
SMPS
■SMPS for LED luminaires
Figure 1. Block diagram
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Contents L6564T
2/33 Doc ID 022671 Rev 1
Contents
1 Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2 Maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
2.1 Absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
2.2 Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
3 Pin connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
4 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
5 Typical electrical performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
6 Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
6.1 Overvoltage protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
6.2 Feedback failure protection (FFP) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
6.3 Voltage feedforward . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
6.4 THD optimizer circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
6.5 Inductor saturation detection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
6.6 Power management/housekeeping functions . . . . . . . . . . . . . . . . . . . . . . 25
7 Application examples and ideas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
8 Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
9 Ordering codes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
10 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
L6564T List of tables
Doc ID 022671 Rev 1 3/33
List of tables
Table 1. Absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Table 2. Thermal data. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Table 3. Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Table 4. Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Table 5. Summary of L6564 idle states . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Table 6. SSO10 mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Table 7. Ordering information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Table 8. Document revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
List of figures L6564T
4/33 Doc ID 022671 Rev 1
List of figures
Figure 1. Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Figure 2. Pin connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Figure 3. IC consumption vs. VCC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Figure 4. IC consumption vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Figure 5. Vcc Zener voltage vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Figure 6. Startup and UVLO vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Figure 7. Feedback reference vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Figure 8. E/A output clamp levels vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Figure 9. UVLO saturation vs. TJ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Figure 10. OVP levels vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Figure 11. Inductor saturation threshold vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Figure 12. Vcs clamp vs. TJ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Figure 13. ZCD sink/source capability vs. TJ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Figure 14. ZCD clamp level vs. TJ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Figure 15. R discharge vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Figure 16. Line drop detection threshold vs. TJ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Figure 17. VMULTpk - VVFF dropout vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Figure 18. PFC_OK threshold vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Figure 19. PFC_OK FFD threshold vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Figure 20. Multiplier characteristics @ VFF = 1 V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Figure 21. Multiplier characteristics @ VFF = 3 V . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Figure 22. Multiplier gain vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Figure 23. Gate drive clamp vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Figure 24. Gate drive output saturation vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Figure 25. Delay to output vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Figure 26. Startup timer period vs. TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Figure 27. Output voltage setting, OVP and FFP functions: internal block diagram . . . . . . . . . . . . . . 18
Figure 28. Voltage feedforward: squarer/divider (1/V2) block diagram and transfer characteristic . . 20
Figure 29. RFF·CFF as a function of 3rd harmonic distortion introduced in the input current . . . . . . . 21
Figure 30. THD optimizer circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Figure 31. THD optimization: standard TM PFC controller (left side) and L6564T (right side) . . . . . . 23
Figure 32. Effect of boost inductor saturation on the MOSFET current and detection method . . . . . . 24
Figure 33. Interface circuits that let DC-DC converter's controller IC disable the L6564T. . . . . . . . . . 25
Figure 34. Demonstration board EVL6564-100W, wide-range mains: electrical schematic . . . . . . . . 27
Figure 35. L6564 100 W TM PFC: compliance with EN61000-3-2 standard . . . . . . . . . . . . . . . . . . . . 28
Figure 36. L6564 100 W TM PFC: compliance with JEITA-MITI standard . . . . . . . . . . . . . . . . . . . . . 28
Figure 37. L6564 100 W TM PFC: input current waveform @230-50 Hz - 100 W load . . . . . . . . . . . . 28
Figure 38. L6564 100W TM PFC: input current waveform @100 V-50 Hz - 100 W load . . . . . . . . . . 28
Figure 39. SSO10 package dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
L6564T Description
Doc ID 022671 Rev 1 5/33
1 Description
The L6564T is a current-mode PFC controller operating in transition mode (TM) and
represents the compact version of the L6563S as it embeds the same driver, reference and
control stages in a very compact 10-pin SO package.
The highly linear multiplier, along with a special correction circuit that reduces crossover
distortion of the mains current, allows wide-range-mains operation with an extremely low
THD even over a large load range.
The output voltage is controlled by means of a voltage-mode error amplifier and an accurate
(1% @TJ = 25 °C) internal voltage reference. The loop stability is optimized by the voltage
feedforward function (1/V2 correction), which in this IC uses a proprietary technique that
also considerably improves line transient response in case of both mains drops and surges
(“bidirectional”).
In addition to overvoltage protection able to control the output voltage during transient
conditions, the IC also provides protection against feedback loop failures or erroneous
settings. Other on-board protection functions allow brownout conditions and boost inductor
saturation to be safely handled.
The totem-pole output stage, capable of a 600 mA source and 800 mA sink current, is
suitable for a high power MOSFET or IGBT drive. This, combined with the other features
and the possibility to operate with ST's proprietary fixed-off-time control, makes the device
an excellent solution for SMPS up to 400 W that requires compliance with EN61000-3-2 and
JEITA-MITI standards.
Maximum ratings L6564T
6/33 Doc ID 022671 Rev 1
2 Maximum ratings
2.1 Absolute maximum ratings
2.2 Thermal data
Table 1. Absolute maximum ratings
Symbol Pin Parameter Value Unit
Vcc 10 IC supply voltage (Icc ≤ 20 mA) Self-limited V
--- 1, 3, 6 Max. pin voltage (Ipin ≤ 1 mA) Self-limited V
--- 2, 4, 5 Analog inputs and outputs -0.3 to 8 V
IZCD 7 Zero current detector max. current -10 (source)
10 (sink) mA
VFF pin 5 Maximum withstanding voltage range
test condition: CDF-AEC-Q100-002
“human body model”
Acceptance criteria: “normal performance”
+/- 1750 V
Other pins 1 to 4
6 to 10 +/- 2000 V
Table 2. Thermal data
Symbol Parameter Value Unit
RthJA Max. thermal resistance, junction-to-ambient 120 °C/W
Ptot Power dissipation @TA = 50 °C 0.75 W
TJJunction temperature operating range -40 to 150 °C
Tstg Storage temperature -55 to 150 °C
L6564T Pin connection
Doc ID 022671 Rev 1 7/33
3 Pin connection
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Table 3. Pin description
n° Name Function
1INV
Inverting input of the error amplifier. The information on the output voltage of the PFC pre-
regulator is fed into the pin through a resistor divider.
The pin normally features high impedance.
2COMP
Output of the error amplifier. A compensation network is placed between this pin and INV (pin
1) to achieve stability of the voltage control loop and ensure high power factor and low THD.
To avoid uncontrolled rise of the output voltage at zero load, when the voltage on the pin falls
below 2.4 V the gate driver output is inhibited (burst-mode operation).
3MULT
Main input to the multiplier. This pin is connected to the rectified mains voltage via a resistor
divider and provides the sinusoidal reference to the current loop. The voltage on this pin is used
also to derive the information on the RMS mains voltage.
4CS
Input to the PWM comparator. The current flowing in the MOSFET is sensed through a resistor,
the resulting voltage is applied to this pin and compared with an internal reference to determine
MOSFET’s turn-off.
A second comparison level at 1.7 V detects abnormal currents (e.g. due to boost inductor
saturation) and, on this occurrence, activates a safety procedure that temporarily stops the
converter and limits the stress of the power components.
5VFF
Second input to the multiplier for 1/V2 function. A capacitor and a parallel resistor must be
connected from the pin to GND. They complete the internal peak-holding circuit that derives the
information on the RMS mains voltage. The voltage at this pin, a DC level equal to the peak
voltage on pin MULT (3), compensates the control loop gain dependence on the mains voltage.
Never connect the pin directly to GND but with a resistor ranging from 100 K ohm (minimum) to
2 M ohm (maximum). This pin is internally connected to a comparator in order to provide the
brownout (AC mains undervoltage) protection. A voltage below 0.8 V shuts down (not latched)
the IC and brings its consumption to a considerably lower level. The IC restarts as the voltage
at the pin goes above 0.88 V.
Pin connection L6564T
8/33 Doc ID 022671 Rev 1
6PFC_OK
PFC pre-regulator output voltage monitoring/disable function. This pin senses the output
voltage of the PFC pre-regulator through a resistor divider and is used for protection purposes.
If the voltage on the pin exceeds 2.5 V the IC stops switching and restarts as the voltage on the
pin falls below 2.4 V. However, if at the same time the voltage of the INV pin falls below 1.66 V,
a feedback failure is assumed. In this case the device is latched off. Normal operation can be
resumed only by cycling Vcc. bringing its value lower than 6 V before moving up to the turn-on
threshold.
If the voltage on this pin is brought below 0.23 V the IC is shut down. To restart the IC the
voltage on the pin must go above 0.27 V. This can be used as a remote on/off control input.
7ZCDBoost inductor’s demagnetization sensing input for transition-mode operation. A negative-going
edge triggers MOSFET’s turn-on.
8 GND Ground. Current return for both the signal part of the IC and the gate driver.
9GD
Gate driver output. The totem pole output stage is able to drive power MOSFET’s and IGBT’s
with a peak current of 600 mA source and 800 mA sink. The high-level voltage of this pin is
clamped at about 12 V to avoid excessive gate voltages.
10 Vcc
Supply voltage of both the signal part of the IC and the gate driver. Sometimes a small bypass
capacitor (0.1 µF typ.) to GND might be useful to get a clean bias voltage for the signal part of
the IC.
Table 3. Pin description (continued)
n° Name Function
L6564T Electrical characteristics
Doc ID 022671 Rev 1 9/33
4 Electrical characteristics
TJ = -40 to 125 °C, VCC = 12 V, CO = 1 nF between pin GD and GND, CFF = 1 µF and
RFF = 1 MΩ between pin VFF and GND; unless otherwise specified.
Table 4. Electrical characteristics
Symbol Parameter Test condition Min. Typ. Max. Unit
Supply voltage
Vcc Operating range After turn-on 10.3 22.5 V
VccOn Turn-on threshold (1) 11 12 13.2 V
VccOff Turn-off threshold (1) 8.7 9.5 10.5 V
Vccrestart Vcc for resuming from latch OVP latched 567V
Hys Hysteresis 2.3 2.7 V
VZZener voltage Icc = 20 mA 22.5 25 28 V
Supply current
Istart-up Startup current Before turn-on, Vcc=10 V 90 180 µA
IqQuiescent current After turn-on, VMULT = 1 V 4 5.5 mA
ICC Operating supply current @ 70 kHz 5 6.0 mA
Iqdis Idle state quiescent current
VPFC_OK> VPFC_OK_S AND VINV <
VFFD
180 320 µA
VPFC_OK<VPFC_OK_D 1.5 2.5 µA
IqQuiescent current VPFC_OK>VPFC_OK_S OR
VCOMP<2.3V 2.2 3.2 µA
Multiplier input
IMULT Input bias current VMULT = 0 to 3 V -0.2 -1 µA
VMULT Linear operation range 0 to 3 V
VCLAMP Internal clamp level IMULT = 1 mA 9 9.5 V
∆Vcs
∆VMULT
Output max. slope VMULT = 0 to 0.4 V, VVFF = 1 V
VCOMP = upper clamp 1.33 1.66 V/V
KMGain (2) VMULT = 1 V, VCOMP= 4 V 0.375 0.45 0.525 V
Error amplifier
VINV Voltage feedback input threshold TJ = 25 °C 2.475 2.5 2.525 V
10.3 V < Vcc < 22.5 V (3) 2.45 2.55
Line regulation Vcc = 10.3 V to 22.5 V 2 5 mV
IINV Input bias current VINV = 0 to 4 V -0.2 -1 µA
VINVCLAMP Internal clamp level IINV = 1 mA 8 9 V
Gv Voltage gain Open loop 60 80 dB
Electrical characteristics L6564T
10/33 Doc ID 022671 Rev 1
GB Gain-bandwidth product 1 MHz
ICOMP
Source current VCOMP = 4 V, VINV = 2.4 V 1.5 4 mA
Sink current VCOMP = 4 V, VINV = 2.6 V 2 4.5 mA
VCOMP
Upper clamp voltage ISOURCE = 0.5 mA 5.7 6.2 6.7
VBurst-mode voltage (3) 2.3 2.4 2.5
Lower clamp voltage ISINK = 0.5 mA (3) 2.1 2.25 2.4
Boost inductor saturation detector
VCS_th Threshold on current sense (3) 1.6 1.7 1.8 V
IINV E/A input pull-up current After VCS > VCS_th, before restarting 5 10 13 µA
Startup timer
tSTART_DEL Startup delay First cycle after wake-up 20 50 100 µs
tSTART Timer period 75 150 350 µs
Restart after VCS > VCS_th 150 300 700
Current sense comparator
ICS Input bias current VCS = 0 1 µA
tLEB Leading edge blanking 70 150 300 ns
td(H-L) Delay to output 70 200 350 ns
VCSclamp Current sense reference clamp VCOMP = upper clamp,
VMULT =1 V, VVFF = 1 V 0.97 1.08 1.2 V
Vcsofst Current sense offset VMULT = 0 V, VVFF = 3 V 40 70 mV
VMULT = 3 V, VVFF = 3 V 20
PFC_OK functions
IPFC_OK Input bias current VPFC_OK = 0 to 2.6 V -0.1 -1 µA
VPFC_OK_C Clamp voltage IPFC_OK = 1 mA 8.5 9.5 V
VPFC_OK_S OVP threshold (1) Voltage rising 2.435 2.5 2.565 V
VPFC_OK_R Restart threshold after OVP (1) Voltage falling 2.34 2.4 2.46 V
VPFC_OK_D Disable threshold (1) Voltage falling 0.08 0.40 V
VPFC_OK_D Disable threshold (1) Voltage falling Tj = 25 °C 0.17 0.23 0.29 V
VPFC_OK_E Enable threshold (1) Voltage rising 0.10 0.43 V
VPFC_OK_E Enable threshold (1) Voltage rising Tj = 25 °C 0.21 0.27 0.32 V
VFFD
Feedback failure detection
threshold (VINV falling) VPFC_OK = VPFC_OK_S 1.61 1.66 1.71 V
Table 4. Electrical characteristics (continued)
Symbol Parameter Test condition Min. Typ. Max. Unit
L6564T Electrical characteristics
Doc ID 022671 Rev 1 11/33
Voltage feedforward
VVFF Linear operation range 1 3 V
∆V Dropout VMULTpk-VVFF
Vcc < VccOn 850 mV
Vcc > or = to VccOn 20
∆VVFF Line drop detection thresh. Below peak value 30 70 110 mV
∆VVFF Line drop detection thresh. Below peak value Tj =25°C 50 70 90 mV
RDISCH Internal discharge resistor Tj = 25 °C 7.5 10 12.5 kΩ
520
VDIS Disable threshold (2) Voltage falling 0.725 0.8 0.875 V
VEN Enable threshold (2) Voltage rising 0.845 0.88 0.915 V
Zero current detector
VZCDH Upper clamp voltage IZCD = 2.5 mA 5.0 5.7 V
VZCDL Lower clamp voltage IZCD = - 2.5 mA -0.3 0 0.3 V
VZCDA
Arming voltage
(positive-going edge) 1.1 1.4 1.9 V
VZCDT
Triggering voltage
(negative-going edge) 0.5 0.7 1 V
IZCDb Input bias current VZCD = 1 to 4.5 V 1 µA
IZCDsrc Source current capability -2.0 -4 mA
IZCDsnk Sink current capability 2.0 5 mA
Gate driver
VOL Output low voltage Isink = 100 mA 0.6 1.4 V
VOH Output high voltage Isource = 5 mA 9.5 10.3 V
Isrcpk Peak source current -0.6 A
Isnkpk Peak sink current 0.8 A
tfVoltage fall time 30 60 ns
trVoltage rise time 45 150 ns
VOclamp Output clamp voltage Isource = 5 mA; Vcc = 20 V 10 12 15 V
UVLO saturation Vcc= 0 to VCCon, Isink= 2 mA 1.2 V
1. Parameters tracking each other.
2. The multiplier output is given by:
3. Parameters tracking each other.
Table 4. Electrical characteristics (continued)
Symbol Parameter Test condition Min. Typ. Max. Unit
()
2
VFF
V
5.2
COMP
V
MULT
V
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cs
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−⋅
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Typical electrical performance L6564T
12/33 Doc ID 022671 Rev 1
5 Typical electrical performance
Figure 3. IC consumption vs. VCC Figure 4. IC consumption vs. TJ
0.001
0.01
0.1
1
10
10 0
0 5 10 1 5 2 0 25 3 0
Vc c [V ]
Icc [mA]
Vcc OFF
Vcc ON
Co=1nF
f =70kHz
Tj = 25°C
0.01
0.1
1
10
- 5 0 -2 5 0 2 5 5 0 75 1 00 1 2 5 1 50 1 7 5
Tj (C)
Ic current (mA)
Op e r a ti n g
Qu ie sc e n t
Dis able d or
du r i n g OV P
Latched off
Befo r e Sta r t up
VCC=1 2V
Co = 1nF
f = 7 0 kH z
Figure 5. Vcc Zener voltage vs. TJFigure 6. Startup and UVLO vs. TJ
22
23
24
25
26
27
28
-50-25 0 25 50 75100125150175
Tj (C)
V
6
7
8
9
10
11
12
13
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
V
VCC
-
ON
VC C -OFF
L6564T Typical electrical performance
Doc ID 022671 Rev 1 13/33
Figure 7. Feedback reference vs. TJFigure 8. E/A output clamp levels vs. TJ
2. 4
2.45
2. 5
2.55
2. 6
-50-25 0 25 5075100125150175
Tj (C)
pi n INV (V )
VC C = 1 2V
0
1
2
3
4
5
6
7
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
VCOMP (V)
U
per
C
lam
p
L
owe r Clamp
V
CC = 12V
Figure 9. UVLO saturation vs. TJFigure 10. OVP levels vs. TJ
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1
-50-25 0 2550 75100125150175
Tj (C)
V
VCC = 0 V
2.36
2.38
2.4
2.42
2.44
2.46
2.48
2.5
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
PFC_OK levels (V )
OVP T h
Resta rt Th
Typical electrical performance L6564T
14/33 Doc ID 022671 Rev 1
Figure 11. Inductor saturation threshold vs. TJFigure 12. Vcs clamp vs. TJ
1.1
1.2
1.3
1.4
1.5
1.6
1.7
1.8
1.9
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
CS pi n (V )
1
1.1
1.2
1.3
1.4
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
VC Sx (V )
VCC = 12V
VCOMP =Upper clamp
Figure 13. ZCD sink/source capability vs. TJFigure 14. ZCD clamp level vs. TJ
-8
-6
-4
-2
0
2
4
6
8
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
IZC Dsrc ( mA
)
VCC = 12V
Source current
Sink curren t
-1
0
1
2
3
4
5
6
7
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
VZCD pin (V)
VC C = 1 2V
Iz c d =
±2.5mV
Upper Clamp
Lower Cl amp
L6564T Typical electrical performance
Doc ID 022671 Rev 1 15/33
Figure 15. R discharge vs. TJFigure 16. Line drop detection threshold vs. TJ
0
2
4
6
8
10
12
14
16
18
20
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
kOhm
0
10
20
30
40
50
60
70
80
90
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
mV
Figure 17. VMULTpk - VVFF dropout vs. TJFigure 18. PFC_OK threshold vs. TJ
-2
-1.5
-1
-0.5
0
0. 5
1
1. 5
2
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
⎯ (m V)
0
0.05
0.1
0.15
0.2
0.25
0.3
0.35
0.4
-50-25 0255075100125150175
Tj (C)
Th (V
)
ON
OFF
Figure 19. PFC_OK FFD threshold vs. TJ
1. 4
1. 5
1. 6
1. 7
1. 8
1. 9
2
-50-25 0 255075100125150175
Tj (C )
VFFD Th (V )
Typical electrical performance L6564T
16/33 Doc ID 022671 Rev 1
Figure 20. Multiplier characteristics
@ VFF = 1 V
Figure 21. Multiplier characteristics
@ VFF = 3 V
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1
1.1
1.2
0 0.10.20.30.40.50.6 0.70.80.9 1 1.1
VM UL T ( V )
VCS (V)
4.5 V
3.5 V
Up p e r vo lt a g e cl am p
3.0
2.6 V
5.0V
4. 0V
5.5
VCOMP
0
100
200
300
400
500
600
700
0 0.5 1 1.5 2 2.5 3 3.5
VMULT (V)
VCS (mV)
2.6V
3.0V
4.5V
4.0V
3.5V
5.0V
5.5V
Upper vo ltage
l
VCOMP
Figure 22. Multiplier gain vs. TJFigure 23. Gate drive clamp vs. TJ
0. 2
0. 3
0. 4
0. 5
-50-25 0 25 5075100125150175
Tj (C)
Gain (1/V)
VC C = 1 2V
VC OMP = 4 V
VMU LT = VFF = 1V
12.65
12.7
12.75
12.8
12.85
12.9
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
V
VCC = 20V
L6564T Typical electrical performance
Doc ID 022671 Rev 1 17/33
Figure 24. Gate drive output saturation vs. TJFigure 25. Delay to output vs. TJ
0
2
4
6
8
10
12
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
V
Low level
Hig h le ve l
50
100
150
200
250
300
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
TD (H - L ) (n s)
VCC = 12V
Figure 26. Startup timer period vs. TJ
0
50
100
150
200
250
300
350
400
450
-50 -25 0 25 50 75 100 125 150 175
Tj (C)
Ti m e (us)
Fi r s t Ci c l e
Timer
Aft e r OC P
Application information L6564T
18/33 Doc ID 022671 Rev 1
6 Application information
6.1 Overvoltage protection
Normally, the voltage control loop keeps the output voltage Vo of the PFC pre-regulator
close to its nominal value, set by the ratio of the resistors R1 and R2 of the output divider. A
pin of the device (PFC_OK) has been dedicated to monitor the output voltage with a
separate resistor divider (R3 high, R4 low, see
Figure 27
). This divider is selected so that
the voltage at the pin reaches 2.5 V if the output voltage exceeds a preset value, usually
larger than the maximum Vo that can be expected.
Example: VO = 400 V, VOX = 434 V. Select: R3 = 8.8 MΩ; then: R4 = 8.8 MΩ ·2.5/(434-2.5) =
51 kΩ.
When this function is triggered, the gate drive activity is immediately stopped until the
voltage on the pin PFC_OK drops below 2.4 V. Note that R1, R2, R3 and R4 can be selected
without any constraints. The unique criterion is that both dividers have to sink a current from
the output bus which needs to be significantly higher than the bias current of both INV and
PFC_OK pins.
Figure 27. Output voltage setting, OVP and FFP functions: internal block diagram
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L6564T Application information
Doc ID 022671 Rev 1 19/33
6.2 Feedback failure protection (FFP)
The OVP function described above handles “normal” overvoltage conditions, i.e. those
resulting from an abrupt load/line change or occurring at startup. In case the overvoltage is
generated by a feedback disconnection, for instance when the upper resistor of the output
divider (R1) fails to open, the comparator detects the voltage at pin INV. If the voltage is
lower than 1.66 V and the OVP is active, the FFP is triggered, the gate drive activity is
immediately stopped, the device is shut down, its quiescent consumption is reduced below
180 µA and the condition is latched as long as the supply voltage of the IC is above the
UVLO threshold. To restart the system it is necessary to recycle the input power, so that the
Vcc voltage of the L6564T goes below 6 V.
The pin PFC_OK doubles its function as a not-latched IC ‘Disable’: a voltage below 0.23 V
shuts down the IC, reducing its consumption below 2 mA. To restart the IC simply let the
voltage at the pin go above 0.27 V.
Note that these functions offer complete protection against not only feedback loop failures or
erroneous settings, but also against a failure of the protection itself. Either resistor of the
PFC_OK divider failing short or open or a PFC_OK pin floating results in shutting down the
IC and stopping the pre-regulator.
6.3 Voltage feedforward
The power stage gain of PFC pre-regulators varies with the square of the RMS input
voltage. So does the crossover frequency FC of the overall open-loop gain because the gain
has a single pole characteristic. This leads to a large trade-off in the design.
For example, setting the gain of the error amplifier to get FC = 20 Hz @ 264 Vac means
having FC 4 Hz @ 88 Vac, resulting in a sluggish control dynamics. Additionally, the slow
control loop causes large transient current flow during rapid line or load changes that are
limited by the dynamics of the multiplier output. This limit is considered when selecting the
sense resistor to let the full load power pass under minimum line voltage conditions, with
some margin. But a fixed current limit allows excessive power input at high line, whereas a
fixed power limit requires the current limit to vary inversely with the line voltage.
Voltage feedforward can compensate for the gain variation with the line voltage and allow
the minimizing of all the above-mentioned issues. It consists in deriving a voltage
proportional to the input RMS voltage, feeding this voltage into a squarer/divider circuit (1/V2
corrector) and providing the resulting signal to the multiplier that generates the current
reference for the inner current control loop (see
Figure 28
).
Application information L6564T
20/33 Doc ID 022671 Rev 1
Figure 28. Voltage feedforward: squarer/divider (1/V2) block diagram and transfer characteristic
In this way a change of the line voltage causes an inversely proportional change of the half
sine amplitude at the output of the multiplier (if the line voltage doubles the amplitude of the
multiplier output is halved and vice versa) so that the current reference is adapted to the new
operating conditions with (ideally) no need for invoking the slow dynamics of the error
amplifier. Additionally, the loop gain is constant throughout the input voltage range, which
significantly improves dynamic behavior at low line and simplifies loop design.
In fact, deriving a voltage proportional to the RMS line voltage implies a form of integration,
which has its own time constant. If it is too small the voltage generated is affected by a
considerable amount of ripple at twice the mains frequency that causes distortion of the
current reference (resulting in high THD and poor PF); if it is too large there is a
considerable delay in setting the right amount of feedforward, resulting in excessive
overshoot and undershoot of the pre-regulator's output voltage in response to large line
voltage changes. Clearly a trade-off was required.
The L6564T realizes a NEW voltage feed forward that, with a technique that makes use of
just two external parts, strongly minimizes this time constant trade-off issue whichever
voltage change occurs on the mains, both surges and drops. A capacitor CFF and a resistor
RFF, both connected from pin VFF (#5) to ground, complete an internal peak-holding circuit
that provides a DC voltage equal to the peak of the rectified sine wave applied on pin MULT
(#3). In this way, in case of sudden line voltage rise, CFF is rapidly charged through the low
impedance of the internal diode; in case of line voltage drop, an internal “mains drop”
detector enables a low impedance switch which suddenly discharges CFF avoiding long
settling time before reaching the new voltage level. The discharge of CFF is stopped as its
voltage equals the voltage on the MULT pin or if the voltage on the VFF pin falls below 0.88
V, to prevent the “Brownout protection” function from being improperly activated (see
Section 6.3
).
As a result of the VFF pin functionality, an acceptably low steady-state ripple and low current
distortion can be achieved with a limited undershoot or overshoot on the pre-regulator's
output.
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L6564T Application information
Doc ID 022671 Rev 1 21/33
The twice-mains-frequency (2•fL) ripple appearing across CFF is triangular with a peak-to-
peak amplitude that, with good approximation, is given by:
where fL is the line frequency. The amount of 3rd harmonic distortion introduced by this
ripple, related to the amplitude of its 2•fL component, is:
Figure 29
shows a diagram that helps choose the time constant RFF·CFF based on the
amount of maximum desired 3rd harmonic distortion. Note that there is a minimum value for
the time constant RFF · CFF below which improper activation of the VFF fast discharge may
occur. In fact, the twice-mains-frequency ripple across CFF under steady-state conditions
must be lower than the minimum line drop detection threshold (∆VFF_min = 40 mV).
Therefore:
Always connect RFF and CFF to the pin, the IC does not work properly if the pin is either left
floating or connected directly to ground.
Figure 29. RFF·CFF as a function of 3rd harmonic distortion introduced in the input
current
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Application information L6564T
22/33 Doc ID 022671 Rev 1
6.4 THD optimizer circuit
The L6564T is provided with a special circuit that reduces the conduction dead-angle
occurring to the AC input current near the zero-crossings of the line voltage (crossover
distortion). In this way the THD (total harmonic distortion) of the current is considerably
reduced.
A major cause of this distortion is the inability of the system to transfer energy effectively
when the instantaneous line voltage is very low. This effect is magnified by the high-
frequency filter capacitor placed after the bridge rectifier, which retains some residual
voltage that causes the diodes of the bridge rectifier to be reverse-biased and the input
current flow to temporarily stop.
To overcome this issue the device forces the PFC pre-regulator to process more energy
near the line voltage zero-crossings as compared to that commanded by the control loop.
This results in both minimizing the time interval where energy transfer is lacking and fully
discharging the high-frequency filter capacitor after the bridge.
Figure 30
shows the internal block diagram of the THD optimizer circuit.
Figure 30. THD optimizer circuit
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L6564T Application information
Doc ID 022671 Rev 1 23/33
Figure 31. THD optimization: standard TM PFC controller (left side) and L6564T (right side)
Essentially, the circuit artificially increases the ON-time of the power switch with a positive
offset added to the output of the multiplier in the proximity of the line voltage zero-crossings.
This offset is reduced as the instantaneous line voltage increases, so that it becomes
negligible as the line voltage moves toward the top of the sinusoid. Furthermore, the offset is
modulated by the voltage on the VFF pin (see
Section 6.3
) so as to have little offset at low
line, where energy transfer at zero crossings is typically quite good, and a larger offset at
high line where the energy transfer gets worse.
The effect of the circuit is shown in
Figure 31
, where the key waveforms of a standard TM
PFC controller are compared to those of this chip.
To take maximum benefit from the THD optimizer circuit, the high-frequency filter capacitor
after the bridge rectifier should be minimized, compatibly with EMI filtering needs. A large
capacitance, in fact, introduces a conduction dead-angle of the AC input current in itself -
even with an ideal energy transfer by the PFC pre-regulator - therefore reducing the
effectiveness of the optimizer circuit.
6.5 Inductor saturation detection
The boost inductor's hard saturation may be a fatal event for a PFC pre-regulator: the
current up-slope becomes so large (50-100 times steeper, see
Figure 32
) that during the
current sense propagation delay the current may reach abnormally high values. The voltage
drop caused by this abnormal current on the sense resistor reduces the gate-to-source
voltage, so that the MOSFET may work in the active region and dissipate a huge amount of
power, which leads to a catastrophic failure after few switching cycles.
Imains
Vdrain
Imains
Vdrain
Input current In pu t cu rren t
MOSFET's drain voltage M OSFE T's dra i n vol t ag e
Rec tified mai ns vol tage Rec tified mains voltage
Input current In pu t cu rren t
Application information L6564T
24/33 Doc ID 022671 Rev 1
However, in some applications such as AC-DC adapters, where the PFC pre-regulator is
turned off at light load for energy saving reasons, even a well-designed boost inductor may
occasionally slightly saturate when the PFC stage is restarted because of a larger load
demand. This happens when the restart occurs at an unfavorable line voltage phase, i.e.
when the output voltage is significantly below the rectified peak voltage. As a result, in the
boost inductor the inrush current coming from the bridge rectifier adds up to the switched
current and, furthermore, there is little or no voltage available for demagnetization.
To cope with a saturated inductor, the L6564T is provided with a second comparator on the
current sense pin (CS, pin 4) that stops the IC if the voltage, normally limited within 1.1 V,
exceeds 1.7 V. After that, the IC is attempted to restart by the internal starter circuitry; the
starter repetition time is twice the nominal value to guarantee lower stress for the inductor
and boost diode. Hence, the system safety is considerably increased.
Figure 32. Effect of boost inductor saturation on the MOSFET current and detection method
L6564T Application information
Doc ID 022671 Rev 1 25/33
6.6 Power management/housekeeping functions
A communication line with the control IC of the cascaded DC-DC converter can be
established via the disable function included in the PFC_OK pin (see
Section 6.2
for more
details). This line is typically used to allow the PWM controller of the cascaded DC-DC
converter to shut down the L6564T in case of light load and to minimize the no-load input
consumption. Should the residual consumption of the chip be an issue, it is also possible to
cut down the supply voltage. The interface circuits are shown in
Figure 32
. Needless to say,
this operation assumes that the cascaded DC-DC converter stage works as the master and
the PFC stage as the slave or, in other words, that the DC-DC stage starts first, it powers
both controllers and enables/disables the operation of the PFC stage.
Figure 33. Interface circuits that let DC-DC converter's controller IC disable the
L6564T
Another function available is the brownout protection which is basically a not-latched
shutdown function that is activated when a condition of mains undervoltage is detected. This
condition may cause overheating of the primary power section due to an excess of RMS
current. Brownout can also cause the PFC pre-regulator to function in open loop and this
may be dangerous to the PFC stage itself and the downstream converter, should the input
voltage return abruptly to its rated value. Another problem is the spurious restarts that may
occur during converter power-down and that cause the output voltage of the converter not to
decay to zero monotonically. For these reasons it is usually preferable to shut down the unit
in case of brownout. The brownout threshold is internally fixed at 0.8 V and is sensed on pin
VFF (5) during the voltage falling and an 80 mV threshold hysteresis prevents rebounding at
input voltage turn-off. In
Tab l e 5
it is possible to find a summary of all of the above
mentioned working conditions that cause the device to stop operating.
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26/33 Doc ID 022671 Rev 1
Table 5. Summary of L6564 idle states
Condition Caused or revealed by IC behavior Restart condition Typical IC
consumption
UVLO Vcc < VccOff Disabled Vcc > VccOn 90 µA
Feedback
disconnected
PFC_OK > VPFC_OK_S
and
INV < 1.66 V
Latched Vcc < Vccrestart then
Vcc > VccOn
180 µA
Standby PFC_OK < VPFC_OK_D Stop switching PFC_OK >
VPFC_OK_E
1.5 mA
AC brownout VFF < VDIS Stop switching RUN > VEN 1.5 mA
OVP PFC_OK > VPFC_OK_S Stop Switching PFC_OK <
VPFC_OK_R
2.2 mA
Low
consumption COMP < 2.4 V Burst mode COMP > 2.4V 2.2 mA
Saturated
boost inductor Vcs > VCS_th Doubled Tstart Auto restart 2.2 mA
L6564T Application examples and ideas
Doc ID 022671 Rev 1 27/33
7 Application examples and ideas
Figure 34. Demonstration board EVL6564-100W, wide-range mains: electrical schematic
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Application examples and ideas L6564T
28/33 Doc ID 022671 Rev 1
Figure 35. L6564 100 W TM PFC: compliance
with EN61000-3-2 standard
Figure 36. L6564 100 W TM PFC: compliance
with JEITA-MITI standard
0.0001
0.001
0.01
0.1
1
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Harmonic Order [n]
Harmonic Current [A]
Meas ured value EN61000-3-2 class-D limits
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1
10
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Harmonic Order [n]
Harmonic Current [ A]
Measur ed value JEITA-MITI Class- D lim its
Figure 37. L6564 100 W TM PFC: input current
waveform @230-50 Hz - 100 W load
Figure 38. L6564 100W TM PFC: input current
waveform @100 V-50 Hz - 100 W
load
L6564T Package mechanical data
Doc ID 022671 Rev 1 29/33
8 Package mechanical data
In order to meet environmental requirements, ST offers these devices in different grades of
ECOPACK® packages, depending on their level of environmental compliance. ECOPACK
specifications, grade definitions and product status are available at: www.st.com.
ECOPACK is an ST trademark.
Table 6. SSO10 mechanical data
Dim.
Databook (mm.)
Min. Typ. Max.
A 1.75
A1 0.10 0.25
A2 1.25
b 0.31 0.51
c 0.17 0.25
D 4.80 4.90 5
E 5.80 6 6.20
E1 3.80 3.90 4
e 1
h 0.25 0.50
L 0.40 0.90
K 0° 8°
Package mechanical data L6564T
30/33 Doc ID 022671 Rev 1
Figure 39. SSO10 package dimensions
8140761 rev. A
L6564T Ordering codes
Doc ID 022671 Rev 1 31/33
9 Ordering codes
Table 7. Ordering information
Order codes Package Packing
L6564TD SSO10 Tube
L6564TDTR Tape and reel
Revision history L6564T
32/33 Doc ID 022671 Rev 1
10 Revision history
Table 8. Document revision history
Date Revision Changes
18-Jan-2012 1 Initial release
L6564T
Doc ID 022671 Rev 1 33/33
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