LM4962
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LM4962 Ceramic Speaker Driver
Check for Samples: LM4962
1FEATURES DESCRIPTION
The LM4962 is an audio power amplifier primarily
2• Click and Pop Circuitry Eliminates Noise designed for driving Ceramic Speaker for applications
During Turn-On and Turn-Off Transitions in Cell Phones, Smart Phones, PDA's and other
• Low Current Shutdown Mode portable applications. It is capable of driving 15Vpp
• Low Quiescent Current (typ) BTL with less than 1% THD+N from a 3.2VDC
power supply. The LM4962 features and low power
• Mono 15Vp-p BTL Output, RL= 2μF+9.4Ω,consumption shutdown mode, an internal thermal
f = 1kHz, 1% THD+N shutdown protection mechanism, along with over
• Over-Current Protection current protection (OCP) and over voltage protection
• Over-Voltage Protection (OVP).
• Unity-Gain Stable Boomer audio power amplifiers were designed
specifically to provide high quality output power with a
• External Gain Configuration Capability minimal number of external components. The
• Including Band Switch Function LM4962 does not require bootstrap capacitors, or
• Leakage Cut Switch (SW-LEAK) snubber circuits.
• Soft-Start Function The LM4962 also features a Band-Switch function
• Space-Saving DSBGA Package (2mm x 2.5mm) which allows the user to use one amplifier device for
both receiver (earpiece) mode and ringer/loudspeaker
APPLICATIONS mode.
• Smart Phones The LM4962 contains advanced click and pop
circuitry that eliminates noises which would otherwise
• Mobile Phones and Multimedia Terminals occur during turn-on and turn-off transitions.
• PDA's, Internet Appliances, and Portable Additionally, the internal boost converter features a
Gaming soft-start function.
• Portable DVD The LM4962 is unity-gain stable and can be
• Digital Still Cameras/Camcorders configured by external gain-setting resistors.
KEY SPECIFICATIONS
• Quiescent Power Supply Current
(Boost Converter + Amplifier): 9 mA (typ)
• Voltage Swing in BTL at 1% THD,
f = 1kHz: 15 Vp-p (typ)
• Shutdown Current: 0.1 μA (typ)
• OVP: 8.5V < VAMP < 9.5 V
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2005–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
D
C
B
A
CCHG
GND BW2 FBBW1
E
SD-
Amp SD-
Boost GND
(SW) SW
1 4
32
Soft-
Start Flag-
out VDD Boot-
Strap
VIN OC/OV
Detect
VAMP
VO1SW-
Leak
VO2
Bypass
LM4962
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Connection Diagram
Top View
Figure 1. DSBGA Package
See Package Number YZR002011A
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Vdd
SD Amp
Bypass
Cchg
Vin
BW1 BW2
Vo1
Vo2
GND
Vamp
SW-LEAK
FB
SW
GND (sw)
Soft-Start
Flagout
SD Boost
Flagout
Vdd
Cs1
4.7 PF
Css
10 nF
Ro2
4.7
Ro1
4.7
Ceramic Speaker
2.0 PF
Cs2
4.7 PF
C2
4.7 PF
R2
25k
C3
100p
L1
10 PH
R5
4.9k
Rf1
200k
RinA
20k
CinA
0.1 PF
Cf1
82 pF
Rf2
20k
V1 = 1.2(1+R2/R5)
OC/OV Detect
Rchg1
1k
Rs
100m
Bootstrap
To LM4951
for stereo
solution
R3
1.6k
Shutdown 2
Shutdown 1
D2
Load
Cb1
1.0 PF
LM4962
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Typical Application
Figure 2. Typical Audio Amplifier Application Circuit
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+
-
PWM
VBG
VDD
Driver
+
-
+
-
Vamp
BIAS, SHUTDOWN,
and
PROTECTION
CIRCUITRY
Shutdown
control
VIH
VIL
flagout
L1 D2
C2
R2
R3
R5
C3
CinA RinA Rf1
Cf1
Rf2
Cf2
Cs2
Ro1
Ro2
Cb1
SW
SW-LEAK
FB
GND(SW)
GND
BW1
BW2
VIN
Cchg
SD Amp
SD Boost
GND
Vo1
Vo2
Rs
LM4962
OC/OV Detect
Cs1
VDD
Css
Rchg1
Bypass
Soft-Start
Flagout
Shutdown
control
VIH
VIL
LM4962
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Block Diagram
Figure 3. LM4962 Block Diagram
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
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Absolute Maximum Ratings(1) (2)(3)
Supply Voltage (VDD) 9.5V
Amplifier Supply Voltage (VAMP) 9.5V
Storage Temperature −65°C to +150°C
Input Voltage −0.3V to VDD + 0.3V
Power Dissipation(4) Internally limited
ESD Susceptibility(5) 2000V
ESD Susceptibility(6) 200V
Junction Temperature 150°C
Thermal Resistance θJA (DSBGA)(7) 73°C/W
(1) All voltages are measured with respect to the GND pin, unless otherwise specified.
(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication
of device performance.
(3) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
(4) The maximum power dissipation must be derated at elevated temperatures and is dictated by TJMAX,θJA, and the ambient temperature,
TA. The maximum allowable power dissipation is PDMAX = (TJMAX −TA) / θJA or the given in Absolute Maximum Ratings, whichever is
lower.
(5) Human body model, 100pF discharged through a 1.5kΩresistor.
(6) Machine Model, 220pF–240pF discharged through all pins.
(7) The value for a θJA is measured with the LM4962 mounted on a 3” x 1.5” 4 layer board. The copper thickness for all 4 layers is 0.5oz
(roughly 0.18mm).
Operating Ratings
Temperature Range (TMIN ≤TA≤TMAX)(1) −40°C ≤TA≤+85°C
Supply Voltage (VDD) 3.0V < VDD < 5.0V
Amplifier Supply Voltage (V1)(2) 2.7V < VAMP < 9.0V
(1) Temperature range is tentative, pending characterization.
(2) An amplifier supply voltage of 9.0V can only be obtained when the over current and over voltage protection circuitry is disabled (OV/OC
Detect pin is disabled).
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Electrical Characteristics
The following specifications apply for VDD = 3.2V, AV-BTL = 26dB, ZL= 2µF+9.4Ω, Cb = 1.0μF, R2= 25KΩ, R5= 4.9KΩunless
otherwise specified. Limits apply for TA= 25°C. LM4962 Units
Parameter Test Conditions (Limits)
Typ(1) Limit(2)(3)
IDD Quiescent Power Supply Current VIN = 0V, 9 12 mA (max)
in Boosted Ringer Mode
Iddrcv Quiescent Power Supply Current SD Boost = GND 3 5 mA (max)
in Receiver Mode SD Amp = VDD
ISD Shutdown Current(4) SD Boost = SD Amp = GND 0.1 2.0 µA (max)
VLH Logic High Threshold Voltage For SD Boost, SD Amp 1.2 V (min)
VLL Logic Low Threshold Voltage For SD Boost, SD Amp 0.4 V (max)
RPULLDOWN Pulldown Resistor For SD Amp, SD Boost 80 60 kΩ(min)
TWUBC Boost Converter Wake-up Time CSS = 10nF 2 5 ms (max)
TWUA Audio Amplifier Wake-up Time (For Vdd = 2.7V to 8.5V) 20 40 msec
VOUT Output Voltage Swing THD = 1% (max), f = 1kHz 15 14 Vpp (min)
THD+N Total Harmonic Distortion + Noise Vout = 14Vpp, f = 1kHz 0.4 1.0 %
εOS Output Noise A-Weighted Filter, VIN = 0V 125 µV
PSRR Power Supply Rejection Ratio VRIPPLE = 200mVp-p, f = 100Hz, 86 71 dB (min)
Input Referred
Ron-sw-leak On Resistance on SW-Leak SD Boost = GND 30 50 Ω(max)
Isink = 100μA
Ron Flagout On resistance Isink = 1mA 50 100 Ω(max)
Vovp Sensitivity of Over Voltage Protection Flagout = GND 9.0 9.5 V (max)
on VAMP 8.5 V (min)
Vocp Sensitivity of Over Current Protection Flagout = GND 185 275 mV (max)
(Voltage Across RS) 75 mV (min)
Ileak Leak Current on Flagout pin Vflagout = VDD 2μA (max)
ISW SW Current Limit 2 2.7 A (max)
1.2 A (min)
TSD Thermal Shutdown Temperature 150 °C (min)
Vos Output Offset Voltage 5 25 mV
VFB Feedback Voltage SD Boost = VDD 1.23 1.15 V (min)
SD Amp = VDD 1.31 V (max)
(1) Typicals are measured at 25°C and represent the parametric norm.
(2) Limits are specified to AOQL (Average Outgoing Quality Level).
(3) Datasheet min/max specification limits are specified by design, test, or statistical analysis.
(4) Shutdown current is measured in a normal room environment. The Shutdown pin should be driven as close as possible to Vin for
minimum shutdown current.
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0.05
0.2
THD+N (%)
OUTPUT VOLTAGE SWING (Vrms)
50m
20m
1
10
10
1200m
0.5
2
5
0.1
0.02
0.01 500m 2 5
f = 100 Hz
f = 1 kHz
f = 10 kHz
0.05
0.2
THD+N (%)
OUTPUT VOLTAGE SWING (Vrms)
50m
20m
1
10
10
1200m
0.5
2
5
0.1
0.02
0.01 500m 2 5
f = 100 Hz
f = 1 kHz
f = 10 kHz
0.05
0.2
THD+N (%)
FREQUENCY (Hz)
300
100
1
10
10k
4k
1k
0.5
2
5
0.1
0.02
0.01 2k
0.05
0.2
THD+N (%)
OUTPUT VOLTAGE SWING (Vrms)
50m
20m
1
10
10
1200m
0.5
2
5
0.1
0.02
0.01 500m 2 5
f = 100 Hz
f = 1 kHz
f = 10 kHz
0.05
0.2
THD+N (%)
FREQUENCY (Hz)
300
100
1
10
10k
4k
1k
0.5
2
5
0.1
0.02
0.01 2k
0.05
0.2
THD+N (%)
FREQUENCY (Hz)
300
100
1
10
10k
4k
1k
0.5
2
5
0.1
0.02
0.01 2k
LM4962
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Typical Performance Characteristics
THD+N vs Frequency THD+N vs Frequency
VDD = 3.2V, VO= 4.95VRMS, ZL= 2μF+9.4ΩVDD = 4.2V, VO= 4.95VRMS, ZL= 2μF+9.4Ω
Figure 4. Figure 5.
THD+N vs Frequency THD+N vs Output Voltage Swing
VDD = 5V, VO= 4.95VRMS, ZL= 2μF+9.4ΩVDD = 3.2V, ZL= 2μF+9.4Ω, f = 1kHz
Figure 6. Figure 7.
THD+N vs Output Voltage Swing THD+N vs Output Voltage Swing
VDD = 4.2V, ZL= 2μF+9.4Ω, f = 1kHz VDD = 5V, ZL= 2μF+9.4Ω, f = 1kHz
Figure 8. Figure 9.
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80
90
BOOST EFFICIENCY (%)
OUTPUT VOLTAGE SWING (Vrms)
10
100
4
3
2
95
85
75
70 5 6
VDD = 5V
VDD = 4.2V
VDD = 3V
0
100
INDUCTOR CURRENT (mA)
OUTPUT VOLTAGE SWING (Vrms)
10
200
432
150
50
5 6
VDD = 4.2V
VDD = 5V
VDD = 3V
20 100 1k 20k
FREQUENCY (Hz)
-28
-24
-20
-16
-12
-8
-4
0
4
8
12
16
20
OUTPUT LEVEL (dB)
2k 5k 10k500200
50
Ci = 1.0 PF
Ci = 0.39 PF
Ci = 0.039 PF
-80
-60
PSRR (dB)
FREQUENCY (Hz)
100
20
-40
-10
0
20k
5k1k
-50
-30
-20
-70
-90
-100
-80
-60
PSRR (dB)
FREQUENCY (Hz)
100
20
-40
-10
0
20k
5k1k
-50
-30
-20
-70
-90
-100
-80
-60
PSRR (dB)
FREQUENCY (Hz)
100
20
-40
-10
0
20k
5k1k
-50
-30
-20
-70
-90
-100
LM4962
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Typical Performance Characteristics (continued)
PSRR vs Frequency PSRR vs Frequency
VDD = 3.2, ZL= 2μF+9.4Ω, VRIPPLE = 200mVP-P VDD = 4.2, ZL= 2μF+9.4Ω, VRIPPLE = 200mVP-P
Figure 10. Figure 11.
PSRR vs Frequency
VDD = 5, ZL= 2μF+9.4Ω, VRIPPLE = 200mVP-P Frequency Response vs Input Capacitor Size
Figure 12. Figure 13.
Boost Efficiency vs Output Voltage Swing Inductor Current vs Output Voltage Swing
f = 1kHz, ZL= 2μF+9.4Ωf = 1kHz, ZL= 2μF+9.4Ω
Figure 14. Figure 15.
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100
200
VOCP (V)
VAMP (V)
4
2 10
86
250
150
50
0
400
800
Rds(on) (m:)
BOOTSTRAP VOLTAGE (V)
3.52.5
1200
6.5
5.5
4.5
1000
600
200
07.5 8.5
1.21
1.23
FEEDBACK VOLTAGE (V)
TEMPERATURE (°C)
-40
-60
1.25
200-20
1.24
1.22
1.20
1.19 40 60 80 100
1.26
4
8
SUPPLY CURRENT (mA)
SUPPLY VOLTAGE (V)
3.02.5
12
4.54.03.5
10
6
2
05.0 5.5
LM4962
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Typical Performance Characteristics (continued)
Supply Current vs Supply Voltage Feedback Voltage vs Temperature
Figure 16. Figure 17.
VOCP vs Vamp Rds(on) vs VBOOTSTRAP
Figure 18. Figure 19.
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APPLICATION INFORMATION
BRIDGE CONFIGURATION EXPLANATION
The Audio Amplifier portion of the LM4962 has two internal amplifiers allowing different amplifier configurations.
The first amplifier’s gain is externally configurable, whereas the second amplifier is internally fixed in a unity-gain,
inverting configuration. The closed-loop gain of the first amplifier is set by selecting the ratio of Rf to Ri while the
second amplifier’s gain is fixed by the two internal 20kΩresistors. Figure 2 shows that the output of amplifier one
serves as the input to amplifier two. This results in both amplifiers producing signals identical in magnitude, but
out of phase by 180°. Consequently, the differential gain for the Audio Amplifier is
AVD = 2 *(Rf/Ri) (1)
By driving the load differentially through outputs Vo1 and Vo2, an amplifier configuration commonly referred to as
“bridged mode” is established. Bridged mode operation is different from the classic single-ended amplifier
configuration where one side of the load is connected to ground.
A bridge amplifier design has a few distinct advantages over the single-ended configuration. It provides
differential drive to the load, thus doubling the output swing for a specified supply voltage.
BOOST CONVERTER POWER DISSIPATION
At higher duty cycles, the increased ON-time of the switch FET means the maximum output current will be
determined by power dissipation within the LM4962 FET switch. The switch power dissipation from ON-time
conduction is calculated by Equation (2).
PD(SWITCH) = DC x IIND(AVE)2x RDS(ON) (2)
where:
DC is the duty cycle.
There will be some switching losses as well, so some derating needs to be applied when calculating IC power
dissipation.
MAXIMUM AMPLIFIER POWER DISSIPATION
Power dissipation is a major concern when designing a successful amplifier, whether the amplifier is bridged or
single-ended. A direct consequence of the increased power delivered to the load by a bridge amplifier is an
increase in internal power dissipation. Since the amplifier portion of the LM4962 has two operational amplifiers,
the maximum internal power dissipation is 4 times that of a single-ended amplifier. The maximum power
dissipation for a given BTL application can be derived from Equation (3).
PDMAX(AMP) = (2VDD 2) / (π2RL) (3)
where:
RL= Ro1+Ro2
MAXIMUM TOTAL POWER DISSIPATION
The total power dissipation for the LM4962 can be calculated by adding Equation (2) and Equation (3) together
to establish Equation (4):
PDMAX(TOTAL) = (2VDD 2) / (π2EFF2RL) (4)
where:
EFF = Efficiency of boost converter
RL= Ro1+Ro2
The result from Equation (4) must not be greater than the power dissipation that results from Equation (5):
PDMAX = (TJMAX - TA) / θJA (5)
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For the LQA28A, θJA = 73°C/W. TJMAX = 125°C for the LM4962. Depending on the ambient temperature, TA, of
the system surroundings, Equation (5) can be used to find the maximum internal power dissipation supported by
the IC packaging. If the result of Equation (4) is greater than that of Equation (5), then either the supply voltage
must be increased, the load impedance increased or TAreduced. For typical applications, power dissipation is
not an issue. Power dissipation is a function of output power and thus, if typical operation is not around the
maximum power dissipation point, the ambient temperature may be increased accordingly.
START-UP SEQUENCE
For the LM4962 correct start-up sequencing is important for optimal device performance. Using the correct start
up sequence will improve click/pop performance as well as avoid transients that could reduce battery life. For
ringer/loudspeaker mode, the supply voltage should be applied first and both the boost converter and the
amplifier should be in shutdown. The boost converter can then be activated followed by the amplifier (see timing
diagram, Figure 20). If the boost converter shutdown is toggled while the amplifier is active a very audible pop
will be heard.
SHUTDOWN FUNCTION
In many applications, a microcontroller or microprocessor output is used to control the shutdown circuitry to
provide a quick, smooth transition into shutdown. Another solution is to use a single-pole, single-throw switch
connected between VDD and Shutdown pins.
BAND SWITCH FUNCTION
The LM4962 features a Band Switch function which allows the user to use one amplifier for both receiver
(earpiece) mode and ringer/loudspeaker mode. When the boost converter and the amplifier are both active the
device is is in ringer mode. This enables the boost converter and sets the externally configurable closed loop
gain selection to BW1. If the boost converter is in the shutdown and the amplifier is active the device is in
receiver mode. In this mode the gain selection is switched to BW2. This allows the amplifier to be powered
directly from the battery minus the voltage drop across the Schottky diode.
SD Boost SD Amp
Receiver Mode (BW2) Low High
Boosted Ringer Mode (BW1) High High
Shutdown Low Low
BOOTSTRAP PIN
The bootstrap pin, featured in the LM4962, provides a voltage supply for the internal switch driver. Connecting
the bootstrap pin to V1 (See Figure 2) allows for a higher voltage to drive the gate of the switch thereby reducing
the Ron. This configuration is necessary in applications with heavier loads. The bootstrap pin can be connected
to VDD when driving lighter loads to improve device performance (Iddq, THD+N, Noise, etc.).
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Vdd
0V
Battery
Voltage
Vdd
0V
Boost
Converter
Shutdown
Vdd
0V
Amplifier
Shutdown
Vdd-Vd*
V1
V1 (output of
Boost
Converter)
(Vdd-Vd)/2*
0V
Vo1/Vo2
(Amplifier
DC Bias)
Vamp/2
Ringer Mode Receiver Mode
*Vd = Voltage drop across diode D2
LM4962
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Figure 20. Power on Sequence Timing Diagram
OVER-CURRENT AND OVER-VOLTAGE PROTECTION FUNCTION
Flagout PinThe Flagout pin indicates a fault when an over current or over voltage condition has been detected.
The Flagout pin is high impedance when inactive. When active, the Flagout pin is pulled down to a 50Ω
short to GND.
Over-Voltage Protection (OVP) OperationWhen a voltage (Vamp) greater than 8.5V (min) is detected at the
OC/OV Detect pin, the LM4962 indicates a fault by activating the Flagout pin. The boost converter
momentarily shutdown and reinitialize the soft-start sequence. The Flagout pin will remain active until both
shutdowns pins are pulled low.
Over-Current Protection (OCP) OperationThe OCP circuitry monitors the voltage across Rocd to detect the
output current of the boost converter. If a voltage greater than 185mV (typ) is detected the device will
shutdown and the Flagout pin will be activated. For the device to return to normal operation both shutdown
pins need to be pulled low to reset the Flagout pin.
Disable OCPThe Over-Current Protection Circuitry can be disabled by shorting out RS. In this configuration the
OVP circuitry is still active.
Disable both OVP and OCPBoth features can be disabled by grounding the OC/OV Detect pin. In this
configuration the Flagout pin will be inactive.
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Vdd
0V
Flagout
t1
Vdd
0V
Amplifier
Shutdown
Vdd
0V
Boost
Shutdown
On
Off
Internal
Boost
Operation
On
Off
Internal
Amplifier
Operation
t1 < 3 Ps
* Vscd refers to the voltage differential across Rs
185 mV
0V
Vscd *
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Timing Diagrams
Figure 21. OCP Timing Diagram
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8.5V
Vdd
0V
Flagout
t2
0V
Vdd
7.5V
Vamp
Vdd
0V
Amplifier
Shutdown
Vdd
0V
Boost
Shutdown
On
Off
Internal
Boost
Operation
On
Off
Internal
Amplifier
Operation
= 3 ms
= 16 ms
<3 Ps
t2t3
t3t1
t3t1
t2
t1
t3
LM4962
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Figure 22. OVP Timing Diagram
PROPER SELECTION OF EXTERNAL COMPONENTS
Proper selection of external components in applications using integrated power amplifiers, and switching DC-DC
converters, is critical for optimizing device and system performance. Consideration to component values must be
used to maximize overall system quality.
The best capacitors for use with the switching converter portion of the LM4962 are multi-layer ceramic
capacitors. They have the lowest ESR (equivalent series resistance) and highest resonance frequency, which
makes them optimum for high frequency switching converters.
When selecting a ceramic capacitor, only X5R and X7R dielectric types should be used. Other types such as
Z5U and Y5F have such severe loss of capacitance due to effects of temperature variation and applied voltage,
they may provide as little as 20% of rated capacitance in many typical applications. Always consult capacitor
manufacturer’s data curves before selecting a capacitor. High-quality ceramic capacitors can be obtained from
Taiyo-Yuden.
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POWER SUPPLY BYPASSING
As with any amplifier, proper supply bypassing is critical for low noise performance and high power supply
rejection. The capacitor location on both V1 and VDD pins should be as close to the device as possible.
SELECTING INPUT CAPACITOR FOR AUDIO AMPLIFIER
One of the major considerations is the closedloop bandwidth of the amplifier. To a large extent, the bandwidth is
dictated by the choice of external components shown in Figure 2. The input coupling capacitor, Ci, forms a first
order high pass filter which limits low frequency response. This value should be chosen based on needed
frequency response for a few distinct reasons.
High value input capacitors are both expensive and space hungry in portable designs. Clearly, a certain value
capacitor is needed to couple in low frequencies without severe attenuation. But ceramic speakers used in
portable systems, whether internal or external, have little ability to reproduce signals below 100Hz to 150Hz.
Thus, using a high value input capacitor may not increase actual system performance.
In addition to system cost and size, click and pop performance is affected by the value of the input coupling
capacitor, Ci. A high value input coupling capacitor requires more charge to reach its quiescent DC voltage
(nominally 1/2 VDD). This charge comes from the output via the feedback and is apt to create pops upon device
enable. Thus, by minimizing the capacitor value based on desired low frequency response, turn-on pops can be
minimized.
SELECTING FEEDBACK CAPACITOR FOR AUDIO AMPLIFIER
The LM4962 is unity-gain stable which gives the designer maximum system flexability. However, to drive ceramic
speakers, a typical application requires a closed-loop differential gain of 10. In this case a feedback capacitor
(Cf2) will be needed as shown in Figure 2 to bandwidth limit the amplifier.
This feedback capacitor creates a low pass filter that eliminates possible high frequency noise. Care should be
taken when calculating the -3dB frequency because an incorrect combination of Rfand Cf2 will cause rolloff
before the desired frequency
SELECTING OUTPUT CAPACITOR (C2) FOR BOOST CONVERTER
A single 4.7µF to 10µF ceramic capacitor will provide sufficient output capacitance for most applications. If larger
amounts of capacitance are desired for improved line support and transient response, tantalum capacitors can
be used. Aluminum electrolytics with ultra low ESR such as Sanyo Oscon can be used, but are usually
prohibitively expensive. Typical AI electrolytic capacitors are not suitable for switching frequencies above 500
kHz because of significant ringing and temperature rise due to self-heating from ripple current. An output
capacitor with excessive ESR can also reduce phase margin and cause instability.
In general, if electrolytics are used, we recommended that they be paralleled with ceramic capacitors to reduce
ringing, switching losses, and output voltage ripple.
SELECTING INPUT CAPACITOR (Cs1) FOR BOOST CONVERTER
An input capacitor is required to serve as an energy reservoir for the current which must flow into the coil each
time the switch turns ON. This capacitor must have extremely low ESR, so ceramic is the best choice. We
recommend a nominal value of 4.7µF, but larger values can be used. Since this capacitor reduces the amount of
voltage ripple seen at the input pin, it also reduces the amount of EMI passed back along that line to other
circuitry.
SETTING THE OUTPUT VOLTAGE (V1) OF BOOST CONVERTER
The output voltage is set using the external resistors R2and R5(see Figure 2). A value of approximately 25kΩis
recommended for R2 to establish the open loop gain of the boost converter.
V1= VFB [1 + (R2/ R5)] (6)
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 15
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LM4962
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www.ti.com
FEED-FORWARD COMPENSATION FOR BOOST CONVERTER
Although the LM4962's internal Boost converter is internally compensated, the external feed-forward capacitor Cf
is required for stability (see Figure 2). Adding this capacitor puts a zero in the loop response of the converter.
The recommended frequency for the zero fz should be approximately 60kHz. C3 can be calculated using the
formula:
C3= 1 / (2πx R2x fz) (7)
SELECTING A SOFT-START CAPACITOR (Css)
The soft-start function charges the boost converter reference voltage slowly, which allows the output of the boost
converter to ramp up slowly thus limiting the transient current at startup.
Selecting a soft-start capacitor (Css) value presents a trade off between the wake-up time of the boost converter
(TWUBC) and the startup transient current. Using a larger capacitor value will increase wake-up time and decrease
startup transient current; on the flip side, using a smaller capacitor value will decrease wake-up time and
increase the transient current seen at startup. A standard rule of thumb is to use a capacitor 1000 times smaller
than the output capacitance of the boost converter (C2+Cs2). A 10nF soft-start capacitor is recommended for a
typical application.
SELECTING A VALUE FOR Rchg
The audio power amplifier integrated in the LM4962 is designed for very fast turn on time. The Cchg pin allows
the input capacitor (CInA) to charge quickly to improve click/pop performance. Resistor, Rchg, protects the Cchg
pin from any over/under voltage conditions caused by excessive input signal, or an active input signal when the
device is in shutdown. The recommended value for Rchg is 1kΩ. If the input signal is less than VDD+0.3V and
greater than -0.3V, and if the input signal is disabled when in shutdown mode, Rchg may be shorted.
SELECTING DIODES
The external diode used in Figure 2 should be a Schottky diode. A 20V diode such as the MBR0520 from
Fairchild Semiconductor is recommended.
The MBR05XX series of diodes are designed to handle a maximum average current of 0.5A. For applications
exceeding 0.5A average but less than 1A, a Microsemi UPS5817 can be used.
DUTY CYCLE
The maximum duty cycle of the boost converter determines the maximum boost ratio of output-to-input voltage
that the converter can attain in continuous mode of operation. The duty cycle for a given boost application is
defined as:
Duty Cycle = (VOUT + VDIODE - VDD)/(VAMP + VDIODE - VSW)
This applies for continuous mode operation.
INDUCTANCE VALUE
The first question we are usually asked is: “How small can I make the inductor.” (because they are the largest
sized component and usually the most costly). The answer is not simple and involves trade-offs in performance.
Larger inductors mean less inductor ripple current, which typically means less output voltage ripple (for a given
size of output capacitor). Larger inductors also mean more load power can be delivered because the energy
stored during each switching cycle is:
E = L/2 x (lp)2 (8)
where:
“lp” is the peak inductor current
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LM4962
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SNAS300D –NOVEMBER 2005–REVISED APRIL 2013
An important point to observe is that the LM4962 will limit its switch current based on peak current. This means
that since lp(max) is fixed, increasing L will increase the maximum amount of power available to the load.
Conversely, using too little inductance may limit the amount of load current which can be drawn from the output.
Best performance is usually obtained when the converter is operated in “continuous” mode at the load current
range of interest, typically giving better load regulation and less output ripple. Continuous operation is defined as
not allowing the inductor current to drop to zero during the cycle. It should be noted that all boost converters shift
over to discontinuous operation as the output load is reduced far enough, but a larger inductor stays “continuous”
over a wider load current range. Taiyo-Yudens NR4012 inductor series is recommended.
MAXIMUM SWITCH CURRENT
The maximum FET switch current available before the current limiter cuts in is dependent on duty cycle of the
application. This is illustrated in a graph in the Typical Performance Characteristics section which shows typical
values of switch current as a function of effective (actual) duty cycle.
CALCULATING OUTPUT CURRENT OF BOOST CONVERTER (IAMP)
The load current of the Boost Converter is related to the average inductor current by the relation:
IAMP = IIND(AVG) x (1 - DC) (9)
Where "DC" is the duty cycle of the application. The switch current can be found by:
ISW = IIND(AVG) + 1/2 (IRIPPLE) (10)
Inductor ripple current is dependent on inductance, duty cycle, supply voltage and frequency:
IRIPPLE = DC x (VDD-VSW) / (f x L) (11)
combining all terms, we can develop an expression which allows the maximum available load current to be
calculated:
IAMP(max) = (1–DC)x(ISW(max)–DC(VDD-VSW))/2fL (12)
The equation shown to calculate maximum load current takes into account the losses in the inductor or turn-OFF
switching losses of the FET and diode.
DESIGN PARAMETERS VSW AND ISW
The value of the FET "ON" voltage (referred to as VSW in Equations (9) thru (12) is dependent on load current. A
good approximation can be obtained by multiplying the "ON Resistance" of the FET times the average inductor
current.
The maximum peak switch current the device can deliver is dependent on duty cycle.
INDUCTOR SUPPLIERS
The recommended inductors for the LM4962 is the Taiyo-Yuden NR4012. When selecting an inductor, make
certain that the continuous current rating is high enough to avoid saturation at peak currents. A suitable core type
must be used to minimize core (switching) losses, and wire power losses must be considered when selecting the
current rating.
Copyright © 2005–2013, Texas Instruments Incorporated Submit Documentation Feedback 17
Product Folder Links: LM4962
LM4962
SNAS300D –NOVEMBER 2005–REVISED APRIL 2013
www.ti.com
REVISION HISTORY
Rev Date Description
1.0 03/31/06 Edited 20142203 and 06, then re-released D/S to the WEB.
Changes from Revision C (April 2013) to Revision D Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 17
18 Submit Documentation Feedback Copyright © 2005–2013, Texas Instruments Incorporated
Product Folder Links: LM4962
PACKAGE OPTION ADDENDUM
www.ti.com 5-Apr-2013
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish MSL Peak Temp
(3)
Op Temp (°C) Top-Side Markings
(4)
Samples
LM4962TL/NOPB ACTIVE DSBGA YZR 20 250 Green (RoHS
& no Sb/Br) SNAGCU Level-1-260C-UNLIM GF7
LM4962TLX/NOPB ACTIVE DSBGA YZR 20 3000 Green (RoHS
& no Sb/Br) SNAGCU Level-1-260C-UNLIM GF7
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a
continuation of the previous line and the two combined represent the entire Top-Side Marking for that device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM4962TL/NOPB DSBGA YZR 20 250 178.0 8.4 2.18 2.69 0.76 4.0 8.0 Q1
LM4962TLX/NOPB DSBGA YZR 20 3000 178.0 8.4 2.18 2.69 0.76 4.0 8.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 8-Apr-2013
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM4962TL/NOPB DSBGA YZR 20 250 210.0 185.0 35.0
LM4962TLX/NOPB DSBGA YZR 20 3000 210.0 185.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 8-Apr-2013
Pack Materials-Page 2
MECHANICAL DATA
YZR0020xxx
www.ti.com
TLA20XXX (Rev D)
0.600±0.075 D
E
A
. All linear dimensions are in millimeters. Dimensioning and tolerancing per ASME Y14.5M-1994.
B. This drawing is subject to change without notice.
4215053/A 12/12
NOTES:
D: Max =
E: Max =
2.49 mm, Min =
1.99 mm, Min =
2.43 mm
1.93 mm
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