Low Cost, 80 MHz FastFET Op Amps AD8033/AD8034 CONNECTION DIAGRAMS AD8033 8 NC 7 +VS +IN 3 6 VOUT -VS 4 5 NC NC = NO CONNECT AD8033 VOUT 1 5 +VS 4 -IN -VS 2 +IN 3 Figure 1. 8-Lead SOIC (R) 02924-002 NC 1 -IN 2 02924-001 FET input amplifier 1 pA typical input bias current Very low cost High speed 80 MHz, -3 dB bandwidth (G = +1) 80 V/s slew rate (G = +2) Low noise 11 nV/Hz (f = 100 kHz) 0.7 fA/Hz (f = 100 kHz) Wide supply voltage range: 5 V to 24 V Low offset voltage: 1 mV typical Single-supply and rail-to-rail output High common-mode rejection ratio: -100 dB Low power: 3.3 mA/amplifier typical supply current No phase reversal Small packaging: 8-lead SOIC, 8-lead SOT-23, and 5-lead SC70 Figure 2. 5-Lead SC70 (KS) VOUT1 1 8 +VS -IN1 2 7 VOUT2 +IN1 3 6 -IN2 -VS 4 5 +IN2 AD8034 02924-003 FEATURES Figure 3. 8-Lead SOIC (R) and 8-Lead SOT-23 (RJ) 24 21 VOUT = 200mV p-p G = +10 18 15 APPLICATIONS 12 GAIN (dB) Instrumentation Filters Level shifting Buffering G = +5 9 6 G = +2 3 G = +1 0 -3 The AD8033/AD8034 FastFETTM amplifiers are voltage feedback amplifiers with FET inputs, offering ease of use and excellent performance. The AD8033 is a single amplifier and the AD8034 is a dual amplifier. The AD8033/AD8034 FastFET op amps in Analog Devices, Inc., proprietary XFCB process offer significant performance improvements over other low cost FET amps, such as low noise (11 nV/Hz and 0.7 fA/Hz) and high speed (80 MHz bandwidth and 80 V/s slew rate). With a wide supply voltage range from 5 V to 24 V and fully operational on a single supply, the AD8033/AD8034 amplifiers work in more applications than similarly priced FET input amplifiers. In addition, the AD8033/AD8034 have rail-to-rail outputs for added versatility. G = -1 -6 -9 0.1 1 10 FREQUENCY (MHz) 100 1000 02924-004 GENERAL DESCRIPTION Figure 4. Small Signal Frequency Response The AD8033/AD8034 amplifiers only draw 3.3 mA/amplifier of quiescent current while having the capability of delivering up to 40 mA of load current. The AD8033 is available in a small package 8-lead SOIC and a small package 5-lead SC70. The AD8034 is also available in a small package 8-lead SOIC and a small package 8-lead SOT-23. They are rated to work over the industrial temperature range of -40C to +85C without a premium over commercial grade products. Despite their low cost, the amplifiers provide excellent overall performance. They offer a high common-mode rejection of -100 dB, low input offset voltage of 2 mV maximum, and low noise of 11 nV/Hz. Rev. D Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c)2002-2008 Analog Devices, Inc. All rights reserved. AD8033/AD8034 TABLE OF CONTENTS Features .............................................................................................. 1 Input Overdrive .......................................................................... 16 Applications ....................................................................................... 1 Input Impedance ........................................................................ 16 General Description ......................................................................... 1 Thermal Considerations............................................................ 16 Connection Diagrams ...................................................................... 1 Layout, Grounding, and Bypassing Considerations .................. 18 Revision History ............................................................................... 2 Bypassing ..................................................................................... 18 Specifications..................................................................................... 3 Grounding ................................................................................... 18 Absolute Maximum Ratings............................................................ 6 Leakage Currents ........................................................................ 18 Maximum Power Dissipation ..................................................... 6 Input Capacitance ...................................................................... 18 Output Short Circuit .................................................................... 6 Applications Information .............................................................. 19 ESD Caution .................................................................................. 6 High Speed Peak Detector ........................................................ 19 Typical Performance Characteristics ............................................. 7 Active Filters ............................................................................... 20 Test Circuits ..................................................................................... 14 Wideband Photodiode Preamp ................................................ 21 Theory of Operation ...................................................................... 16 Outline Dimensions ....................................................................... 23 Output Stage Drive and Capacitive Load Drive ..................... 16 Ordering Guide .......................................................................... 24 REVISION HISTORY 9/08--Rev. C to Rev. D Deleted Usable Input Range Parameter, Table 1 ........................... 3 Deleted Usable Input Range Parameter, Table 2 ........................... 4 Deleted Usable Input Range Parameter, Table 3 ........................... 5 4/08--Rev. B to Rev. C Changes to Format ............................................................. Universal Changes to Features and General Description ............................. 1 Changes to Figure 13 Caption and Figure 14 Caption ................ 8 Changes to Figure 22 and Figure 23 ............................................... 9 Changes to Figure 25 and Figure 28 ............................................. 10 Changes to Input Capacitance Section ........................................ 18 Changes to Active Filters Section ................................................. 21 Changes to Outline Dimensions................................................... 23 Changes to Ordering Guide .......................................................... 24 8/02--Rev. 0 to Rev. A Added AD8033 ................................................................... Universal VOUT = 2 V p-p Deleted from Default Conditions ......... Universal Added SOIC-8 (R) and SC70 (KS) ..................................................1 Edits to General Description Section .............................................1 Changes to Specifications .................................................................2 New Figure 2 ......................................................................................5 Edits to Maximum Power Dissipation Section ..............................5 Changes to Ordering Guide .............................................................5 Change to TPC 3 ...............................................................................6 Change to TPC 6 ...............................................................................6 Change to TPC 9 ...............................................................................7 New TPC 16 .......................................................................................8 New TPC 17 .......................................................................................8 New TPC 31 .................................................................................... 11 New TPC 35 .................................................................................... 11 New Test Circuit 9 .......................................................................... 13 SC70 (KS) Package Added ............................................................ 19 2/03--Rev. A to Rev. B Changes to Features.......................................................................... 1 Changes to Connection Diagrams ................................................. 1 Changes to Specifications ................................................................ 2 Changes to Absolute Maximum Ratings ....................................... 4 Replaced TPC 31............................................................................. 11 Changes to TPC 35 ......................................................................... 11 Changes to Test Circuit 3 ............................................................... 12 Updated Outline Dimensions ....................................................... 19 Rev. D | Page 2 of 24 AD8033/AD8034 SPECIFICATIONS TA = 25C, VS = 5 V, RL = 1 k, gain = +2, unless otherwise noted. Table 1. Parameter DYNAMIC PERFORMANCE -3 dB Bandwidth Input Overdrive Recovery Time Output Overdrive Recovery Time Slew Rate (25% to 75%) Settling Time to 0.1% NOISE/HARMONIC PERFORMANCE Distortion Second Harmonic Third Harmonic Crosstalk, Output-to-Output Input Voltage Noise Input Current Noise DC PERFORMANCE Input Offset Voltage Conditions Min Typ G = +1, VOUT = 0.2 V p-p G = +2, VOUT = 0.2 V p-p G = +2, VOUT = 2 V p-p -6 V to +6 V input -3 V to +3 V input, G = +2 G = +2, VOUT = 4 V step G = +2, VOUT = 2 V step G = +2, VOUT = 8 V step 65 80 30 21 135 135 80 95 225 MHz MHz MHz ns ns V/s ns ns -82 -85 -70 -81 -86 11 0.7 dBc dBc dBc dBc dB nV/Hz fA/Hz 55 fC = 1 MHz, VOUT = 2 V p-p RL = 500 RL = 1 k RL = 500 RL = 1 k f = 1 MHz, G = +2 f = 100 kHz f = 100 kHz VCM = 0 V TMIN - TMAX 1 Input Offset Voltage Match Input Offset Voltage Drift Input Bias Current Open-Loop Gain INPUT CHARACTERISTICS Common-Mode Input Impedance Differential Input Impedance Input Common-Mode Voltage Range FET Input Range Common-Mode Rejection Ratio OUTPUT CHARACTERISTICS Output Voltage Swing Output Short-Circuit Current Capacitive Load Drive POWER SUPPLY Operating Range Quiescent Current per Amplifier Power Supply Rejection Ratio TMIN - TMAX VOUT = 3 V 89 VCM = -3 V to +1.5 V -89 4.75 30% overshoot, G = +1, VOUT = 400 mV p-p 4 1.5 50 92 -90 Rev. D | Page 3 of 24 2 3.5 2.5 27 11 Unit mV mV mV V/C pA pA dB 1000||2.3 1000||1.7 G||pF G||pF -5.0 to +2.2 -100 V dB 4.95 40 35 V mA pF 5 VS = 2 V Max 3.3 -100 24 3.5 V mA dB AD8033/AD8034 TA = 25C, VS = 5 V, RL = 1 k, gain = +2, unless otherwise noted. Table 2. Parameter DYNAMIC PERFORMANCE -3 dB Bandwidth Input Overdrive Recovery Time Output Overdrive Recovery Time Slew Rate (25% to 75%) Settling Time to 0.1% NOISE/HARMONIC PERFORMANCE Distortion Second Harmonic Third Harmonic Crosstalk, Output to Output Input Voltage Noise Input Current Noise DC PERFORMANCE Input Offset Voltage Conditions Min Typ G = +1, VOUT = 0.2 V p-p G = +2, VOUT = 0.2 V p-p G = +2, VOUT = 2 V p-p -3 V to +3 V input -1.5 V to +1.5 V input, G = +2 G = +2, VOUT = 4 V step G = +2, VOUT = 2 V step 70 80 32 21 180 200 70 100 MHz MHz MHz ns ns V/s ns -80 -84 -70 -80 -86 11 0.7 dBc dBc dBc dBc dB nV/Hz fA/Hz 55 fC = 1 MHz, VOUT = 2 V p-p RL = 500 RL = 1 k RL = 500 RL = 1 k f = 1 MHz, G = +2 f = 100 kHz f = 100 kHz VCM = 0 V TMIN - TMAX 1 Input Offset Voltage Match Input Offset Voltage Drift Input Bias Current Open-Loop Gain INPUT CHARACTERISTICS Common-Mode Input Impedance Differential Input Impedance Input Common-Mode Voltage Range FET Input Range Common-Mode Rejection Ratio OUTPUT CHARACTERISTICS Output Voltage Swing Output Short-Circuit Current Capacitive Load Drive POWER SUPPLY Operating Range Quiescent Current per Amplifier Power Supply Rejection Ratio TMIN - TMAX VOUT = 0 V to 3 V 87 VCM = 1.0 V to 2.5 V -80 RL = 1 k 0.16 to 4.83 30% overshoot, G = +1, VOUT = 400 mV p-p 4 1 50 92 -80 Rev. D | Page 4 of 24 2 3.5 2.5 30 10 Unit mV mV mV V/C pA pA dB 1000||2.3 1000||1.7 G||pF G||pF 0 to 2.0 -100 V dB 0.04 to 4.95 30 25 V mA pF 5 VS = 1 V Max 3.3 -100 24 3.5 V mA dB AD8033/AD8034 TA = 25C, VS = 12 V, RL = 1 k, gain = +2, unless otherwise noted. Table 3. Parameter DYNAMIC PERFORMANCE -3 dB Bandwidth Input Overdrive Recovery Time Output Overdrive Recovery Time Slew Rate (25% to 75%) Settling Time to 0.1% NOISE/HARMONIC PERFORMANCE Distortion Second Harmonic Third Harmonic Crosstalk, Output to Output Input Voltage Noise Input Current Noise DC PERFORMANCE Input Offset Voltage Conditions Min Typ G = +1, VOUT = 0.2 V p-p G = +2, VOUT = 0.2 V p-p G = +2, VOUT = 2 V p-p -13 V to +13 V input -6.5 V to +6.5 V input, G = +2 G = +2, VOUT = 4 V step G = +2, VOUT = 2 V step G = +2, VOUT = 10 V step 65 80 30 21 100 100 80 90 225 MHz MHz MHz ns ns V/s ns ns -80 -82 -70 -82 -86 11 0.7 dBc dBc dBc dBc dB nV/Hz fA/Hz 55 fC = 1 MHz, VOUT = 2 V p-p RL = 500 RL = 1 k RL = 500 RL = 1 k f = 1 MHz, G = +2 f = 100 kHz f = 100 kHz VCM = 0 V TMIN - TMAX 1 Input Offset Voltage Match Input Offset Voltage Drift Input Bias Current Open-Loop Gain INPUT CHARACTERISTICS Common-Mode Input Impedance Differential Input Impedance Input Common-Mode Voltage Range FET Input Range Common-Mode Rejection Ratio OUTPUT CHARACTERISTICS Output Voltage Swing Output Short-Circuit Current Capacitive Load Drive POWER SUPPLY Operating Range Quiescent Current per Amplifier Power Supply Rejection Ratio TMIN - TMAX VOUT = 8 V VCM = 5 V 88 -92 11.52 30% overshoot, G = +1 4 2 50 96 -85 Rev. D | Page 5 of 24 2 3.5 2.5 24 12 Unit mV mV mV V/C pA pA dB 1000||2.3 1000||1.7 G||pF G||pF -12.0 to +9.0 -100 V dB 11.84 60 35 V mA pF 5 VS = 2 V Max 3.3 -100 24 3.5 V mA dB AD8033/AD8034 ABSOLUTE MAXIMUM RATINGS Rating 26.4 V See Figure 5 26.4 V 1.4 V -65C to +125C -40C to +85C 300C PD = (VS x IS) + (VS/4)2/RL In single-supply operation with RL referenced to VS-, worst case is VOUT = VS/2. 2.0 Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. MAXIMUM POWER DISSIPATION 1.5 SOT-23-8 1.0 SC70-5 0.5 0 -60 The maximum safe power dissipation in the AD8033/AD8034 packages is limited by the associated rise in junction temperature (TJ) on the die. The plastic that encapsulates the die locally reaches the junction temperature. At approximately 150C, which is the glass transition temperature, the plastic changes its properties. Even temporarily exceeding this temperature limit can change the stresses that the package exerts on the die, permanently shifting the parametric performance of the AD8033/ AD8034. Exceeding a junction temperature of 175C for an extended period can result in changes in silicon devices, potentially causing failure. The still-air thermal properties of the package and PCB (JA), ambient temperature (TA), and the total power dissipated in the package (PD) determine the junction temperature of the die. The junction temperature can be calculated as TJ = TA + (PD x JA) PD is the sum of the quiescent power dissipation and the power dissipated in the package due to the load drive for all outputs. The quiescent power is the voltage between the supply pins (VS) times the quiescent current (IS). Assuming the load (RL) is referenced to midsupply, the total drive power is VS/2 x IOUT, some of which is dissipated in the package and some in the load (VOUT x IOUT). The difference between the total drive power and the load power is the drive power dissipated in the package SOIC-8 -40 -20 0 20 40 60 AMBIENT TEMPERATURE (C) 80 100 02924-005 Parameter Supply Voltage Power Dissipation Common-Mode Input Voltage Differential Input Voltage Storage Temperature Range Operating Temperature Range Lead Temperature (Soldering 10 sec) If the rms signal levels are indeterminate, consider the worst case, when VOUT = VS/4 for RL to midsupply MAXIMUM POWER DISSIPATION (W) Table 4. Figure 5. Maximum Power Dissipation vs. Ambient Temperature for a 4-Layer Board Airflow increases heat dissipation, effectively reducing JA. In addition, more metal directly in contact with the package leads from metal traces, through holes, ground, and power planes reduces the JA. Care must be taken to minimize parasitic capacitances at the input leads of high speed op amps as discussed in the Layout, Grounding, and Bypassing Considerations section. Figure 5 shows the maximum power dissipation in the package vs. the ambient temperature for the 8-lead SOIC (125C/W), 5-lead SC70 (210C/W), and 8-lead SOT-23 (160C/W) packages on a JEDEC standard 4-layer board. JA values are approximations. OUTPUT SHORT CIRCUIT Shorting the output to ground or drawing excessive current for the AD8033/AD8034 will likely cause catastrophic failure. ESD CAUTION PD = Quiescent Power + (Total Drive Power - Load Power) PD = [VS x IS] + [(VS/2) x (VOUT/RL)] - [VOUT2/RL] RMS output voltages should be considered. If RL is referenced to -VS, as in single-supply operation, the total drive power is VS x IOUT. Rev. D | Page 6 of 24 AD8033/AD8034 TYPICAL PERFORMANCE CHARACTERISTICS Default conditions: VS = 5 V, CL = 5 pF, RL = 1 k, TA = 25C. 24 21 8 VOUT = 200mV p-p G = +10 G = +2 7 18 12 5 GAIN (dB) 9 G = +2 6 VOUT = 1V p-p 4 3 3 G = +1 0 VOUT = 4V p-p 2 -3 G = -1 -9 0.1 1 1 100 10 FREQUENCY (MHz) 1000 VOUT = 2V p-p 0 0.1 02924-006 -6 1 10 FREQUENCY (MHz) 100 02924-009 GAIN (dB) VOUT = 0.2V p-p 6 G = +5 15 Figure 9. Frequency Response for Various Output Amplitudes (See Figure 45) Figure 6. Small Signal Frequency Response for Various Gains 8 1 VS = +5V 7 0 VS = 5V 6 5 VS = 12V -2 GAIN (dB) -3 -4 VS = 12V 1 G = +1 VOUT = 200mV p-p 0.1 1 10 FREQUENCY (MHz) 100 0 1 10 FREQUENCY (MHz) 100 VS = 12V 6 VS = 12V VS = 5V VS = +5V 5 GAIN (dB) VS = 5V VS = +5V -1 -2 -3 4 3 2 -4 1 -5 1 10 FREQUENCY (MHz) 100 G = +2 VOUT = 2V p-p 02924-008 GAIN (dB) 1 7 G = +1 VOUT = 2V p-p 0 -6 0.1 0.1 Figure 10. Small Signal Frequency Response for Various Supplies (See Figure 45) Figure 7. Small Signal Frequency Response for Various Supplies (See Figure 44) 2 G = +2 VOUT = 200mV p-p 02924-010 -6 3 2 02924-007 -5 VS = 5V 4 0 0.1 Figure 8. Large Signal Frequency Response for Various Supplies (See Figure 44) 1 10 FREQUENCY (MHz) 100 Figure 11. Large Signal Frequency Response for Various Supplies (See Figure 45) Rev. D | Page 7 of 24 02924-011 GAIN (dB) -1 VS = +5V AD8033/AD8034 8 10 VOUT = 200mV p-p G = +1 CL = 100pF 9 6 VOUT = 200mV p-p G = +2 CL = 100pF 8 CL = 100pF RSNUB = 25 2 0 6 5 CL = 33pF CL = 2pF 3 CL = 2pF 2 -4 1 100 10 0 0.1 02924-012 0.1 Figure 12. Small Signal Frequency Response for Various CL (See Figure 44) 8 CF = 0pF 7 CF = 1pF 100 VOUT = 200mV p-p G = +2 RL = 1k 7 6 6 5 GAIN (dB) CF = 1.5pF 5 CF = 2pF 4 RL = 500 4 3 3 2 2 1 10 FREQUENCY (MHz) 100 0 0.1 02924-013 0 0.1 Figure 13. Small Signal Frequency Response for Various CF (See Figure 45) 1 10 FREQUENCY (MHz) 100 Figure 16. Small Signal Frequency Response for Various RL (See Figure 45) 100 VOUT = 200mV p-p VS = 12V 180 150 80 10 1 G = +1 60 120 40 90 PHASE 20 60 0 30 PHASE (Degrees) GAIN G = +2 GAIN (dB) IMPEDANCE () 02924-016 1 1 100 10 FREQUENCY (MHz) Figure 15. Small Signal Frequency Response for Various CL (See Figure 45) 9 VOUT = 200mV p-p RF = 3k 8 G = +2 1 02924-015 1 FREQUENCY (MHz) GAIN (dB) CL = 33pF 4 -2 -6 CL = 51pF 7 GAIN (dB) GAIN (dB) 4 1k 10k 100k 1M 10M 100M FREQUENCY (Hz) -20 100 02924-014 0.01 100 1k 10k 100k 1M FREQUENCY (Hz) Figure 14. Output Impedance vs. Frequency (See Figure 47) Figure 17. Open-Loop Response Rev. D | Page 8 of 24 10M 0 100M 02924-017 0.1 AD8033/AD8034 -40 G = +2 HD3 RL = 500 -50 -60 -60 -70 -70 -80 DISTORTION (dBc) DISTORTION (dBc) -50 HD3 RL = 1k -90 HD2 RL = 500 -100 -110 HD3 G = +2 -90 HD2 G = +2 -100 -110 HD3 G = +1 HD2 RL = 1k 1 FREQUENCY (MHz) 5 -120 0.1 02924-018 -120 0.1 -20 G = +2 HD3 VS = 5V -50 HD3 VOUT = 10V p-p -50 -70 DISTORTION (dBc) DISTORTION (dBc) G = +2 -40 HD2 V S = 5V HD3 VS = 24V -90 5 -30 -60 -80 1 FREQUENCY (MHz) Figure 21. Harmonic Distortion vs. Frequency for Various Gains Figure 18. Harmonic Distortion vs. Frequency for Various Loads (See Figure 45) -40 HD2 G = +1 -80 02924-021 -40 -100 HD2 VOUT = 20V p-p HD3 V OUT = 20V p-p -60 -70 HD2 V OUT = 10V p-p -80 -90 HD3 VOUT = 2V p-p -100 HD2 VS = 24V -110 1 FREQUENCY (MHz) 5 02924-019 -120 0.1 HD2 V OUT = 2V p-p -120 0.1 Figure 19. Harmonic Distortion vs. Frequency for Various Supply Voltages (See Figure 45) 1 FREQUENCY (MHz) Figure 22. Harmonic Distortion vs. Frequency for Various Amplitudes (See Figure 45), VS = 24 V 1000 80 G = +1 VS = +5V POSITIVE SIDE PERCENT OVERSHOOT (%) 70 100 60 VS = +5V NEGATIVE SIDE 50 40 VS = 5V NEGATIVE SIDE 30 20 VS = 5V POSITIVE SIDE 10 10 100 1k 10k 100k 1M FREQUENCY (Hz) 1 0M 10M 100M Figure 20. Voltage Noise vs. Frequency 0 10 30 50 70 CAPACITIVE LOAD (pF) 90 110 Figure 23. Percent Overshoot vs. Capacitive Load (See Figure 44) Rev. D | Page 9 of 24 02924-023 10 02924-020 NOISE (nV/Hz) 5 02924-022 -110 AD8033/AD8034 G = +1 20ns/DIV 25mV/DIV 02924-024 38pF 15pF 80mV/DIV Figure 24. Small Signal Transient Response 5 V (See Figure 44) 80ns/DIV 02924-027 G = +1 Figure 27. Small Signal Transient Response 5 V (See Figure 44) G = +1 VOUT = 8V p-p VOUT = 8V p-p VOUT = 2V p-p VOUT = 2V p-p 3V/DIV 320ns/DIV 02924-025 VOUT = 20V p-p 320ns/DIV 3V/DIV Figure 25. Large Signal Transient Response (See Figure 44) 02924-028 G = +2 VOUT = 20V p-p Figure 28. Large Signal Transient Response (See Figure 45) G = -1 G = +1 350ns/DIV 1.5V/DIV Figure 26. Output Overdrive Recovery (See Figure 46) VIN 350ns/DIV Figure 29. Input Overdrive Recovery (See Figure 44) Rev. D | Page 10 of 24 02924-029 1.5V/DIV VOUT VOUT 02924-026 VIN AD8033/AD8034 VIN = 1V VIN = 1V VOUT - 2VIN +0.1% +0.1% 02924-030 1.5s/DIV 2mV/DIV Figure 33. 0.1% Short-Term Settling Time 7.0 0 6.9 QUIESCENT SUPPLY CURRENT (mA) -5 -10 -Ib -20 +Ib -25 -30 -35 6.6 6.5 VS = 5V 6.4 6.3 VS = +5V 6.2 6.1 30 35 40 45 50 55 60 65 TEMPERATURE (C) 70 75 80 85 -20 0 20 40 TEMPERATURE (C) 80 Figure 34. Quiescent Supply Current vs. Temperature for Various Supply Voltages 4.0 BJT INPUT RANGE 3.5 30 18 NORMALIZED OFFSET (mV) -Ib 24 +Ib 12 6 -Ib VS = 12V 3.0 2.5 2.0 1.5 1.0 0.5 VS = 5V VS = +5V 0 -0.5 0 2 4 6 8 COMMON-MODE VOLTAGE (V) 10 12 02924-032 0 FET INPUT RANGE 10 +Ib 5 0 -5 -10 -15 -20 -25 -30 -12 -10 -8 -6 -4 -2 60 02924-034 25 02924-031 20 5.9 -40 36 Ib (A) VS = 12V 6.0 Figure 31. Ib vs. Temperature Ib (pA) 6.8 6.7 Figure 32. Ib vs. Common-Mode Voltage Range -1.0 -14 -12 -10 -8 -6 -4 -2 0 2 4 6 8 COMMON-MODE VOLTAGE (V) 10 12 14 Figure 35. Input Offset Voltage vs. Common-Mode Voltage Rev. D | Page 11 of 24 02924-035 Ib (pA) -15 42 -0.1% 20ns/DIV 2mV/DIV Figure 30. Long-Term Settling Time -40 VOUT - 2VIN t=0 02924-033 -0.1% t=0 AD8033/AD8034 105 -20 100 -30 OPEN-LOOP GAIN (dB) 95 CMRR (dB) -40 -50 -60 90 RL = 500 85 RL = 1k RL = 2k 80 75 70 -70 1 10 FREQUENCY (MHz) 100 60 -12 -10 02924-036 -80 0.1 -4 -2 0 2 4 OUTPUT VOLTAGE (V) 6 8 10 12 -40 1.0 -50 0.8 SOT-23 A/B VCC - VOH CROSSTALK (dB) 0.6 0.4 VOL - VEE 0.2 -60 SOIC A/B -70 SOT-23 B/A SOIC B/A -80 -90 5 10 15 20 25 30 ILOAD (mA) -100 0.1 02924-037 0 1 FREQUENCY (MHz) 10 50 02924-040 OUTPUT SATURATION (V) -6 Figure 39. Open-Loop Gain vs. Output Voltage for Various RL Figure 36. CMRR vs. Frequency (See Figure 50) 0 -8 02924-039 65 Figure 40. Crosstalk (See Figure 52) Figure 37. Output Saturation Voltage vs. Load Current 0 180 -10 150 -20 -30 FREQUENCY PSRR (dB) -PSRR -40 -50 +PSRR -60 120 90 60 -70 -80 30 0.001 0.01 0.1 1 10 100 FREQUENCY (MHz) Figure 38. PSRR vs. Frequency (See Figure 49 and Figure 51) 0 -1.5 -1.0 -0.5 0 0.5 VOS (mV) Figure 41. Initial Offset Rev. D | Page 12 of 24 1.0 1.5 02924-041 -100 0.0001 02924-038 -90 AD8033/AD8034 VOUT 1s/DIV 1.2V/DIV VIN 1s/DIV Figure 43. G = +2 Response, VS = 5 V Figure 42. G = +1 Response, VS = 5 V Rev. D | Page 13 of 24 02924-043 VIN 02924-042 1.2V/DIV VOUT AD8033/AD8034 TEST CIRCUITS +VS +VS 1F 1F + + 10nF 10nF RSNUB VIN AD8033/AD8034 49.9 976 CLOAD VOUT AD8033/AD8034 49.9 10nF 10nF VSINE 0.2V p-p 02924-047 -VS - + 1F 02924-044 + 1F + -VS Figure 47. Output Impedance, G = +1 Figure 44. G = +1 CF 1k 1k RF +VS 1k +VS 1F 1F + + 10nF 499 VIN 49.9 RSNUB 976 AD8033/AD8034 CLOAD 10nF VOUT AD8033/AD8034 49.9 10nF 10nF VSINE 0.2V p-p -VS -VS Figure 45. G = +2 1k Figure 48. Output Impedance, G = +2 1k +VS 1F + 10nF 976 AD8033/AD8034 499 VOUT 49.9 10nF + 1F -VS 02924-046 VIN - + 1F 02924-045 + 1F + Figure 46. G = -1 Rev. D | Page 14 of 24 02924-048 1k AD8033/AD8034 1V p-p +VS - + +VS AC 1F + +VS 49.9 10nF AD8033/AD8034 VOUT VOUT AD8033/AD8034 10nF 02924-049 49.9 -VS AC + 1F 02924-051 -VS 1V p-p + - -VS Figure 49. Negative PSRR 1k Figure 51. Positive PSRR 1k 1k 1k -VS +VS - 1F TO PORT 1 499 + 50 VIN - + 49.9 10nF 976 1k AD8033/AD8034 VOUT 499 TO PORT 2 1k -VS +VS + 49.9 10nF + 1F 1k -VS A - +VS 02924-050 1k + B 1k Figure 50. CMRR Figure 52. Crosstalk Rev. D | Page 15 of 24 1k 02924-052 VIN AD8033/AD8034 THEORY OF OPERATION The incorporation of JFET devices into the Analog Devices high voltage XFCB process has enabled the ability to design the AD8033/AD8034. The AD8033/AD8034 are voltage feedback rail-to-rail output amplifiers with FET inputs and a bipolarenhanced common-mode input range. The use of JFET devices in high speed amplifiers extends the application space into both the low input bias current and low distortion, high bandwidth areas. Using N-channel JFETs and a folded cascade input topology, the common-mode input level operates from 0.2 V below the negative rail to within 3.0 V of the positive rail. Cascading of the input stage ensures low input bias current over the entire common-mode range as well as CMRR and PSRR specifications that are above 90 dB. Additionally, long-term settling issues that normally occur with high supply voltages are minimized as a result of the cascading. OUTPUT STAGE DRIVE AND CAPACITIVE LOAD DRIVE The common emitter output stage adds rail-to-rail output performance and is compensated to drive 35 pF (30% overshoot at G = +1). Additional capacitance can be driven if a small snub resistor is put in series with the capacitive load, effectively decoupling the load from the output stage, as shown in Figure 12. The output stage can source and sink 20 mA of current within 500 mV of the supply rails and 1 mA within 100 mV of the supply rails. INPUT OVERDRIVE An additional feature of the AD8033/AD8034 is a bipolar input pair that adds rail-to-rail common-mode input performance specifically for applications that cannot tolerate phase inversion problems. Under normal common-mode operation, the bipolar input pair is kept reversed, maintaining Ib at less than 1 pA. When the input common-mode operation comes within 3.0 V of the positive supply rail, I1 turns off and I4 turns on, supplying tail current to the bipolar pair Q25 and Q27. With this configuration, the inputs can be driven beyond the positive supply rail without any phase inversion (see Figure 53). As a result of entering the bipolar mode of operation, an offset and input bias current shift occurs (see Figure 32 and Figure 35). After re-entering the JFET common-mode range, the amplifier recovers in approximately 100 ns (refer to Figure 29 for input overload behavior). Above and below the supply rails, ESD protection diodes activate, resulting in an exponentially increasing input bias current. If the inputs are driven well beyond the rails, series input resistance should be included to limit the input bias current to <10 mA. INPUT IMPEDANCE The input capacitance of the AD8033/AD8034 forms a pole with the feedback network, resulting in peaking and ringing in the overall response. The equivalent impedance of the feedback network should be kept small enough to ensure that the parasitic pole falls well beyond the -3 dB bandwidth of the gain configuration being used. If larger impedance values are desired, the amplifier can be compensated by placing a small capacitor in parallel with the feedback resistor. Figure 13 shows the improvement in frequency response by including a small feedback capacitor with high feedback resistance values. THERMAL CONSIDERATIONS Because the AD8034 operates at up to 12 V supplies in the small 8-lead SOT-23 package (160C/W), power dissipation can easily exceed package limitations, resulting in permanent shifts in device characteristics and even failure. Likewise, high supply voltages can cause an increase in junction temperature even with light loads, resulting in an input bias current and offset drift penalty. The input bias current doubles for every 10C shown in Figure 31. Refer to the Maximum Power Dissipation section for an estimation of die temperature based on load and supply voltage. Rev. D | Page 16 of 24 AD8033/AD8034 +VS I2 R2 R3 V2 + - Q4 + - Q1 Q13 Q7 Q6 VTH V4 Q14 R14 J1 D4 Q25 J2 Q27 VCC +IN -IN D5 VOUT Q11 Q9 Q29 R7 I4 I3 R8 02924-053 I1 Q28 -VS Figure 53. Simplified AD8033/AD8034 Input Stage Rev. D | Page 17 of 24 AD8033/AD8034 LAYOUT, GROUNDING, AND BYPASSING CONSIDERATIONS BYPASSING LEAKAGE CURRENTS Power supply pins are actually inputs, and care must be taken so that a noise-free stable dc voltage is applied. The purpose of bypass capacitors is to create low impedances from the supply to ground at all frequencies, thereby shunting or filtering a majority of the noise. Decoupling schemes are designed to minimize the bypassing impedance at all frequencies with a parallel combination of capacitors. The chip capacitors, 0.01 F or 0.001 F (X7R or NPO), are critical and should be placed as close as possible to the amplifier package. Larger chip capacitors, such as the 0.1 F capacitor, can be shared among a few closely spaced active components in the same signal path. The 10 F tantalum capacitor is less critical for high frequency bypassing, and in most cases, only one per board is needed at the supply inputs. Poor PCB layout, contaminants, and the board insulator material can create leakage currents that are much larger than the input bias currents of the AD8033/AD8034. Any voltage differential between the inputs and nearby runs set up leakage currents through the PCB insulator, for example, 1 V/100 G = 10 pA. Similarly, any contaminants on the board can create significant leakage (skin oils are a common problem). To significantly reduce leakages, put a guard ring (shield) around the inputs and input leads that is driven to the same voltage potential as the inputs. This way there is no voltage potential between the inputs and surrounding area to set up any leakage currents. For the guard ring to be completely effective, it must be driven by a relatively low impedance source and should completely surround the input leads on all sides, above, and below using a multilayer board. GROUNDING A ground plane layer is important in densely packed PCBs to spread the current, thereby minimizing parasitic inductances. However, an understanding of where the current flows in a circuit is critical to implementing effective high speed circuit design. The length of the current path is directly proportional to the magnitude of the parasitic inductances and, thus, the high frequency impedance of the path. High speed currents in an inductive ground return create unwanted voltage noise. The length of the high frequency bypass capacitor leads is most critical. A parasitic inductance in the bypass grounding works against the low impedance created by the bypass capacitor. Place the ground leads of the bypass capacitors at the same physical location. Because load currents flow from the supplies as well, the ground for the load impedance should be at the same physical location as the bypass capacitor grounds. For the larger value capacitors that are intended to be effective at lower frequencies, the current return path distance is less critical. Another effect that can cause leakage currents is the charge absorption of the insulator material itself. Minimizing the amount of material between the input leads and the guard ring helps to reduce the absorption. In addition, low absorption materials such as Teflon(R) or ceramic may be necessary in some instances. INPUT CAPACITANCE Along with bypassing and ground, high speed amplifiers can be sensitive to parasitic capacitance between the inputs and ground. A few pF of capacitance reduces the input impedance at high frequencies, in turn it increases the gain of the amplifier and can cause peaking of the overall frequency response or even oscillations if severe enough. It is recommended that the external passive components that are connected to the input pins be placed as close as possible to the inputs to avoid parasitic capacitance. The ground and power planes must be kept at a distance of at least 0.05 mm from the input pins on all layers of the board. Rev. D | Page 18 of 24 AD8033/AD8034 APPLICATIONS INFORMATION Using two amplifiers, the difference between the peak and the current input level is forced across R2 instead of either amplifier's input pins. In the event of a rising pulse, the first amplifier compensates for the drop across D2 and D3, forcing the voltage at Node 3 equal to Node 1. D1 is off and the voltage drop across R2 is zero. Capacitor C3 speeds up the loop by providing the charge required by the input capacitance of the first amplifier, helping to maintain a minimal voltage drop across R2 in the sampling mode. A negative going edge results in D2 and D3 turning off and D1 turning on, closing the loop around the first amplifier and forcing VOUT - VIN across R2. R4 makes the voltage across D2 zero, minimizing leakage current and kickback from D3 from affecting the voltage across C2. HIGH SPEED PEAK DETECTOR The low input bias current and high bandwidth of the AD8033/ AD8034 make the parts ideal for a fast settling, low leakage peak detector. The classic fast-low leakage topology with a diode in the output is limited to ~1.4 V p-p maximum in the case of the AD8033/AD8034 because of the protection diodes across the inputs, as shown in Figure 54. AD8033/ AD8034 VOUT VIN 02924-054 ~1.4V p-p MAX The rate of the incoming edge must be limited so that the output of the first amplifier does not overshoot the peak value of VIN before the output of the second amplifier can provide negative feedback at the summing junction of the first amplifier. This is accomplished with the combination of R1 and C1, which allows the voltage at Node 1 to settle to 0.1% of VIN in 270 ns. The selection of C2 and R3 is made by considering droop rate, settling time, and kickback. R3 prevents overshoot from occurring at Node 3. The time constants of R1, C1 and R3, C2 are roughly equal to achieve the best performance. Slower time constants can be selected by increasing C2 to minimize droop rate and kickback at the cost of increased settling time. R1 and C1 should also be increased to match, reducing the incoming pulse's effect on kickback. Figure 54. High Speed Peak Detector with Limited Input Range Using the AD8033/AD8034, a unity gain peak detector can be constructed that captures a 300 ns pulse while still taking advantage of the low input bias current and wide commonmode input range of the AD8033/AD8034, as shown in Figure 55. C3 10pF R2 1k D1 LS4148 C4 4.7pF R4 6k +VS +VS 1/2 R1 1k R5 49.9 AD8034 D3 D2 LS4148 LS4148 AD8034 C1 39pF/ 120pF -VS VOUT -VS C2 180pF/ 560pF R3 200 02924-056 VIN 1/2 Figure 55. High Speed, Unity Gain Peak Detector Using AD8034 Rev. D | Page 19 of 24 AD8033/AD8034 The Sallen-Key topology is the least dependent on the active device, requiring that the bandwidth be flat to beyond the stopband frequency because it is used simply as a gain block. In the case of high Q filter stages, the peaking must not exceed the openloop bandwidth and the linear input range of the amplifier. INPUT OUTPUT Using an AD8033/AD8034, a 4-pole cascaded Sallen-Key filter can be constructed with fC = 1 MHz and over 80 dB of stop-band attenuation, as shown in Figure 58. 2 02924-055 C3 33pF R1 4.22k VIN Figure 56. Peak Detector Response 4 V, 300 ns Pulse Figure 56 shows the peak detector in Figure 55 capturing a 300 ns, 4 V pulse with 10 mV of kickback and a droop rate of 5 V/s. For larger peak-to-peak pulses, increase the time constants of R1, C1 and R3, C3 to reduce overshoot. The best droop rate occurs by isolating parasitic resistances from Node 3, which can be accomplished using a guard band connected to the output of the second amplifier that surrounds its summing junction (Node 3). Increasing both time constants by a factor of 3 permits a larger peak pulse to be captured and increases the output accuracy. R2 6.49k R5 49.9 AD8034 -VS C1 27pF C4 82pF R4 4.99k +VS 1/2 R3 4.99k AD8034 -VS C2 10pF INPUT VOUT 02924-058 1V/DIV 100ns/DIV +VS 1/2 Figure 58. 4-Pole Cascade Sallen-Key Filter Component values are selected using a normalized cascaded, 2-stage Butterworth filter table and Sallen-Key 2-pole active filter equations. The overall frequency response is shown in Figure 59. OUTPUT 0 -10 2 02924-057 -20 1V/DIV 200ns/DIV Figure 57. Peak Detector Response 5 V, 1 s Pulse Figure 57 shows a 5 V peak pulse being captured in 1 s with less than 1 mV of kickback. With this selection of time constants, up to a 20 V peak pulse can be captured with no overshoot. REF LEVEL (dB) -30 -40 -50 -60 -70 -80 ACTIVE FILTERS -90 -100 10k Rev. D | Page 20 of 24 100k 1M 10M FREQUENCY (Hz) Figure 59. 4-Pole Cascade Sallen-Key Filter Response 02924-059 The response of an active filter varies greatly depending on the performance of the active device. Open-loop bandwidth and gain, along with the order of the filter, determines the stop-band attenuation as well as the maximum cutoff frequency, while input capacitance can set a limit on which passive components are used. Topologies for active filters are varied, and some are more dependent on the performance of the active device than others are. AD8033/AD8034 Filter cutoff frequencies can be increased beyond 1 MHz using the AD8033/AD8034 but limited open-loop gain and input impedance begin to interfere with the higher Q stages. This can cause early roll-off of the overall response. Additionally, the stop-band attenuation decreases with decreasing open-loop gain. Keeping these limitations in mind, a 2-pole Sallen-Key Butterworth filter with fC = 4 MHz can be constructed that has a relatively low Q of 0.707 while still maintaining 15 dB of attenuation an octave above fC and 35 dB of stop-band attenuation. The filter and response are shown in Figure 60 and Figure 61, respectively. C3 22pF R2 2.49k AD8033 VOUT = I PHOTO x R F 1 + sC F R F where IPHOTO is the output current of the photodiode, and the parallel combination of RF and CF sets the signal bandwidth. CF RF IPHOTO CD CM CS VOUT -VS C1 10pF R5 49.9 The basic transfer function is RSH = 1011 +VS R1 2.49k Figure 62 shows an I/V converter with an electrical model of a photodiode. CM 02924-060 VIN WIDEBAND PHOTODIODE PREAMP VB Figure 60. 2-Pole Butterworth Active Filter CF + CS VOUT RF 02924-062 When selecting components, the common-mode input capacitance must be taken into consideration. Figure 62. Wideband Photodiode Preamp 5 The stable bandwidth attainable with this preamp is a function of RF, the gain bandwidth product of the amplifier, and the total capacitance at the summing junction of the amplifier, including CS and the amplifier input capacitance. RF and the total capacitance produce a pole in the loop transmission of the amplifier that can result in peaking and instability. Adding CF creates a zero in the loop transmission that compensates for the effect of the pole and reduces the signal bandwidth. It can be shown that the signal bandwidth resulting in a 45phase margin (f(45)) is defined by the expression 0 -5 -15 -20 -25 -30 -35 f ( 45) = -40 -45 100k 1M 10M FREQUENCY (Hz) Figure 61. 2-Pole Butterworth Active Filter Response 100M 02924-061 GAIN (dB) -10 f CR 2 x R F x C S where: fCR is the amplifier crossover frequency. RF is the feedback resistor. CS is the total capacitance at the amplifier summing junction (amplifier + photodiode + board parasitics). The value of CF that produces f(45) is CF = CS 2 x R F x f CR The frequency response in this case shows about 2 dB of peaking and 15% overshoot. Doubling CF and cutting the bandwidth in half results in a flat frequency response, with about 5% transient overshoot. Rev. D | Page 21 of 24 AD8033/AD8034 Keeping the input terminal impedances matched is recommended to eliminate common-mode noise peaking effects that add to the output noise. Integrating the square of the output voltage noise spectral density over frequency and then taking the square root results in the total rms output noise of the preamp. f1 = 1 2R F (CF + CS + CM + 2CD) f2 = 1 2RF CF f f3 = (C + C + CR S M 2CD + CF)/CF RF NOISE f2 VEN (CF + CS + CM + 2CD)/CF f3 f1 VEN NOISE DUE TO AMPLIFIER FREQUENCY (Hz) Figure 63. Photodiode Voltage Noise Contributions Rev. D | Page 22 of 24 02924-063 The pole in the loop transmission translates to a zero in the noise gain of the amplifier, leading to an amplification of the input voltage noise over frequency. The loop transmission zero introduced by CF limits the amplification. The bandwidth of the noise gain extends past the preamp signal bandwidth and is eventually rolled off by the decreasing loop gain of the amplifier. VOLTAGE NOISE (nV/Hz) The output noise over frequency of the preamp is shown in Figure 63. AD8033/AD8034 OUTLINE DIMENSIONS 5.00 (0.1968) 4.80 (0.1890) 5 1 6.20 (0.2441) 5.80 (0.2284) 4 1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040) 1.75 (0.0688) 1.35 (0.0532) 0.51 (0.0201) 0.31 (0.0122) COPLANARITY 0.10 SEATING PLANE 0.50 (0.0196) 0.25 (0.0099) 45 8 0 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157) COMPLIANT TO JEDEC STANDARDS MS-012-A A CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. 012407-A 8 4.00 (0.1574) 3.80 (0.1497) Figure 64. 8-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-8) Dimensions shown in millimeters and (inches) 2.20 2.00 1.80 1.35 1.25 1.15 5 4 1 2 3 PIN 1 2.40 2.10 1.80 0.65 BSC 1.00 0.90 0.70 1.10 0.80 0.30 0.15 0.10 MAX 0.40 0.10 0.46 0.36 0.26 0.22 0.08 SEATING PLANE 0.10 COPLANARITY COMPLIANT TO JEDEC STANDARDS MO-203-AA Figure 65. 5-Lead Thin Shrink Small Outline Transistor Package [SC70] (KS-5) Dimensions shown in millimeters 2.90 BSC 8 7 6 5 1 2 3 4 1.60 BSC 2.80 BSC PIN 1 INDICATOR 0.65 BSC 1.95 BSC 1.30 1.15 0.90 1.45 MAX 0.15 MAX 0.38 0.22 0.22 0.08 SEATING PLANE 8 4 0 COMPLIANT TO JEDEC STANDARDS MO-178-BA Figure 66. 8-Lead Small Outline Transistor Package [SOT-23] (RJ-8) Dimensions shown in millimeters Rev. D | Page 23 of 24 0.60 0.45 0.30 AD8033/AD8034 ORDERING GUIDE Model AD8033AR AD8033AR-REEL AD8033AR-REEL7 AD8033ARZ 1 AD8033ARZ-REEL1 AD8033ARZ-REEL71 AD8033AKS-R2 AD8033AKS-REEL AD8033AKS-REEL7 AD8033AKSZ-R21 AD8033AKSZ-REEL1 AD8033AKSZ-REEL71 AD8034AR AD8034AR-REEL7 AD8034AR-REEL AD8034ARZ1 AD8034ARZ-REEL1 AD8034ARZ-REEL71 AD8034ART-R2 AD8034ART-REEL AD8034ART-REEL7 AD8034ARTZ-R21 AD8034ARTZ-REEL1 AD8034ARTZ-REEL71 AD8034CHIPS 1 Temperature Range -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C Package Description 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 5-Lead SC70 5-Lead SC70 5-Lead SC70 5-Lead SC70 5-Lead SC70 5-Lead SC70 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOT-23 8-Lead SOT-23 8-Lead SOT-23 8-Lead SOT-23 8-Lead SOT-23 8-Lead SOT-23 DIE Z = RoHS Compliant Part, # denotes RoHS compliant product may be top or bottom marked. (c)2002-2008 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D02924-0-9/08(D) Rev. D | Page 24 of 24 Package Option R-8 R-8 R-8 R-8 R-8 R-8 KS-5 KS-5 KS-5 KS-5 KS-5 KS-5 R-8 R-8 R-8 R-8 R-8 R-8 RJ-8 RJ-8 RJ-8 RJ-8 RJ-8 RJ-8 Branding H3B H3B H3B H3C H3C H3C HZA HZA HZA HZA# HZA# HZA#