RF/IF Vector Multiplier
ADL5390
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
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registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700 www.analog.com
Fax: 781.326.8703 © 2004 Analog Devices, Inc. All rights reserved.
FEATURES
Matched pair of multiplying VGAs
Broad frequency range 20 MHz to 2.4 GHz
Continuous magnitude control from +5 dB to −30 dB
Output third-order intercept 24 dBm
Output 1 dB compression point 11 dBm
Output noise floor −148 dBm/Hz
Adjustable modulation bandwidth up to 230 MHz
Fast output power disable
Single-supply voltage 4.75 V to 5.25 V
APPLICATIONS
PA linearization and predistortion
Amplitude and phase modulation
Variable matched attenuator and/or phase shifter
Cellular base stations
Radio links
Fixed wireless access
Broadband/CATV
RF/IF analog multiplexer
FUNCTIONAL BLOCK DIAGRAM
VPS2OBBMQBBP
INPI
INMI
DSOPIBBMIBBP
RFOP
RFOM
04954-001
VPRF
CMOP
INMQ
INPQ
CMRF
Figure 1.
GENERAL DESCRIPTION
The ADL5390 vector multiplier consists of a matched pair of
broadband variable gain amplifiers whose outputs are summed.
The separate gain controls for each amplifier are linear-in-
magnitude. If the two input RF signals are in quadrature, the
vector multiplier can be configured as a vector modulator or as
a variable attenuator/phase shifter by using the gain control pins
as Cartesian variables. In this case, the output amplitude can be
controlled from a maximum of +5 dB to less than –30 dB, and
the phase can be shifted continuously over the entire 360°
range. Since the signal paths are linear, the original modulation
on the inputs is preserved. If the two signals are independent,
then the vector multiplier can function as a 2:1 multiplexer or
can provide fading from one channel to another.
The ADL5390 operates over a wide frequency range of 20 MHz
to 2400 MHz. For a maximum gain setting on one channel at
380 MHz, the ADL5390 delivers an OP1dB of 11 dBm, an OIP3
of 24 dBm, and an output noise floor of −148 dBm/Hz. The gain
and phase matching between the two VGAs is better than 0.5 dB
and 1°, respectively, over most of the operating range.
The gain control inputs are dc-coupled with a +/−500 mV dif-
ferential full-scale range centered about a 500 mV common
mode. The maximum modulation bandwidth is 230 MHz,
which can be reduced by adding external capacitors to limit the
noise bandwidth on the control lines.
Both the RF inputs and outputs can be used differentially or
single-ended and must be ac-coupled. The impedance of each
VGA RF input is 250 Ω to ground, and the differential output
impedance is nominally 50 Ω over the operating frequency
range. The DSOP pin allows the output stage to be disabled
quickly to protect subsequent stages from overdrive. The
ADL5390 operates off supply voltages from 4.75 V to 5.25 V
while consuming 135 mA.
The ADL5390 is fabricated on Analog Devices’ proprietary,
high performance 25 GHz SOI complementary bipolar IC
process. It is available in a 24-lead, Pb-free CSP package and
operates over a −40°C to +85°C temperature range. Evaluation
boards are available.
ADL5390
Rev. 0 | Page 2 of 24
TABLE OF CONTENTS
Specifications..................................................................................... 3
Absolute Maximum Ratings............................................................ 5
ESD Caution.................................................................................. 5
Pin Configuration and Function Descriptions............................. 6
Typical Performance Characteristics ............................................. 7
General Structure ........................................................................... 11
Theory of Operation .................................................................. 11
Noise and Distortion.................................................................. 11
Applications..................................................................................... 12
Using the ADL5390.................................................................... 12
RF Input and Matching.............................................................. 12
RF Output and Matching .......................................................... 13
Driving the I-Q Baseband Gain Controls............................... 13
Interfacing to High Speed DACs.............................................. 14
Generalized Modulator ............................................................. 15
Vector Modulator ....................................................................... 15
Vector Modulator Example—CDMA2000 ............................. 15
Quadrature Modulator .............................................................. 17
RF Multiplexer............................................................................ 18
Evaluation Board ............................................................................ 19
Outline Dimensions ....................................................................... 23
Ordering Guide .......................................................................... 23
REVISION HISTORY
10/04—Revision 0: Initial Version
ADL5390
Rev. 0 | Page 3 of 24
SPECIFICATIONS
VS = 5 V, TA = 25°C, ZO = 50 Ω, FRF = 380 MHz, single-ended source drive to INPI and INPQ, and INMI and INMQ are ac-coupled to
common, unless otherwise noted. 66.5 Ω termination resistors before ac-coupling capacitors on INPI and INPQ. The specifications refer
to one active channel with the other channel input terminated in 50 Ω. The common-mode level for the gain control inputs is 0.5 V. A
maximum gain setpoint of 1.0 refers to a differential gain control voltage of 0.5 V.
Table 1.
Parameter Conditions Min Typ Max Unit
OVERALL FUNCTION
Frequency Range 20 2400 MHz
Gain Control Range Relative to maximum gain 35 dB
GAIN CONTROL INTERFACE (I and Q) QBBP, QBBM, IBBM, IBBP (Pins 4, 5, 14, 15)
Gain Scaling 3.5 1/V
Modulation Bandwidth 500 mV p-p, sinusoidal baseband input single-
ended
230 MHz
Second Harmonic Distortion 500 mV p-p, 1 MHz, sinusoidal baseband input
differential
45 dBc
Third Harmonic Distortion 500 mV p-p, 1 MHz, sinusoidal baseband input
differential
55 dBc
Step Response For gain from −15 dB to +5 dB 45 ns
For gain from +5 dB to −15 dB 47 ns
FRF = 70 MHz
Maximum Gain Maximum gain setpoint 4.6 dB
Gain Conformance Over gain setpoint of 0.2 to 1.0 0.25 dB
Output Noise Floor Maximum gain setpoint, no RF input −149 dBm/Hz
RF PIN = −5 dBm, frequency offset = 20 MHz −146 dBm/Hz
Output IP3 FRF1 = 70 MHz, FRF2 = 72.5 MHz, maximum gain
setpoint
23 dBm
Output 1 dB Compression Point Maximum gain setpoint 10.7 dBm
Input 1 dB Compression Point Gain setpoint = 0.1 6.7 dBm
Gain Flatness Over any 60 MHz bandwidth 0.25 dB
Gain Matching At maximum gain setpoint 0.5 dB
Phase Matching At maximum gain setpoint ±0.25 Degrees
Input Impedance INPI, INMI, INMQ, INMP (Pins 20, 21, 22, 23) 250||1 Ohms||pF
Output Return Loss RFOP, RFOM (Pins 9, 10) measured through
balun
9.7 dB
FRF = 140 MHz
Maximum Gain Maximum gain setpoint 4.5 dB
Gain Conformance Over gain setpoint of 0.2 to 1.0 0.25 dB
Output Noise Floor Maximum gain setpoint, no RF input −144 dBm/Hz
RF PIN = −5 dBm, frequency offset = 20 MHz −145 dBm/Hz
Output IP3 FRF1 = 140 MHz, FRF2 = 142.5 MHz, maximum
gain setpoint
24.4 dBm
Output 1 dB Compression Point Maximum gain setpoint 11 dBm
Input 1 dB Compression Point Gain setpoint = 0.1 7.1 dBm
Gain Flatness Over any 60 MHz bandwidth 0.25 dB
Gain Matching At maximum gain setpoint 0.5 dB
Phase Matching At maximum gain setpoint ±0.25 Degrees
Input Impedance INPI, INMI, INMQ, INMP (Pins 20, 21, 22, 23) 250||1 Ohms||pF
Output Return Loss RFOP, RFOM (Pins 9, 10) measured through
balun
9.6 dB
FRF = 380 MHz
Maximum Gain Maximum gain setpoint 4.1 dB
Gain Conformance Over gain setpoint of 0.2 to 1.0 0.25 dB
ADL5390
Rev. 0 | Page 4 of 24
Parameter Conditions Min Typ Max Unit
Output Noise Floor Maximum gain setpoint, no RF input −147.5 dBm/Hz
RF PIN = −5 dBm, frequency offset = 20 MHz −146 dBm/Hz
Output IP3 FRF1 = 380 MHz, FRF2 = 382.5 MHz, maximum
gain setpoint
24.2 dBm
Output 1 dB Compression Point Maximum gain setpoint 11.3 dBm
Input 1 dB Compression Point Gain setpoint = 0.1 8.3 dBm
Gain Flatness Over any 60 MHz bandwidth 0.25 dB
Gain Matching At maximum gain setpoint 0.5 dB
Phase Matching At maximum gain setpoint ±0.5 Degrees
Input Impedance INPI, INMI, INMQ, INMP (Pins 20, 21, 22, 23) 200||1 Ohms||pF
Output Return Loss RFOP, RFOM (Pins 9, 10) measured through
balun
8.5 dB
FRF = 900 MHz
Maximum Gain Maximum gain setpoint 4.5 dB
Gain Conformance Over gain setpoint of 0.2 to 1.0 0.4 dB
Output Noise Floor Maximum gain setpoint, no RF input −149.5 dBm/Hz
RF PIN = −5 dBm, frequency offset = 20 MHz −148 dBm/Hz
Output IP3 FRF1 = 900 MHz, FRF2 = 902.5 MHz, maximum
gain setpoint
23.3 dBm
Output 1 dB Compression Point Maximum gain setpoint 11.5 dBm
Input 1 dB Compression Point Gain setpoint = 0.1 8.5 dBm
Gain Flatness Over any 60 MHz bandwidth 0.25 dB
Gain Matching At maximum gain setpoint 0.6 dB
Phase Matching At maximum gain setpoint ±1 Degrees
Input Impedance INPI, INMI, INMQ, INMP (Pins 20, 21, 22, 23) 180||0.6 Ohms||pF
Output Return Loss RFOP, RFOM (Pins 9, 10) measured through
balun
6.8 dB
FRF = 2400 MHz
Maximum Gain Maximum gain setpoint 7.0 dB
Gain Conformance Over gain setpoint of 0.2 to 1.0 0.5 dB
Output Noise Floor Maximum gain setpoint, no RF input −147 dBm/Hz
RF PIN = −5 dBm, frequency offset = 20 MHz −144 dBm/Hz
Output IP3 FRF1 = 2400 MHz, FRF2 = 2402.5 MHz, maximum
gain setpoint
18.7 dBm
Output 1 dB Compression Point Maximum gain setpoint 9.6 dBm
Input 1 dB Compression Point Gain setpoint = 0.1 4.3 dBm
Gain Flatness Over any 60 MHz bandwidth 0.25 dB
Gain Matching At maximum gain setpoint 0.8 dB
Phase Matching At maximum gain setpoint ±2.5 Degrees
Input Impedance INPI, INMI, INMQ, INMP (Pins 20, 21, 22, 23) 140||0.5 Ohms||pF
Output Return Loss RFOP, RFOM (Pins 9, 10) measured through
balun
13.5 dB
POWER SUPPLY VPRF, VPS2 (Pin 1, 18, 6); RFOP, RFOM (Pins 9, 10)
Positive Supply Voltage 4.75 5 5.25 V
Total Supply Current Includes load current 135 mA
OUTPUT DISABLE DSOP (Pin 13)
Disable Threshold 2.5 V
Maximum Attenuation DSOP = 5 V 40 dB
Enable Response Time Delay following high-to-low transition until
device meets full specifications
15 ns
Disable Response Time Delay following low-to-high transition until
device produces full attenuation
10 ns
ADL5390
Rev. 0 | Page 5 of 24
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameters Rating
Supply Voltage VPRF, VPS2 5.5 V
DSOP 5.5 V
IBBP, IBBM, QBBP, QBBM 2.5 V
RFOP, RFOM 5.5 V
RF Input Power at Maximum Gain 10 dBm for 50 Ω
(INPI or INPQ, Single-Ended Drive)
Equivalent Voltage 2.0 V p-p
Internal Power Dissipation 825 mW
θJA (With Pad Soldered to Board) 59°C/W
Maximum Junction Temperature 125°C
Operating Temperature Range −40°C to +85°C
Storage Temperature Range −65°C to +150°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational sec-
tion of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
ADL5390
Rev. 0 | Page 6 of 24
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
24
1
2
3
4
5
6
18
17
16
15
14
13
23 22 21 20 19
7 8 9 101112
TOP VIEW
(Not to Scale)
ADL5390
VPRF
QFLP
QFLM
QBBP
QBBM
VPS2
VPRF
IFLP
IFLM
IBBP
IBBM
DSOP
04954-002
CMRF
INPQ
INMQ
INMI
INPI
CMRF
CMOP
CMOP
RFOP
RFOM
CMOP
CMOP
Figure 2. LFCSP Pin Configuration
Table 3. Pin Function Descriptions
Pin No. Mnemonic Description
2, 3 QFLP, QFLM Q Baseband Input Filter Pins. Connect optional capacitor to reduce Q baseband gain control channel low-
pass corner frequency.
4, 5 QBBP, QBBM Q Channel Differential Baseband Gain Control Inputs. Typical common-mode bias level of 0.5 V.
6, 1, 18 VPS2, VPRF Positive Supply Voltage. VP of 4.75 V to 5.25 V.
7, 8, 11, 12,
19, 24
CMOP, CMRF Device Common. Connect via lowest possible impedance to external circuit common.
9, 10 RFOP, RFOM Differential RF Outputs. Must be ac-coupled. Differential impedance 50 Ω nominal.
13 DSOP Output Disable. Pull high to disable output stage. Connect to common for normal operation.
14, 15 IBBM, IBBP I Channel Differential Baseband Gain Control Inputs. Typical common-mode bias level of 0.5 V.
16, 17 IFLM, IFLP I Baseband Input Filter Pins. Connect optional capacitor to reduce I baseband gain control channel low-
pass corner frequency.
20, 21 INPI, INMI I Channel Differential RF Inputs. Must be ac-coupled. 250 Ω impedance to common on each pin. These
inputs can be driven single-ended without any performance degradation.
22, 23 INMQ, INPQ Q Channel Differential RF Inputs. Must be ac-coupled. 250 Ω impedance to common on each pin. These
inputs can be driven single-ended without any performance degradation.
Exposed
Paddle
GND The exposed paddle on the underside of the package should be soldered to a low thermal and electrical
impedance ground plane.
ADL5390
Rev. 0 | Page 7 of 24
TYPICAL PERFORMANCE CHARACTERISTICS
10
5
0
–5
–10
–15
–20
–25
–30 0.250 0.50 0.75 1.00
GAIN (dB)
GAIN SETPOINT
04954-003
F
RF
= 70MHz
F
RF
= 140MHz
F
RF
= 380MHz
F
RF
= 900MHz
F
RF
= 2400MHz
Figure 3. Gain Magnitude vs. Gain Setpoint, RF Frequency = 70 MHz,
140 MHz, 380 MHz, 900 MHz, 2400 MHz (Channel I or Channel Q)
10
5
0
–5
–10
–15
–20
–25
–300 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
GAIN (dB)
GAIN SETPOINT
TEMP = –40°C
TEMP = +25°C
TEMP = +85°C
04954-004
Figure 4. Gain Magnitude vs. Gain Setpoint, Temp = +85°C, +25°C, −40°C,
RF Frequency = 380 MHz (Channel I or Channel Q)
0 0.25 0.50 0.75 1.0
4
3
2
1
0
–1
–2
–3
–4
GAIN ERROR (dB)
GAIN SETPOINT
F
RF
= 70MHz
F
RF
= 140MHz
F
RF
= 380MHz
F
RF
= 900MHz
F
RF
= 2400MHz
04954-005
+3
σ
= DASH LINE
–3
σ
= SOLID LINE
Figure 5. Gain Conformance Error vs. Gain Setpoint, RF Frequency = 70 MHz,
140 MHz, 380 MHz, 900 MHz, 2400 MHz
5
4
3
2
1
0
–1
–2
–3
–4
–5 0 300 600 900 1200 1500 1800 2100 2400
CHANNEL GAIN MATCH (dB)
FREQUENCY (MHz)
+3
σ
–3
σ
04954-006
Figure 6. Channel Gain Matching (I to Q) vs. RF Frequency,
Gain Setpoint = 1.0
0 300 600 900 1200 1500 1800 2100 2400
9
8
7
6
5
4
3
2
1
0
GAIN (dB)
FREQUENCY (MHz)
TEMP = +85°C
TEMP = +25°C
TEMP = –40°C
04954-007
Figure 7. Channel Gain vs. RF Frequency, Temp = +85°C, +25°C, −40°C,
Gain Setpoint = 1.0
0 0.25 0.50 0.75 1.0
5
0
–5
–10
–15
–20
PHASE ERROR (Degrees)
GAIN SETPOINT
04954-008
F
RF
= 70MHz
F
RF
= 140MHz
F
RF
= 380MHz
F
RF
= 900MHz
F
RF
= 2400MHz
+3
σ
= DASH LINE
–3
σ
= SOLID LINE
Figure 8. Single-Channel Phase Deviation vs. Gain Setpoint, Normalized to
Gain Setpoint = 1.0, RF Frequency = 70 MHz, 140 MHz, 380 MHz,
900 MHz, 2400 MHz
ADL5390
Rev. 0 | Page 8 of 24
25
20
15
10
5
0
–5
–10
–15
–200 0.2 0.4 0.6 0.8 1.0
PHASE DIFFERRENCE (Degrees)
GAIN SETPOINT
04954-009
F
RF
= 70MHz
F
RF
=140MHz
F
RF
=380MHz
F
RF
= 900MHz
F
RF
= 2400MHz
+3
σ
= DASH LINE
–3
σ
= SOLID LINE
Figure 9. Channel-to-Channel Phase Matching vs. Gain Setpoint,
RF Frequency = 70 MHz, 140 MHz, 380 MHz, 900 MHz, 2400 MHz
10
8
6
4
2
0
–2
–4
–6
–8
–10
PHASE DIFFERENCE (Degrees)
0 600 1200 1800 2400
FREQUENCY (MHz)
TEMP = –40°C
TEMP = +25°C
TEMP = +85°C
04954-010
+3
σ
= DASH LINE
–3
σ
= SOLID LINE
Figure 10. Channel-to-Channel Phase Matching vs. RF Frequency, Temp =
+85°C, +25°C, −40°C, Gain Setpoint = 1.0
–142
–143
–144
–145
–146
–147
–148
–149
–150
–151
–152
NOISE (dBm/Hz)
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
GAIN SETPOINT
04954-011
F
RF
= 70MHz
F
RF
= 140MHz
F
RF
= 380MHz
F
RF
= 900MHz
F
RF
= 2400MHz
Figure 11. Output Noise Floor vs. Gain Setpoint, No RF Carrier,
RF Frequency = 70 MHz, 140 MHz, 380 MHz, 900 MHz, 2400 MHz
NO CARRIER
P
IN
=–15dBm
P
IN
=–10dBm
P
IN
= –5dBm
–142
–143
–144
–145
–146
–147
–148
–149
–150
–151
–152
NOISE (dBm/Hz)
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
GAIN SETPOINT
04954-012
Figure 12. Output Noise Floor vs. Gain Setpoint, No Carrier, with Carrier
(20 MHz Offset), RF PIN = −5, −10, −15, No Carrier, RF Frequency = 380 MHz
04954-013
–142
–143
–144
–145
–146
–147
–148
–149
–150
–151
–152
NOISE (dBm/Hz)
0 400 800 1200 1600 2000 2400
GAIN SETPOINT = 1.0
FREQUENCY (MHz)
Figure 13. Output Noise Floor vs. RF Frequency, Gain Setpoint = 1.0,
No RF Carrier
–20
–15
–10
–5
0
5
10
0 400 800 1200 1600 2000 2400
FREQUENCY (MHz)
GAIN (dB)
04954-014
GAIN SETPOINT = 1.0
GAIN SETPOINT = 0.5
GAIN SETPOINT = 0.1
Figure 14. Gain vs. RF Frequency, Gain Setpoint = 1.0, 0.5, 0.1
ADL5390
Rev. 0 | Page 9 of 24
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–1000 100 200 300 400 500 600 700 800 900 1000
RF OUTPUT SIDEBAND POWER (dBm)
FUNDAMENTAL
2ND HARMONIC
3RD HARMONIC
DIFF. BASEBAND INPUT LEVEL (mV p-p)
04954-015
Figure 15. Baseband Harmonic Distortion, (Channel I and Channel Q),
RF PIN = −5 dBm, (Balun and Cable Losses Not Included)
0400 800 1200 1600 2000 2400
12
11
10
9
8
7
6
FREQUENCY (MHz)
OP1dB (dBm)
TEMP = –40°C
TEMP = +25°C
TEMP = +85°C
04954-016
Figure 16. Output 1 dB Compression Point vs. RF Frequency,
Temp = +85°C, +25°C, −40°C, Gain Setpoint = 1.0
0400 800 1200 1600 2000 2400
30
28
26
24
22
20
18
16
14
12
10
TEMP = –40°C
TEMP = +25°C
TEMP = +85°C
FREQUENCY (MHz)
OIP3 (dBm)
04954-017
Figure 17. Output IP3 vs. RF Frequency, Temp = +85°C, +25°C, −40°C,
Gain Setpoint = 1.0
0 50 100 150 200 250 300 350 400
0
–5
–10
–15
–20
–25
–30
–35
1V p-p BASEBAND INPUT
500mV p-p BASEBAND INPUT
250mV p-p BASEBAND INPUT
BB FREQUENCY (MHz)
SIDEBAND POWER (dBm)
04954-018
Figure 18. IQ Modulation Bandwidth vs. Baseband Magnitude
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
15
10
5
0
–5
–10
–15
–20
–25
GAIN SETPOINT
OP1dB (dBm)
04954-019
F
RF
= 70MHz
F
RF
= 140MHz
F
RF
= 380MHz
F
RF
= 900MHz
F
RF
= 2400MHz
Figure 19. Output 1 dB Compression vs. Gain Setpoint, RF Frequency =
70 MHz, 140 MHz, 380 MHz, 900 MHz, 2400 MHz
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
15
10
5
0
–5
–10
–15
20
25
30
OIP3 (dBm)
GAIN SETPOINT
04954-020
F
RF
= 70MHz
F
RF
= 140MHz
F
RF
= 380MHz
F
RF
= 900MHz
F
RF
= 2400MHz
Figure 20. Output IP3 vs. Gain Setpoint, RF Frequency = 70 MHz, 140 MHz,
380 MHz, 900 MHz, 2400 MHz
ADL5390
Rev. 0 | Page 10 of 24
20 220 420 620 820 1020 1220 1420 1620 1820 2020 2220
300
250
200
150
100
50
F
RF
(MHz)
INPUT SHUNT RESISTANCE (Ω)
04954-021
1.25
1.00
0.75
0.50
0.25
0
INPUT SHUNT CAPACITANCE (pF)
SHUNT RESISTANCE (Ω)
SHUNT CAPACITANCE (pF)
Figure 21. S11 of RF Input (Shunt R/C Representation)
|GRIDZ|
S
11TERMANG
I
→
0
1
04954-022
0180
30
330
60
ADL5390 SDD22
90
270
ARG(
GRIDZ
), RADS(S
11
TERMANG
I
)
300
120
240
150
210
IMPEDANCE CIRCLE
SDD22 NOM WFR DUT 1
S22 NOM WFR
10MHz
3GHz
2.7GHz
10MHz
Figure 22. S22 of RF Output (Differential and Single-Ended through Balun)
139
138
137
136
135
134
133
132
–40 –25 –10 5 20 35 50 65 80
SUPPLY CURRENT (mA)
TEMPERATURE (°C)
04954-023
Vp = 5.25
Vp = 4.75
Vp = 5
Figure 23. Supply Current vs. Temperature
0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0
0
–5
–10
–15
–20
–25
–30
–35
–40
–45
–50
–55
–60
–65
–70
–75
DSOP VOLTAGE (V)
RF OUTPUT POWER (dBm)
04954-024
Figure 24. Power Shutdown Attenuation, RF = 380 MHz
TEK RUN ENVELOPE 12 AUG 04 16:48:01
CH1 500mV Ω
CH3 2.0V Ω D
S
M 10.0ns 5.0GS/s ET 200ps/pt
A CH3 760mV
1
04954-025
Figure 25. Power Shutdown Response Time, RF = 380 MHz
ADL5390
Rev. 0 | Page 11 of 24
GENERAL STRUCTURE
THEORY OF OPERATION
The simplified block diagram given in Figure 26 shows a
matched pair of variable gain channels whose outputs are
summed and presented to the final output. The RF/IF signals
propagate from the left to the right, while the baseband gain
controls are placed above and below. The proprietary linear-
responding variable attenuators offer excellent linearity, low
noise, and greater immunity from mismatches than other
commonly used methods.
Since the two independent RF/IF inputs can be combined in
arbitrary proportions, the overall function can be termed
“vector multiplication” as expressed by
VOUT = VIRF × (VIBB/VO) + VQRF × (VQBB/VO)
where:
VIRF and VQRF are the RF/IF input vectors.
VIBB and VQBB are the baseband input scalars.
VO is the built-in normalization factor, which is designed to be
0.285 V (1/3.5 V).
The overall voltage gain, in linear terms, of the I and Q channels
is proportional to its control voltage and scaled by the normali-
zation factor, i.e., a full-scale gain of 1.75 (5 dB) for VI (Q)BB of
500 mV. A full-scale voltage gain of 1.75 defines a gain setpoint
of 1.0.
Due to its versatile functional form and wide signal dynamic
range, the ADL5390 can form the core of a variety of useful
functions such as quadrature modulators, gain and phase ad-
justers, and multiplexers. At maximum gain on one channel, the
output 1 dB compression point and noise floor referenced to
50 Ω are 11 dBm and −148 dBm/Hz, respectively. The broad
frequency response of the RF/IF and gain control ports allows
the ADL5390 to be used in a variety of applications at different
frequencies. The bandwidth for the RF/IF signal path extends
from approximately 20 MHz to beyond 2.4 GHz, while the gain
controls signals allow for modulation rates greater than 200 MHz.
Matching between the two gain channels is ensured by careful
layout and design. Since they are monolithic and arranged
symmetrically on the die, thermal and process gradients are
minimized. Typical gain and phase mismatch at maximum gain
are <0.5 dB and <0.5°.
04954-026
LINEAR
ATTENUATOR
LINEAR
ATTENUATOR
I-V
V
QBB
Q CHANNEL
BASEBAND INPUT
V
IRF
,
I CHANNEL
SINGLE-ENDED
OR DIFFERENTIAL
V
IBB
OUTPUT
DISABLE
SINGLE-ENDED
OR DIFFERENTIAL
50Ω OUTPUT
V
QRF
,
Q CHANNEL
SINGLE-ENDED
OR DIFFERENTIAL
I CHANNEL
BASEBAND INPUT
V-I
V-I
Figure 26. Simplified Architecture of the ADL5390
NOISE AND DISTORTION
The signal path for a particular channel of the ADL5390 con-
sists basically of a preamplifier followed by a variable attenuator
and then an output driver. Each subblock contributes some level
of noise and distortion to the desired signal. As the channel gain
is varied, these relative contributions change. The overall effect
is a dependence of output noise floor and output distortion
levels on the gain setpoint.
For the ADL5390, the distortion is always determined by the
preamplifier. At the highest gain setpoint, the signal capacity, as
described by the 1 dB compression point (P1dB) and the third-
order intercept (OIP3), are at the highest levels. As the gain is
reduced, the P1dB and OIP3 are reduced in exact proportion.
At the higher gain setpoints, the output noise is dominated by
the preamplifier as well. At lower gains, the contribution from
the preamplifier is correspondingly reduced and eventually a
noise floor, set by the output driver, is reached. As Figure 27
illustrates, the overall dynamic range defined as a ratio of OIP3
to output noise floor remains constant for the higher gain
setpoints. At some gain level, the noise floor levels off and the
dynamic range degrades commensurate with the gain reduction.
175
170
165
160
155
150
145
140
DYNAMIC RANGE (dB
×
Hz)
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0
GAIN SETPOINT
04954-027
DYNAMIC RANGE = OIP3 – (OUTPUT NOISE
FLOOR (NO CARRIER))
Figure 27. Dynamic Range Variation with Gain Setpoint
ADL5390
Rev. 0 | Page 12 of 24
APPLICATIONS
USING THE ADL5390
The ADL5390 is designed to operate in a 50 Ω impedance system.
Figure 29 illustrates an example where the RF/IF inputs are
driven in a single-ended fashion, while the differential RF out-
put is converted to a single-ended output with a RF balun. The
baseband gain controls for the I and Q channels are typically
driven from differential DAC outputs. The power supplies,
VPRF and VPS2, should be bypassed appropriately with 0.1 µF
and 100 pF capacitors. Low inductance grounding of the CMOP
and CMRF common pins is essential to prevent unintentional
peaking of the gain. The exposed paddle on the underside of the
package should be soldered to a low thermal and electrical
impedance ground plane.
RF INPUT AND MATCHING
The RF/IF inputs present 250 Ω resistive terminations to
ground. In general, the input signals should be ac-coupled
through dc-blocking capacitors. The inputs may be driven dif-
ferentially or single-ended, in which case the unused inputs are
connected to common via the dc-blocking capacitors. The
ADL5390’s performance is not degraded by driving these inputs
single-ended. The input impedance can be reduced by placing
external shunt termination resistors to common on the source
side of the dc-blocking capacitors so that the quiescent dc-bias
level of the ADL5390 inputs is not affected, as shown in Figure
29. Capacitive reactance at the RF inputs can be compensated
for with series inductance. In fact, the customer evaluation
board has high impedance line traces between the shunt termi-
nation pads and the device input pins, which provides series in-
ductance and improves the return loss at 1.9 GHz to better than
−15 dB with the shunt termination removed, as shown in Figure
28.
S11 MATCH WITHOUT
66.5Ω TERMINATION
S11 MATCH WITH
TERMINATION
0
–5
–10
–15
–20
–25
–30
–35
dB
20.0
305.6
591.2
876.8
1162.4
1448.0
1733.6
2019.2
2400.0
115.2
210.4
400.8
496.0
686.4
781.6
972.0
1067.2
1257.6
1352.8
1543.2
1638.4
1828.8
1924.0
2114.4
2209.6
2304.8
FREQUENCY (MHz)
04954-028
Figure 28. ADL5390 Customer Evaluation Board RF Input Return Loss.
04954-029
C12
(SEE TEXT)
C11
(SEE TEXT)
C8
100pF
VP
C7
0.1µF
VP C4
0.1µF
B
A
SW1
R8
10kΩ
VP
IBBMIBBP
QBBP QBBM
L2
0Ω
C5
10nF
RFIN_Q R22
66.5Ω
L1
0Ω
C2
10nF
RFIN_I R2
66.5Ω
C1
10nF
C6
10nF
L4
120nH L3
120nH
C14
0.1µF
C10
100pF C9
0.1µF
VP
C17
10nF
C18
10nF
RFOP
15
34
T1
ETC1-1-13
(M/A-COM)
IBBP
IBBM
VPS2
DSOP
QBBP
QBBM
VPRF
CMRF
INMI
IPMQ
CMOP
CMOP
RFOM
RFOP
QFLP
QFLM
CMOP
CMOP
CMRF
VPRF
IFLP
IFLM
ADL5390
INPI
INPQ
C3
100pF
Figure 29. Basic Connections
ADL5390
Rev. 0 | Page 13 of 24
RF OUTPUT AND MATCHING
The RF/IF outputs of the ADL5390, RFOP and RFOM, are open
collectors of a transimpedance amplifier that need to be pulled
up to the positive supply, preferably with RF chokes, as shown
in Figure 30. The nominal output impedance looking into each
individual output pin is 25 Ω. Consequently, the differential
output impedance is 50 Ω.
04954-030
50Ω
DIFFERENTIAL
10pF 1:1 RF
OUTPUT
RFOM
RFOP
R
T
R
T
120nH
10pF
V
P
G
M
±
I
SIG
0.1µF
Figure 30. RF Output Interface to the ADL5390 Showing
Coupling Capacitors, Pull-Up RF Chokes, and Balun
Since the output dc levels are at the positive supply, ac-coupling
capacitors are usually needed between the ADL5390 outputs
and the next stage in the system.
A 1:1 RF broadband output balun, such as the ETC1-1-13 (M/A-
COM), converts the differential output of the ADL5390 into a
single-ended signal. Note that the loss and balance of the balun
directly impact the apparent output power, noise floor, and gain/
phase errors of the ADL5390. In critical applications, narrow-band
baluns with low loss and superior balance are recommended.
If the output is taken in a single-ended fashion directly into a 50 Ω
load through a coupling capacitor, there will be an impedance
mismatch. This can be resolved with a 1:2 balun to convert the
single-ended 25 Ω output impedance to 50 Ω. If loss of signal
swing is not critical, a 25 Ω back termination in series with the
output pin can also be used. The unused output pin must still be
pulled up to the positive supply. The user may load it through a
coupling capacitor with a dummy load to preserve balance. The
mismatched gain of the ADL5390 when the output is single-
ended varies slightly with dummy load value, as shown in Figure
31.
10
–10
–4
–2
0
2
4
6
8
10 100 1000 10000
04954-043
FREQUENCY (MHz)
GAIN (dB)
R
L2
= OPEN
R
L2
= 50Ω
R
L2
= SHORT
R
L
= 50Ω
Figure 31. Gain of the ADL5390 Using a Single-Ended Output with Different
Dummy Loads, RL2 on the Unused Output, Gain Setpoint = 1.0
The RF output signal can be disabled by raising the DSOP pin
to the positive supply. The output disable function provides
>40 dB attenuation of the input signal, even at full gain. The
interface to DSOP is high impedance and the output disable
and output enable response times are <100 ns. If the output
disable function is not needed, the DSOP should be tied to
ground.
DRIVING THE I-Q BASEBAND GAIN CONTROLS
The I and Q gain control inputs to the ADL5390 set the gain for
each channel. These inputs are differential and should normally
have a common-mode level of 0.5 V. However, when differen-
tially driven, the common mode can vary from 250 mV to
750 mV while still allowing full gain control. Each input pair
has a nominal input swing of ±0.5 V differential around the
common-mode level. The maximum gain is achieved if the
differential voltage is equal to +500 mV or −500 mV. So with a
common-mode level of 500 mV, IBBP and IBBM will each
swing between 250 mV and 750 mV.
The I and Q gain control inputs can also be driven with a single-
ended signal. In this case, one side of each input should be tied
to a low noise 0.5 V voltage source (a 0.1 µF decoupling capaci-
tor located close to the pin is recommended), while the other
input swings from 0 V to 1 V. Low speed, single-ended drive can
easily be achieved using 12-bit voltage output DACs such as
AD8303 (serial SPI® interface) or AD8582 (parallel interface)
DACs. A reference voltage should also be supplied. Differential
drive generally offers superior even-order distortion and lower
noise than single-ended drive.
The bandwidth of the baseband controls exceeds 200 MHz even
at full-scale baseband drive. This allows for very fast gain modu-
lation of the RF input signal. In cases where lower modulation
bandwidths are acceptable or desired, external filter capacitors
can be connected across Pins IFLP to IFLM and Pins QFLP to
QFLM to reduce the ingress of baseband noise and spurious
signal into the control path.
ADL5390
Rev. 0 | Page 14 of 24
The 3 dB bandwidth is set by choosing CFLT according to the
following equation:
pF0.5
nF10kHz45
dB3 +
×
≈
external
C
f
This equation has been verified for values of CFLT from 10 pF to
0.1 µF (bandwidth settings of approximately 4.5 kHz to 43 MHz).
INTERFACING TO HIGH SPEED DACs
The AD977x family of dual DACs is well suited to driving the I
and Q gain controls of the ADL5390 with fast modulating sig-
nals. While these inputs can in general be driven by any DAC,
the differential outputs and bias level of the ADI TxDAC® fam-
ily allows for a direct connection between DAC and modulator.
The AD977x family of dual DACs has differential current out-
puts. The full-scale current is user programmable and is usually
set to 20 mA, that is each output swings from 0 mA to 20 mA.
The basic interface between the AD9777 DAC outputs and the
ADL5390 I and Q gain control inputs is shown in Figure 32.
Resistors R1 and R2 (R1 = R2) set the dc bias level according to
the following equation:
Bias Level = Average Output Current × R1
For example, if the full-scale current from each output is 20 mA,
each output will have an average current of 10 mA. Therefore,
to set the bias level to the recommended 0.5 V, R1 and R2
should be set to 50 Ω each. R1 and R2 should always be equal.
If R3 is omitted, this will result in an available swing from the
DAC of 2 V p-p differential, which is twice the maximum voltage
range required by the ADL5390. DAC resolution can be maxi-
mized by adding R3, which scales down this voltage according
to the following equation:
=SwingScaleFull
()()
⎥
⎥
⎦
⎤
⎢
⎢
⎣
⎡
+
−×+× R3R2
R2
R3R2R1I MAX 1||2
OPTIONAL
LOW-PASS
FILTER
04954-032
R1
R2 R3
I
OUTB2
I
OUTA2
QBBM
QBBP
I
OUTB1
I
OUTA1
IBBM
IBBP
AD9777 ADL5390
OPTIONAL
LOW-PASS
FILTER
R1
R2 R3
Figure 32. Basic AD9777-to-ADL5390 Interface
R3 (Ω)
04954-033
13050 55 60 65 70 75 80 85 90 100 105 115120110 12595
DIFFERENTIAL PEAK-PEAK SWING (R3) (V p-p)
1.15
1.08
1.10
1.13
1.00
1.02
1.05
0.95
0.97
0.88
0.90
0.92
0.77
0.80
0.82
0.85
0.70
0.75
0.72
Figure 33. Peak-Peak DAC Output Swing vs.
Swing Scaling Resistor R3 (R1 = R2 = 50 Ω)
Figure 33 shows the relationship between the value of R3 and
the peak baseband voltage with R1 and R2 equal to 50 Ω. As
shown in Figure 33, a value of 100 Ω for R3 will provide a
peak-peak swing of 1 V p-p differential into the ADL5390’s
I and Q inputs.
When using a DAC, low-pass image reject filters are typically
used to eliminate the Nyquist images produced by the DAC.
They also provide the added benefit of eliminating broadband
noise that might feed into the modulator from the DAC.
ADL5390
Rev. 0 | Page 15 of 24
GENERALIZED MODULATOR
The ADL5390 can be configured as a traditional IQ quadrature
modulator or as a linear vector modulator by applying signals
that are in quadrature to the RF/IF input channels. Since the
quadrature generation is performed externally, its accuracy and
bandwidth are determined by the user. The user-defined band-
width is attractive for multioctave or lower IF applications
where on-chip, high accuracy quadrature generation is tradi-
tionally difficult or impractical. The gain control pins (IBBP/M
and QBBP/M) become the in-phase (I) and quadrature (Q) base-
band inputs for the quadrature modulator and the gain/phase
control for the vector modulator. The wide modulation band-
widths of the gain control interface allow for high fidelity base-
band signals to be generated for the quadrature modulator and
for high speed gain and phase adjustments to be generated for
the vector modulator.
RF/IF signals can be introduce to the ADL5390 in quadrature
by using a two-way 90o power splitter such as the Mini-Circuits
QCN-12. Each output of an ideal 90o power splitter is 3 dB
smaller than the input and has a 90o phase difference from the
other output. In reality, the 90o power splitter will have its own
insertion loss, which can be different for each output, causing a
magnitude imbalance. Furthermore, quadrature output will not
be maintained over a large frequency range, introducing a phase
imbalance. The type of 90o power splitter that should be used
for a particular application will be determined by the frequency,
bandwidth, and accuracy needed. In some applications minor
magnitude and phase imbalances can be adjusted for in the
I/Q gain control inputs.
VECTOR MODULATOR
04954-034
|A|
θ
A
+0.5–0.5
+0.5
–0.5
Vi
Vq
MIN GAIN < –30dB
MAX GAIN = 5dB
Figure 34. Vector Gain Representation
The ADL5390 can be used as a vector modulator by driving the
RF I and Q inputs single-ended through a 90o power splitter. By
controlling the relative amounts of I and Q components that are
summed, continuous magnitude and phase control of the gain
is possible. Consider the vector gain representation of the
ADL5390 expressed in polar form in Figure 34. The attenuation
factors for the RF I and Q signal components are represented on
the x-axis and y-axis, respectively, by the baseband gain control
inputs VIBB and VQBB. The resultant of their vector sum represents
the vector gain, which can also be expressed as a magnitude and
phase. By applying different combinations of baseband inputs,
any vector gain within the unit circle can be programmed. The
magnitude and phase (with respect to 90o) accuracy of the 90o
power splitter will directly affect this representation and could
be seen as an offset and skew of the circle.
A change in sign of VIBB or VQBB can be viewed as a change in
sign of the gain or as a 180° phase change. The outermost circle
represents the maximum gain magnitude. The circle origin
implies, in theory, a gain of 0. In practice, circuit mismatches
and unavoidable signal feedthrough limit the minimum gain to
approximately −30 dB. The phase angle between the resultant
gain vector and the positive x-axis is defined as the phase shift.
Note that there is a nominal, systematic insertion phase through
the ADL5390 to which the phase shift is added. In the following
discussions, the systematic insertion phase is normalized to 0°.
The correspondence between the desired gain and phase and
the Cartesian inputs VIBB and VQBB is given by simple trigono-
metric identities
(
)
()
[
]
22 // OQBBO
IBB VVVVGain +=
(
)
IBB
QBB VVarctanPhase /
=
where:
VO is the baseband scaling constant (285 mV).
VIBB and VQBB are the differential I and Q baseband voltages
centered around 500 mV, respectively (VIBB = VIBBP − VIBBM;
VQBB = VQBBP − VQBBM).
Note that when evaluating the arctangent function, the proper
phase quadrant must be selected. For example, if the principal
value of the arctangent (known as arctangent(x)) is used, quad-
rants 2 and 3 would be interpreted mistakenly as quadrants 4
and 1, respectively. In general, both VIBB and VQBB are needed in
concert to modulate the gain and the phase.
Pure amplitude modulation is represented by radial movement
of the gain vector tip at a fixed angle, while pure phase modula-
tion is represented by rotation of the tip around the circle at a
fixed radius. Unlike traditional I-Q modulators, the ADL5390 is
designed to have a linear RF signal path from input to output.
Traditional I-Q modulators provide a limited LO carrier path
through which any amplitude information is removed.
VECTOR MODULATOR EXAMPLE—CDMA2000
The ADL5390 can be used as a vector modulator by driving the
RF I and Q inputs (INPI and INPQ) single-ended through a 90o
power splitter and controlling the magnitude and phase using
the gain control inputs. To demonstrate operation as a vector
modulator, an 880 MHz single-carrier CDMA2000 test model
signal (forward pilot, sync, paging, and six traffic as per
ADL5390
Rev. 0 | Page 16 of 24
3GPP2 C.S0010-B, Table 6.5.2.1) was applied to the ADL5390.
A cavity-tuned filter was used to reduce noise from the signal
source being applied to the device. The 4.6 MHz pass band of
this filter is apparent in the subsequent spectral plots.
Figure 35 shows the output signal spectrum for a programmed
gain and phase of 5 dB and 45 o. POUT is equal to 0 dBm and
VIBB = VQBB = 0.353 V (centered around 500 mV), i.e., VIBBP −
VIBBM = VQBBP − VQBBM = 0.353 V. Adjacent channel power is
measured in 30 kHz resolution bandwidth at 750 kHz and
1.98 MHz carrier offset. Noise floor is measured at ±4 MHz
carrier offset in a 1 MHz resolution bandwidth.
1
CU2
CU2
CU1
C0
C0
Cl1
Cl1
Cl2
Cl2
CU1
04954-035
A
CENTER 880MHz 500kHz/ SPAN 5MHz
REF LVL
5dBm
0.7dB OFFSET
1AVG
MARKER 1 [T1]
–14.38dBm
880.00755511MHz
RBW 30kHz
VWB 300kHz
SWT 2s
RF ATT 20dB
MIXER –10dBm
UNIT dB
1 [T1] –14.38dBm
880.00755511MHz
CH PWR
0
–70
–80
–90
–
100
–60
–50
–40
–30
–20
–10
1 [T1] –14.38dBm
880.00755511MHz
CH PWR 0.13dBm
ACP UP –62.00dB
ACP LOW –61.98dB
ALT1 UP –87.02dB
ALT1 LOW –87.04dB 1RM
EXT
Figure 35. Output Spectrum, Single-Carrier CDMA2000 Test Model at −5 dBm,
VI = VQ = 0.353 V, ACP Measured at 750 kHz and 1.98 MHz Carrier Offset,
Input Signal–Filtered Using a Cavity-Tuned Filter (Pass Band = 4.6 MHz)
Holding the I and Q gain control voltages steady at 0.353 V,
input power was swept. Figure 36 shows the resulting output
power, noise floor, and adjacent channel power ratio. The noise
floor is presented as noise in a 1 MHz bandwidth as defined by
the 3GPP2 specification.
–30
–100
–90
–80
–70
–60
–50
–40
–30
–100
–90
–80
–70
–60
–50
–40
–30 –25 –20 –15 –10 –5 0 5
04954-044
OUTPUT POWER (dBm)
ACP (dBc)
NOISE (dBm @ 4MHz Carrier Offset)
ACP: 750kHz OFFSET, 30kHz RBW
ACP: 1.98MHz OFFSET, 30kHz RBW
NOISE: 4MHz OFFSET, 1MHz RBW
Figure 36. Noise and ACP vs. Output Power, Single-Carrier CDMA2000 Test
Model, VI = VQ = 0.353, ACP Measured in 30 kHz RBW at ±750 kHz and
±1.98 MHz Carrier Offset, Noise Measured at ±4 MHz Carrier Offset
ACP is still in compliance with the standard (<−45 dBc @
750 kHz and <−60 dBc @ 1.98 MHz) even with output powers
greater than +3 dBm. At low output power levels, ACP at
1.98 MHz carrier offset degrades as the noise floor of the
ADL5390 becomes the dominant contributor to measured ACP.
Measured noise at 4 MHz carrier offset begins to increase
sharply above 2 dBm output power. This increase is not due to
noise, but results from increased carrier-induced distortion. As
output power drops below 2 dBm, the noise floor drops towards
−90 dBm.
With a fixed input power of 2.16 dBm, the output power was
again swept by changing VIBB and VQBB from 0 V to 500 mV.
The resulting output power, ACP, and noise floor are shown in
Figure 37.
5
–44
–37
–30
–23
–16
–9
–2
–30
–100
–90
–80
–70
–60
–50
–40
0 0.1 0.2 0.3 0.4 0.5
04954-045
V
I(Q)BB
RF OUTPUT POWER (dBm)
NOISE (dBm @ 4MHz Carrier Offset)
ACP (dBc)
V
OUT
vs. V
IBB
/V
QBB
ACP: 750kHz OFFSET, 30kHz RBW
NOISE: 4MHz OFFSET, 1MHz RBW
ACP: 1.98MHz OFFSET, 30kHz RBW
Figure 37. Output Power, Noise, and ACP vs. I and Q Control Voltages,
CDMA2000 Test Model, VI = VQ, ACP Measured in 30 kHz RBW at ±750 MHz
and ±1.98 MHz Carrier Offset, Noise Measured at ±4 MHz Carrier Offset
In contrast to Figure 36, Figure 37 shows that for a fixed input
power, ACP remains fairly constant as gain and phase are
changed (this is not true for very high RF input powers) until
the noise floor of the ADL5390 becomes the dominant con-
tributor to the measured ACP.
ADL5390
Rev. 0 | Page 17 of 24
QUADRATURE MODULATOR
The ADL5390 can be used as a quadrature modulator by driving
the RF I and Q inputs (INPI and INPQ) single-ended through a
90o phase splitter to serve as the LO input. I/Q modulation is
applied to the baseband I and Q gain control inputs (IBBP/IBBM
and QBBP/QBBM). A simplified schematic is shown in Figure
38.
04954-038
90° PHASE
SPLITTER
SUM
PORT
TERM
PORT
PORT 1
PORT 2
QCN-12
QBBP
QBBM
IBBP
IBBM
RFOM
RFOP
INPI
Q DATA
I DATA
INPQ
15
34
ROFP
LO IN
ADL5390
10nF
10nF
10nF
10nF
66.5Ω
66.5Ω
50Ω
ETC1-1-13
(M/A-COM)
Figure 38. Quadrature Modulator Application
Single sideband performance of a quadrature modulator is
determined by the magnitude and phase balance (compared to
a 90o offset) at the summation point of the I and Q signals.
Because the ADL5390 has matched amplifiers and mixers in
the I and Q channel, most of the single sideband performance
will be determined by the external 90o phase splitter. Good
single sideband performance can be achieved by choosing a
well-balanced 90o phase splitter. However, phase and magnitude
differences in the 90o phase splitter can be corrected by adjusting
the magnitude and phase of the I and Q data. Figure 39 shows
the performance of the ADL5390 used in conjunction with Mini-
Circuits QCN-12 90o power splitter. Figure 40 shows the single
sideband improvement as the I and Q data is adjusted in magnitude
and phase to achieve better single sideband performance.
For maximum dynamic range, the ADL5390 should be driven
as close to the output 1 dB compression point as possible. The
output power of the ADL5390 increases linearly with the RF
(LO) input power and baseband gain control input voltage until
the ADL5390 reaches compression. At the 1 dB compression
point, the lower sideband starts to increase. Figure 41 demon-
strates the output spectrum of a 3-carrier CDMA2000 signal
applied to the I/Q baseband gain control inputs. As the RF (LO)
power is increased, the relative amount of noise is reduced until
the ADL5390 goes into compression. At this point, the relative
noise increases, as shown in Figure 42.
Analog Devices has several quadrature/vector modulators that
have highly accurate integrated 90o phase splitters—AD8340,
AD8341, AD8345, AD8346, AD8349—that cover a variety of
frequency bands.
04954-039
-
9
0
-80
1
2
34
A
CENTER 900MHz 700kHz/ SPAN 7MHz
REF 7dBm ATT 35dB
*
RBW 3kHz
VWB 10kHz
SWT 780ms
1 AP
CLRWR
0
–70
–60
–50
–40
–30
–20
–10
DESIRED SIDEBAND
–16.20dBm
900.998397436MHz
UNDESIRED SIDEBAND
–23.27dB
–1.996794872MHz
THIRD BASEBAND HARMONIC
–37.38dB
–4.004807692MHz
LO FEEDTHROUGH
–41.27dB
–998.397435897kHz
Figure 39. SSB Quadrature Modulator Result Using External 90° Phase Splitter,
RF PIN = −15 dBm, VIBB = VQBB = 0.5 V
(With Reference to a Common-Mode Voltage of 0.5 V)
04954-046
A
CENTER 900MHz 700kHz/ SPAN 7MHz
REF 7dBm ATT 35dB
*
RBW 3kHz
VWB 10kHz
SWT 780ms
1 AP
CLRWR
0
–70
–80
–90
–60
–50
–40
–30
–20
–10
1
2
3
4
DESIRED SIDEBAND
–16.78dBm
900.998397436MHz
UNDESIRED SIDEBAND
–51.81dB
–1.996794872MHz
THIRD BASEBAND HARMONIC
–38.45dB
–4.004807692MHz
LO FEEDTHROUGH
–41.49dB
–998.397435897kHz
Figure 40. SSB Modulator Applications with Gain and Phase Errors Corrected,
RF Pin = −15 dBm, VIBB = VQBB = 0.5 V (With Reference to a Common-Mode
Voltage of 0.5 V), I/Q Phase Offset by 3o, and Magnitude Offset by 0.5 V
ADL5390
Rev. 0 | Page 18 of 24
04954-037
A
–30
–40
–50
–60
–70
–80
–90
–100
–110
–120
REF –26.6dBm
*
ATT 5dB
*
RBW 10kHz
*
VBW 300kHz
*
SWT 4s
CENTER 880MHz 1.29MHz/ SPAN 12.9MHz
NOR
1 RM
AVG
*
POS –26.623dBm
STANDARD: CDMA IS95C CLASS 0 FWD
Tx CHANNELS
CH1 (REF)
CH2
CH3
TOTAL
–22.93dBm
–22.83dBm
–22.95dBm
–18.13dBm
ADJACENT CHANNEL
LOWER –59.25dB
UPPER –59.43dB
ALTERNATE CHANNEL
LOWER –62.79dB
UPPER –63.11dB
2ND ALTERNATE CHANNEL
LOWER –60.71dB
UPPER –60.32dB
RF MULTIPLEXER
The ADL5390 may also be used as an RF multiplexer. In this
application, two RF signals are applied to the INPI and INPQ
inputs, and the baseband voltages control which of the two RF
signals appears at the output. Figure 43 illustrates this applica-
tion and shows that with VIBB = 0.5 and VQBB = 0.0 (with refer-
ence to a common-mode voltage of 0.5 V). The INPI signal is
presented to the output. Then, when VIBB transitions to VIBB =
0.0 and VQBB remains equal to zero, there is no RF output.
Lastly, as VQBB transitions to 0.5 V, the INPQ signal appears at
the output. With VIBB = 0.0 and VQBB = 0.0, the isolation to the
output is typically >40 dB at 380 MHz.
04954-047
RF OUTPUT
V
IBB
V
QBB
TEK STOPPED 200 Acqs 24 AUG 04 11:08:40
CH1 200mV ΩCH2 500mV Ω D
S
CH3 500mV Ω D
S
M 40.0ns 1.25GS/s 800ps/pt
A CH2 780mV
1
2
CH2
CH3
CH1
3
Figure 41. ADL5390 as a Quadrature Modulator with the Use of an External
90° Phase Splitter, RF/LO Power = −1 dBm and Gain Control Inputs Driven
Differentially with 0.353 VP-P, 3-Carrier CDMA2000 I/Q Data
–78
–77
–76
–75
–74
–73
–72
–71
–70
–69
–25 –20 –15 –10 –5 0
OUTPUT POWER (dBm)
NOISE (dBC) (1MHz RBW)
NOISE (dBc) - 8MHz OFFSET - 1MHz RBW
04954-031
Figure 43. ADL5390 in RF Multiplexer Application
Figure 42. Noise vs. Output Power
ADL5390
Rev. 0 | Page 19 of 24
EVALUATION BOARD
The evaluation board circuit schematic for the ADL5390 is
shown in Figure 44.
The evaluation board is configured to be driven from a
single-ended 50 Ω source. Although the input of the ADL5390
is differential, it may be driven single-ended with no loss of
performance.
The low-pass corner frequency of the baseband I and Q chan-
nels can be reduced by installing capacitors in the C11 and C12
positions. The low-pass corner frequency for either channel is
approximated by
pF0.5
nF10kHz45
dB3 +
×
≈
external
C
f
On this evaluation board, the I and Q baseband circuits are
identical to each other, so the following description applies to
each. The connections and circuit configuration for the I/Q
baseband inputs are described in Table 4.
The baseband input of the ADL5390 requires a differential volt-
age drive. The evaluation board is set up to allow such a drive
by connecting the differential voltage source to QBBP and
QBBM. The common-mode voltage should be maintained at
approximately 0.5 V. For this configuration, Jumpers W1 to W4
should be removed.
The baseband input of the evaluation board may also be driven
with a single-ended voltage. In this case, a bias level is provided
to the unused input from Potentiometer R10 by installing either
W1 or W2.
Setting SW1 in Position B disables the ADL5390 output amplifier.
With SW1 set to Position A, the output amplifier is enabled. With
SW1 set to Position A, an external voltage signal, such as a pulse,
can be applied to the DSOP SMA connector to exercise the
output amplifier enable/disable function.
ADL5390
Rev. 0 | Page 20 of 24
Table 4. Evaluation Board Configuration Options
Component Function Default Conditions
R7, R9, R11, R14,
R15, R19, R20,
R21, C15, C19,
W3, W4
I Channel Baseband Interface. Resistors R7 and R9 may be installed to
accommodate a baseband source that requires a specific terminating
impedance. Capacitors C15 and C19 are bypass capacitors. For single-ended
baseband drive, the Potentiometer R11 can be used to provide a bias level to
the unused input (install either W3 or W4).
R7, R9 = not installed
R11 = potentiometer, 2 kΩ, 10 turn
(Bourns)
R14 = 4 kΩ (size 0603)
R15 = 44 kΩ (size 0603)
R19, R20, R21 = 0 Ω (size 0603)
C15, C19 = 0.1 µF
(Size 0603)
W3 = jumper (installed)
W4 = jumper (open)
R1, R3, R10, R12,
R13, R16, R17,
R18, C16, C20,
W1, W2
Q Channel Baseband Interface. See the I Channel Baseband Interface section. R1, R3 = not installed
R10 = potentiometer, 2 kΩ, 10 turn
(Bourns)
R12 = 4 kΩ (size 0603)
R13 = 44 kΩ (size 0603)
R16, R17, R18 = 0 Ω
(size 0603)
C16, C20 = 0.1 µF (size 0603)
W1 = jumper (installed)
W2 = jumper (open)
C11, C12 Baseband Low-Pass Filtering. By adding capacitor C11 between QFLP and
QFLM, and capacitor C12 between IFLP and IFLM, the 3 dB low-pass corner
frequency of the baseband interface can be reduced from 230 MHz (nominal)
as given by the equation in the Evaluation Board section.
C11, C12 = not installed
T1, C17, C18, L3,
L4
Output Interface. The 1:1 balun transformer, T1, converts the 50 Ω differential
output to 50 Ω single-ended. C17 and C18 are dc blocks. L3 and L4 provide dc
bias for the output.
C17, C18 = 10 nF (size 0603)
T1 = ETC1-1-13 (M/A-COM)
L3, L4 = 120 nH (size 0603)
C2, C1, R2
C5, C6, R22
I and Q Channel RF Input Interface. The single-ended impedance to the
ADL5390 RF inputs is 200 Ω. Shunt terminations R2 and R22 of 66.5 Ω bring the
impedances to 50 Ω. C2 and C5 are dc blocks. C1 and C6 are used to ac-couple
the unused side of the differential inputs to common.
C2 = C1 = 10 nF (size 0603)
R2 = 66.5 Ω 10 (size 0603)
C5 = C6 = 10 nF (size 0603)
R22 = 66.5 Ω 10 (size 0603)
R4, R6, R5, C4, C7
C9, C3, C8, C10
Power Supply Decoupling. R4, R6, R5 = 0 Ω (size 0603)
C4, C7 C9 = 0.1uF (size 0603)
C3, C8, C10 = 100 pF (size 0603)
R8, SW1 Output Disable Interface. The output stage of the ADL5390 is disabled by
applying a high voltage to the DSOP pin by moving SW1 to Position B. The
output stage is enabled moving SW1 to Position A. The output disable
function can also be exercised by applying an external high or low voltage to
the DSOP SMA connector with SW1 in Position A.
R8 = 10 kΩ (size 0603)
SW1 = SPDT (Position A, output
enabled)
ADL5390
Rev. 0 | Page 21 of 24
04954-040
C12
(OPEN)
C11
(OPEN)
C8
100pF
VP
C7
0.1µFR5
0Ω
VP
R4
0Ω
C3
100pF
C4
0.1µF
B
A
SW1
R8
10kΩ
VP
TEST POINT GND
TEST POINT
DSOP
IBBMIBBP
VP
R14
4kΩR11
2kΩR15
44kΩ
C15
0.1µF
W4
R21
0Ω
R19
0Ω
W3 R20
0Ω
R9
(OPEN) R7
(OPEN)
C19
0.1µF
QBBP QBBM
VP
R12
4kΩR10
2kΩR13
44kΩ
C16
0.1µF
W2
R17
0ΩR16
0Ω
W1
R18
0Ω
R1
(OPEN) R3
(OPEN)
C20
0.1µF
L2
0Ω
C5
10nF
RFIN_Q R22
66.5Ω
L1
0Ω
C2
10nF
RFIN_I R2
66.5Ω
C1
10nF
C6
10nF
L4
120nH L3
120nH
C14
0.1µF
C10
100pF C9
0.1µF
VP
C17
10nF
C18
10nF
RFOP
R6
0Ω
15
34
T1
ETC1-1-13
(M/A-COM)
IBBP
IBBM
VPS2
DSOP
QBBP
QBBM
VPRF
CMRF
INMI
IPMQ
CMOP
CMOP
RFOM
RFOP
QFLP
QFLM
CMOP
CMOP
CMRF
VPRF
IFLP
IFLM
ADL5390
INPI
INPQ
Figure 44. Evaluation Board Schematic
ADL5390
Rev. 0 | Page 22 of 24
04954-041
Figure 45. Component Side Layout
04954-042
Figure 46. Component Side Silkscreen
ADL5390
Rev. 0 | Page 23 of 24
OUTLINE DIMENSIONS
1
24
6
7
13
19
18
12
2.45
2.30 SQ*
2.15
0.60 MAX
0.50
0.40
0.30
0.30
0.23
0.18
2.50 REF
0.50
BSC
12° MAX 0.80 MAX
0.65 TYP 0.05 MAX
0.02 NOM
1.00
0.85
0.80
SEATING
PLANE
PIN 1
INDICATOR TOP
VIEW 3.75
BSC SQ
4.00
BSC SQ PIN 1
INDICATOR
0.60 MAX
COPLANARITY
0.08
0.20 REF
0.23 MIN
EXPOSED
PA D
(BOTTOMVIEW)
*COMPLIANT TO JEDEC STANDARDS MO-220-VGGD-2
EXCEPT FOR EXPOSED PAD DIMENSION
Figure 47. 24-Lead Lead Frame Chip Scale Package [LFCSP]
4 × 4 mm Body
(CP-24-2)
Dimensions shown in millimeters
ORDERING GUIDE
Models Temperature Range Package Description Package Option Order Multiple
ADL5390ACPZ-WP1, 2 −40°C to +85°C 24-Lead Lead Frame Chip Scale Package (LFCSP) CP-24-2 64
ADL5390ACPZ-REEL71 −40°C to +85°C 24-Lead Lead Frame Chip Scale Package (LFCSP) CP-24-2 1,500
ADL5390-EVAL Evaluation Board 1
1 Z = Pb-free part.
2 WP = waffle pack.
ADL5390
Rev. 0 | Page 24 of 24
NOTES
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and regis-
tered trademarks are the property of their respective owners.
D04954–0–10/04(0)
