LT8315 560VIN Micropower No-Opto Isolated Flyback Converter with 630V/300mA Switch DESCRIPTION FEATURES Wide Input Voltage Range: 18V to 560V nn 630V/300mA Integrated Power Switch nn No Opto-Isolator Required for Regulation nn Quasi-Resonant Boundary Mode Operation nn Constant-Current and Constant-Voltage Regulation nn Low Ripple Light Load Burst Mode(R) Operation nn Low Quiescent Current: 70A nn Programmable Current Limit and Soft-Start nn TSSOP Package with High Voltage Spacing The LT(R)8315 is a high voltage flyback converter with integrated 630V/300mA switch. No opto-isolator is needed for regulation. The device samples the output voltage from the isolated flyback waveform appearing across a third winding on the transformer. Quasi-resonant boundary mode operation improves load regulation, reduces transformer size and maintains high efficiency. nn APPLICATIONS At start-up, the LT8315 charges its INTVCC capacitor via a current source attached to the DRAIN pin. During normal operation, the current source turns off and the device draws its power from a third winding on the transformer. Isolated Telecom, Automotive, Industrial, Medical Power Supplies nn Isolated Off Line Housekeeping Power Supplies nn Electric Vehicles and Battery Stacks The LT8315 operates from a wide range of input supply voltages and can deliver up to 15W of power. It is available in a thermally enhanced 20-pin TSSOP package with four pins removed for high voltage spacing. nn All registered trademarks and trademarks are the property of their respective owners. TYPICAL APPLICATION 20VIN to 450VIN Isolated 12VOUT Supply Efficiency VIN 20V TO 450V 90 80 5:1:2 600 20k 93.1k 47pF 10F BIAS FB 61.9k INTVCC SMODE DRAIN VC 100k 470pF 22nF 5.11k TC LT8315 10F 160H DCM EN/UVLO VOUT+ 12V 640H 4mH 200F VOUT- 5mA to 220mA (VIN = 50V) 5mA to 320mA (VIN = 100V) 5mA to 380mA (VIN = 150V) 5mA to 440mA (VIN > 250V) SOURCE IREG/SS GND EFFICIENCY (%) 0.44F 70 60 VIN = 50V VIN = 100V VIN = 150V VIN = 250V VIN = 350V VIN = 450V 50 40 30 0 100 200 300 400 LOAD CURRENT (mA) 500 8315 TA01b 330m 121k 8315 TA01a 8315fa For more information www.linear.com/LT8315 1 LT8315 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) DRAIN......................................................................630V BIAS, EN/UVLO..........................................................40V INTVCC.......................................................................15V SMODE................................................................ INTVCC SOURCE, TC, FB, VC, IREG/SS.....................................4V DCM.................................................................... 100mA Operating Junction Temperature (Note 2) LT8315E, LT8315I............................... -40C to 125C LT8315H............................................. -40C to 150C Storage Temperature Range................... -65C to 150C Lead Temperature (Soldering, 10 sec)................... 300 C ORDER INFORMATION TOP VIEW DRAIN 1 20 GND DRAIN 2 19 NC DRAIN 3 18 SOURCE 17 EN/UVLO 21 GND 16 SMODE 15 GND 14 IREG/SS INTVCC 8 13 VC BIAS 9 12 FB DCM 10 11 TC FE PACKAGE 20(16)-LEAD PLASTIC TSSOP JA = 38C/W, JC = 10C/W EXPOSED PAD (PIN 21) IS GND, MUST BE SOLDERED TO PCB http://www.linear.com/product/LT8315#orderinfo LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT8315EFE#PBF LT8315EFE#TRPBF LT8315FE 20-Lead Plastic TSSOP -40C to 125C LT8315IFE#PBF LT8315IFE#TRPBF LT8315FE 20-Lead Plastic TSSOP -40C to 125C LT8315HFE#PBF LT8315HFE#TRPBF LT8315FE 20-Lead Plastic TSSOP -40C to 150C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container Consult LTC Marketing for information on nonstandard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through designated sales channels with #TRMPBF suffix. 8315fa 2 For more information www.linear.com/LT8315 LT8315 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. BIAS = 40V, VEN/UVLO = 40V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN BIAS Chip Bias Voltage Supply Range After Startup IQ BIAS Quiescent Current Burst Mode Operation Active ISHDN DRAIN Shutdown Current VEN/UVLO < 0.3V, BIAS = Floating l VDRAIN(MIN) Minimum Drain Voltage for Startup BIAS = Floating l ISTARTUP Startup Current through Depletion FET VDRAIN = 18V, BIAS = Floating l VUVLO EN/UVLO Threshold EN/UVLO Hysteresis INTVCC UVLO Rising Threshold 40 V 70 470 150 700 A 8 15 A 9.5 UNITS 18 V 300 VEN/UVLO Falling VEN/UVLO Rising 1.18 30 1.22 65 1.26 120 V mV Startup Current through Depletion FET 11.1 12 13.1 V INTVCC UVLO Falling Threshold FB Regulation Voltage GM Voltage Error Amplifier Transconductance VFB = 1.22V 20mV VTC TC Voltage TC Voltage Temperature Coefficient TA = 25C ITC TC Sinking/Sourcing Current IIREG/SS IREG/SS Current A 7.7 8.2 8.7 V l 1.19 1.22 1.25 V l 75 100 125 S 1.16 1.22 +4.1 1.28 V mV/C 9.9 9.5 10 10.1 10.5 A -140 -65 -170 -85 -200 -105 A A 7 10 500 300 800 100 Current Out-of-Pin l Flyback Collapse Detection Threshold Resonant Valley Detection Threshold MAX 130 VREG IDCM TYP IDCM Rising IDCM Falling A RSW Power MOSFET Resistance ISW(MAX) Maximum Switch Current VSOURCE(MIN) Minimum Current Voltage Threshold 15 20 25 mV VSOURCE(MAX) Maximum Current Voltage Threshold 90 100 110 mV VSOURCE(ILIM) Over-Current Voltage Threshold 250ns Blanking Period; Restarts Chip 110 120 130 mV FSW(MIN) Minimum Switching Frequency Burst Mode Standby Mode 3 187 3.5 220 4 250 kHz Hz FSW(MAX) Maximum Switching Frequency 138 140 142 kHz l Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT8315E is guaranteed to meet performance specifications from 0C to 125C junction temperature. Specifications over the -40C to 125C operating junction temperature range are assured by design mA mA characterization and correlation with statistical process controls. The LT8315I is guaranteed over the full -40C to 125C operating junction temperature range. The LT8315H is guaranteed over the full -40C to 150C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125C. 8315fa For more information www.linear.com/LT8315 3 LT8315 TYPICAL PERFORMANCE CHARACTERISTICS FRONT PAGE APPLICATION 12.3 12.3 12.2 12.2 OUTPUT VOLTAGE (V) OUTPUT VOLTAGE (V) Output Voltage vs Temperature 12.4 12.1 12.0 11.9 VIN = 50V VIN = 100V VIN = 150V VIN = 250V VIN = 350V VIN = 450V 11.8 11.7 11.6 0 100 200 300 400 LOAD CURRENT (mA) 12.0 11.9 11.8 11.6 -50 -25 FREQUENCY (kHz) 40 VSOURCE 100mV/DIV VOUT AC COUPLED 50mV/DIV VOUT AC COUPLED 50mV/DIV FRONT PAGE APPLICATION 200 300 400 LOAD CURRENT (mA) 0 75 150 225 300 LOAD CURRENT (mA) 500 375 450 Discontinuous Mode Waveforms VSOURCE 100mV/DIV 10s/DIV 100 VIN = 100V VIN = 150V VIN = 250V VIN = 350V VIN = 450V 8315 G03 VDRAIN 200V/DIV 20 0 0 VDRAIN 200V/DIV 60 0 25 50 75 100 125 150 TEMPERATURE (C) Boundary Mode Waveforms VIN = 50V VIN = 100V VIN = 150V VIN = 250V VIN = 350V VIN = 450V 80 6 3 IOUT = 100mA IOUT = 200mA IOUT = 440mA 0 9 8315 G02 Switching Frequency 100 FRONT PAGE APPLICATION RIREG/SS = 66.5k 12 12.1 8315 G01 120 FRONT PAGE APPLICATION VIN = 350V 11.7 500 CV/CC Operation 15 OUTPUT VOLTAGE (V) Load and Line Regulation 12.4 TA = 25C, unless otherwise noted. 8315 G05 10s/DIV FRONT PAGE APPLICATION VIN = 350V, IOUT = 440mA FRONT PAGE APPLICATION VIN = 350V, IOUT = 50mA Load Transient Response Startup Waveforms 8315 G06 8315 G04 Burst Mode Waveforms VIN 350V/DIV VOUT 10V/DIV IOUT 500mA/DIV VDRAIN 200V/DIV VSOURCE 100mV/DIV VINTVCC 10V/DIV VOUT AC COUPLED 500mV/DIV VOUT AC COUPLED 50mV/DIV 100s/DIV FRONT PAGE APPLICATION VIN = 350V, IOUT = 3mA 8315 G07 VBIAS 10V/DIV 10ms/DIV 8315 G08 FRONT PAGE APPLICATION VIN = 350V, IOUT = 10mA TO 440mA 50ms/DIV 8315 G09 FRONT PAGE APPLICATION VIN = 350V, ROUT = 28 8315fa 4 For more information www.linear.com/LT8315 LT8315 TYPICAL PERFORMANCE CHARACTERISTICS DRAIN Shutdown Current BIAS Quiescent Current Depletion Startup Current 160 VDRAIN = 100V VDRAIN = 630V 1.6 -55C 25C 150C 1.4 45 30 15 120 1.2 IINTVCC (mA) QUIESCENT CURRENT (A) SHUTDOWN CURRENT (A) 60 TA = 25C, unless otherwise noted. 80 40 1.0 0.8 0.6 0.4 0.2 0 -50 -25 0 0 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 0 8315 G10 FB Regulation Voltage 3 6 9 VINTVCC (V) EN/UVLO Rising EN/UVLO Falling 12 15 8315 G12 TC Pin Voltage 1.240 1.80 1.235 1.30 1.25 1.20 1.60 1.230 TC VOLTAGE (V) FB REGULATION VOLTAGE (V) ENABLE THRESHOLD (V) 0 8315 G11 EN/UVLO Threshold 1.35 0 25 50 75 100 125 150 TEMPERATURE (C) 1.225 1.220 1.215 1.210 1.40 1.20 1.00 1.205 1.15 -50 -25 0 1.200 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 0 IREG/SS Pin Current Switching Frequency Limit Switch Current Limit 150 110 10.4 145 105 10.3 135 FREQUENCY (kHz) 10.0 9.9 9.8 130 125 5 4 3 9.7 100 MAXIMUM SWITCHING FREQUENCY SOURCE VOLTAGE (mV) 140 10.1 MINIMUM SWITCHING FREQUENCY 95 85 25 20 15 2 10 1 5 9.5 -50 -25 0 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 8315 G16 0 25 50 75 100 125 150 TEMPERATURE (C) 8315 G17 MAXIMUM CURRENT LIMIT 90 9.6 0 25 50 75 100 125 150 TEMPERATURE (C) 8315 G15 10.5 10.2 0 8315 G14 8315 G13 IREG/SS CURRENT (A) 0.80 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 0 -50 -25 MINIMUM CURRENT LIMIT 0 25 50 75 100 125 150 TEMPERATURE (C) 8315 G18 8315fa For more information www.linear.com/LT8315 5 LT8315 TYPICAL PERFORMANCE CHARACTERISTICS Minimum Switch-Off Time 400 900 350 850 300 800 250 750 OFF TIME (ns) ON TIME (ns) Minimum Switch-On Time 200 150 700 650 100 600 50 550 0 -50 -25 0 TA = 25C, unless otherwise noted. 500 -50 -25 25 50 75 100 125 150 TEMPERATURE (C) 0 8315 G19 8315 G20 DCM Pin Threshold 16 0 -20 RESISTANCE () DCM CURRENT (A) 12 -60 RESONANT VALLEY DETECT -100 -120 -140 -160 10 8 6 4 FLYBACK COLLAPSE DETECT -180 2 -200 -220 -50 -25 Switch RDS(ON) 14 -40 -80 25 50 75 100 125 150 TEMPERATURE (C) 0 25 50 75 100 125 150 TEMPERATURE (C) 0 -50 -25 8315 G21 0 25 50 75 100 125 150 TEMPERATURE (C) 8315 G22 8315fa 6 For more information www.linear.com/LT8315 LT8315 PIN FUNCTIONS DRAIN (Pins 1,2,3): Drain of the 630V Internal Power Switch and Startup FET. Design a compact layout with the transformer and input capacitor, and minimize trace area to reduce EMI and voltage spikes. INTVCC (Pin 8): Internal Gate Driver Bias Voltage. During start-up, current from the DRAIN charges this pin to 12V. During operation, a linear regulator from BIAS maintains this voltage at 10V. Bypass locally with a 2.2F ceramic 15V capacitor. BIAS (Pin 9): Unregulated Input Voltage for the IC. This pin derives power from a third winding on the transformer to provide power to INTVCC. Bypass locally with a capacitor. DCM (Pin 10): Discontinuous Conduction Mode Detector. This pin detects the dV/dt of the switching waveform, ensuring accurate output voltage sampling and quasiresonant boundary-mode switching. Connect a capacitor with series resistance from this pin to the third winding. TC (Pin 11): Temperature Compensation Pin. This pin presents a proportional-to-absolute-temperature (PTAT) voltage, which is equal to the internal 1.22V reference voltage at 25C and rises with temperature by 4.1mV/C, to compensate for the output rectifier diode. Connect an appropriate resistor from this pin to FB. FB (Pin 12): FeedBack Pin. The voltage appearing on this pin is sampled and regulated to equal the internal 1.22V reference voltage. Connect this pin to a resistor divider from the third winding to regulate the output voltage. IREG/SS (Pin 14): Current Regulation/Soft-Start Pin. A 10A current flows out of this pin. The resulting voltage sets the output current regulation point, as determined by an internal current regulation loop. Program the current with a resistor to GND, or connect a capacitor to implement soft-start. SMODE (Pin 16): Standby Mode Pin. Connect this pin to INTVCC to enable Standby Mode, which reduces the minimum switching frequency to 220Hz for ultralow quiescent power consumption. Connect to GND to disable. EN/UVLO (Pin 17): Enable/Undervoltage Lockout Pin. The chip will operate only if the voltage on this pin is greater than the internal 1.22V reference voltage. Connect to a resistor divider as desired, or connect to BIAS or INTVCC if UVLO functionality is not desired. SOURCE (Pin 18): Source of 630V Internal Power Switch. The voltage appearing on this pin is used for peak currentmode control and current limiting. Connect a currentsensing resistor to GND to program the current limit. Design a compact layout with the transformer and input capacitor to reduce EMI and voltage spikes. NC (Pin 19): No-Connect. This pin is electrically disconnected. Leave floating. GND (Pins 15, 20, 21): Ground. Solder the exposed pad (Pin 21) to a ground plane for heat sinking. VC (Pin 13): Loop Compensation Pin. An internal GM transconductance amplifier feeds this pin with an error current depending on the sampled FB voltage. The resulting voltage determines the switching frequency and peak current limit for power delivery. Connect a series R-C network to stabilize the regulator. 8315fa For more information www.linear.com/LT8315 7 LT8315 BLOCK DIAGRAM NPS :1 VIN CIN ZSNUB LPRI DSNUB * * DOUT VOUT + COUT LSEC VOUT - 9 BIAS LDO CBIAS - 10V + VUVLO DRAIN 1, 2, 3 630V DEPLETION FET M2 + DBIAS 8 17 MASTER LATCH EN/UVLO BIAS/REF CONTROL TSD CINTVCC 16 - INTVCC SMODE 630V POWER FET S Q M1 DRIVER R CDCM :NTS * RDCM 10 RFB2 12 LTER RFB1 DCM FB BOUNDARY DETECT x1 VOLTAGE CONTROLLED OSCILLATOR SOURCE CURRENT COMPARATOR + RSNS x10 GND 15, 20, 21 - S&H 18 RTC 11 TC x1 +4.1mV/C CURRENT ERROR AMP VOLTAGE ERROR AMP - GM 1.22V 1.25x(1-D) - + 10A + 13 VC RC 14 IREG/SS 8315 BD RIREG CC 8315fa 8 For more information www.linear.com/LT8315 LT8315 OPERATION The LT8315 is a high-voltage current-mode switching regulator designed for the isolated flyback topology. The special problem normally encountered in such circuits is that information relating to the output voltage on the isolated secondary side of the transformer must be communicated to the primary side in order to achieve regulation. This is often performed by opto-isolator circuits, which waste output power, require extra components that increase the cost and physical size of the power supply, and exhibit trouble due to limited dynamic response, nonlinearity, unit-to-unit variation, and aging over life. The LT8315 does not need an opto-isolator because it derives its information about the isolated output voltage by examining the flyback pulse waveform appearing on a tertiary winding on the transformer. The output voltage is easily programmed with two resistors. Boundary Mode Operation Boundary mode is a variable frequency, current-mode switching scheme. The internal N-channel MOSFET turns on and the inductor current increases until it reaches the limit determined by the voltage on the VC pin and the sense resistor's value. After the internal MOSFET turns off, the voltage on the tertiary winding rises to the output voltage multiplied by the transformer tertiary-to-secondary turns ratio. After the current through the output diode falls to zero, the voltage on the tertiary winding falls. A boundary mode detection comparator on the DCM pin detects the negative dV/dt associated with the falling voltage and triggers the sample-and-hold circuit to sample the FB voltage. When the tertiary voltage reaches its minimum and stops falling, the boundary mode comparator turns the internal MOSFET back on for minimal switching energy loss. The LT8315 features a boundary mode control method (also called critical conduction mode), where the part operates at the boundary between continuous conduction mode and discontinuous conduction mode. Due to boundary mode operation, the output voltage can be determined from the tertiary winding's voltage when the secondary current is almost zero. This method improves load regulation without extra resistors and capacitors. Boundary mode operation returns the secondary current to zero every cycle, so parasitic resistive voltage drops do not cause load regulation errors. Boundary mode also allows the use of a smaller transformer compared to continuous conduction mode and does not exhibit subharmonic oscillation. The Block Diagram shows an overall view of the system. Many of the blocks are similar to those found in traditional switching regulators, including current comparator, internal reference, LDO, logic, timers, and an N-channel MOSFET. The novel sections include a special sampling error amplifier, a temperature compensation circuit, an output current regulator, and a depletion-mode startup FET. As the load gets lighter, the peak switch current decreases. Maintaining boundary mode requires the switching frequency to increase. An excessive switching frequency increases switching and gate charge losses. To limit these losses, the LT8315 features an internal oscillator which limits the maximum switching frequency to 140kHz. Once the switching frequency hits this limit, the part starts to reduce its switching frequency and operates in discontinuous conduction mode. Depletion Startup FET The LT8315 features an internal depletion mode MOSFET. At startup, this transistor charges the INTVCC capacitor so that the LT8315 has power to begin switching. This removes the need for an external bleeder resistor or other components. Discontinuous Conduction Mode Operation Low Ripple Burst Mode Operation Unlike traditional flyback converters, the internal MOSFET has to turn on and off to generate a flyback pulse in order to update the sampled output voltage. The duration of a well-formed flyback pulse must exceed the minimum-off time for proper sampling. To this end, a minimum switch turn-off current is necessary to ensure a flyback pulse of sufficient duration. 8315fa For more information www.linear.com/LT8315 9 LT8315 OPERATION As the load gets very light, the LT8315 reduces switching frequency while maintaining the minimum current limit in order to reduce current delivery while still properly sampling the output voltage. Because flyback pulses must be generated to regulate the output, a minimum switching frequency of 3.5kHz is enforced. The minimum switching frequency determines how often the output voltage is sampled and introduces a minimum load requirement. Tying the SMODE pin to INTVCC enables Standby Mode, which reduces the minimum switching frequency to 220Hz, reducing the minimum load requirement at the expense of a longer period between samples. CV/CC Regulation Like a traditional voltage regulator, the LT8315 implements a GM transconductance amplifier that regulates the output voltage. In addition, the LT8315 includes a current regulation loop which regulates the estimated output current to a point set by the voltage on the IREG/ SS pin. Below the current setpoint, the output voltage is regulated for constant-voltage (CV) regulation. Below the voltage setpoint, the output current is regulated for constant-current (CC) regulation. APPLICATIONS INFORMATION The LT8315 is designed to be an easy-to-use, yet fullyfeatured flyback regulator. With proper technique, it is simple to build an efficient and robust power solution. The depletion FET is current-limited to avoid destructive power levels. To ensure start-up, do not load INTVCC or BIAS with excessive current while the chip is starting. However, don't let the simplicity beguile you into sloppy lab practices: the voltage and power levels involved can be lethal. Milliamperes from a high voltage power supply can cause heart fibrillation and death. Never touch conductive nodes while the circuit is active, and keep one hand behind your back while probing. Conduct lab work in the presence of an assistant, who can perform first aid in case of emergency. ENABLE and Undervoltage Lockout (UVLO) Depletion Startup FET The LT8315 features an internal depletion-mode FET, which has a negative threshold voltage and is therefore normally on. At startup, this FET charges the INTVCC capacitor to 12V so that the LT8315 has power to begin switching. This removes the need for an external bleeder resistor or other startup components. Once INTVCC is charged, the depletion-mode FET turns off. A resistive divider from VIN to the EN/UVLO pin implements undervoltage lockout (UVLO). The EN/UVLO pin threshold is set at 1.22V. Upon startup, the EN/UVLO pin exhibits a ~65mV hysteresis voltage to prevent oscillations. The EN/UVLO pin can also be driven with logic levels and set by the output pin of a digital controller. Otherwise, EN/UVLO can also be tied to BIAS or INTVCC to keep the chip enabled. Output Voltage The output voltage is programmed by the RFB1 and RFB2 resistors depicted in the Block Diagram. The LT8315 operates similarly to traditional current-mode switchers, except in the use of a unique sample-and-hold error amplifier, which regulates the isolated output voltage from the sampled flyback pulse. 8315fa 10 For more information www.linear.com/LT8315 LT8315 APPLICATIONS INFORMATION Operation is as follows: when the power switch M1 turns off, the voltage across the tertiary winding rises. The amplitude of the flyback pulse is given as: VFLBK = (VOUT + VF + ISEC * ESR) * NTS, With a fixed value for RFB1 (such as 10k) chosen, rearrangement of the expression for VOUT yields the starting value for RFB2: +V V RFB2 = RFB1 * OUT F * NTS - 1 1.22V where VF = Output diode (DOUT) forward-biased voltage where ISEC = Transformer secondary current VOUT = Desired output voltage ESR = Parasitic resistance of secondary circuit VF = Output diode (DOUT) forward voltage 300mV NTS = Transformer tertiary-to-secondary turns ratio NTS = Transformer tertiary-to-secondary turns ratio The voltage divider formed by RFB1 and RFB2 feeds a scaled version of the flyback pulse to the FB pin, where it is sampled and fed to the error amplifier. Because the sample-and-hold circuit samples the voltage when the secondary current is nearly zero, the (ISEC * ESR) term in the VFLBK equation can be ignored. Power up the application with the final power components installed and the starting RFB2 value, and measure the regulated output voltage, VOUT(MEAS). The final RFB2 value can be adjusted to: The internal 1.22V reference voltage feeds the non-inverting input of the error amplifier. The high gain of the overall loop causes the FB voltage to be nearly equal to the reference voltage. The resulting flyback voltage VFLBK can be expressed as: R VFLBK = 1+ FB2 * 1.22V RFB1 Combining with the previous VFLBK equation and solving for VOUT yields: R 1.22V VOUT = 1+ FB2 * - VF RFB1 NTS Due to the fast nature of the flyback pulse, it is recommended to keep RFB1 between 1k and 10k in order to preserve the resistor divider's dynamic response. RFB2(FINAL) (RFB2 + RFB1) * VOUT VOUT(MEAS) - RFB1 Once the final RFB2 value is selected, the regulation accuracy from board to board for a given application will be very consistent, typically within 5% when including device variation of all the components in the system (assuming resistor tolerances and transformer windings matching within 1%). However, if the transformer or the output diode is changed, or the layout is dramatically altered, there may be some change in VOUT. Example: Consider a 12V output supply with an output diode whose forward voltage at nearly zero current is 300mV at room temperature. If the tertiary-to-secondary ratio NTS is 1 and RFB1 is 10k, then RFB2 is calculated as 90.9k. The application is powered up and the output is slightly high at 12.2V, so RFB2 is adjusted to 88.7k. Output Diode Temperature Compensation Reiterating the equation for VOUT, Selecting the Actual RFB2 Resistor Value The LT8315 uses a unique sampling scheme to regulate the isolated output voltage. Due to its sampling nature, the scheme exhibits repeatable delays and error sources, which will affect the output voltage and force a re-evaluation of the resistor values. VOUT = 1+ RFB2 1.22V * NTS RFB2 VF The first term in the VOUT equation is insensitive to temperature, but the output diode forward voltage VF has a significant negative temperature coefficient (from -1mV/C 8315fa For more information www.linear.com/LT8315 11 LT8315 APPLICATIONS INFORMATION to -2mV/C). Such a temperature coefficient produces approximately 200mV to 400mV output voltage variation across operating temperature. At higher output voltages, the resulting variation may be unimportant as it represents a small fraction of the total output. However, for lower output voltages, the diode temperature coefficient accounts for a large output voltage error. To correct this error, the TC pin provides a buffered proportional-to-absolute-temperature (PTAT) voltage. At room temperature, this voltage is equal to the internal 1.22V reference, and it has a +4.1mV/C temperature coefficient. The output diode's temperature coefficient TCF can easily be found experimentally by applying a uniform temperature to both the output diode and the LT8315. First, RFB1 and RFB2 are adjusted to give the desired output voltage at room temperature. The temperature is then raised or lowered by a known amount to a new temperature, and the diode temperature coefficient is found as: TCF = VOUT(25C) - VOUT(TNEW) TNEW - 25C where VOUT(25C) = VOUT measured at room temperature VOUT(TNEW) = VOUT measured at new temperature TNEW = New temperature in Celsius Alternatively, TCF can be found more accurately by measuring VOUT at two extremes of temperature and computing: TCF = - VOUT T ing a nominal TCF value (such as -1.5mV/C) may yield a satisfactory result. With the output diode's temperature coefficient known, a resistor RTC is then attached from the TC pin to the FB pin. Its value can be calculated as: RTC = -RFB2 * 4.1mV / C TCF * NTS Example: If the output diode's temperature coefficient TCF is found experimentally to be -1.9mV/C, then with RFB2 = 88.7k, a RTC value of 191k will yield a temperatureinvariant output voltage. Sense Resistor Selection The resistor RSNS between the SOURCE pin and GND should be selected to provide an adequate switch current to drive the application without exceeding the current limit threshold. At maximum current delivery, current limit occurs when the SOURCE pin voltage is 100mV. In boundary mode, the maximum output current will depend on the duty cycle D and is given by: IOUT(MAX) 100mV * (1 D) * NPS 2 * RSNS where NPS = Transformer primary-to-secondary turns ratio D ( VOUT + VF ) * NPS ( VOUT + VF ) * NPS + VIN VIN = Power supply voltage. It should be noted that for this measurement, it is critical that the entire board be heated or cooled uniformly, for example by an oven. A heat gun or freeze spray will not suffice, since the heating and cooling will not be uniform, and dramatic temperature mismatch between the LT8315 and the output diode will cause significant error. It should be noted that the worst-case occurs at minimum VIN, so DVIN(MIN) should be calculated assuming VIN = VIN(MIN). Solving for the sense resistor value: If no method is available to apply uniform heat or cooling, extrapolating data from the diode's data sheet or assum- A factor of 80% is introduced to compensate for system delays and tolerances, but it may need adjustment for the final application. RSNS = 1- DVIN(MIN) IOUT(MAX) * 50mV * NPS * 80% 8315fa 12 For more information www.linear.com/LT8315 LT8315 APPLICATIONS INFORMATION Example: A 12V output voltage is generated from a VIN = 350V input that can drop as low as VIN(MIN) = 250V. If a transformer with primary-to-secondary turns ratio NPS = 10 is selected and it is to supply a maximum output current IOUT(MAX) = 750mA, then the duty cycle is DVIN(MIN) 33% and the sense resistor is calculated RSNS = 356m. A 330m resistor is selected. A more accurate value for RSNS can be obtained by finding D experimentally with an oscilloscope and electronic load. Output Power Compared with a buck or a boost converter, a flyback converter has a complicated relationship between the input and output currents. Boost converters have relatively constant maximum input current regardless of input voltage, while buck converters have relatively constant maximum output current regardless of input voltage, owing to the fact that they have continuous input and output currents respectively. A flyback converter, however, has both discontinuous input and output currents. The duty cycle affects both input and output currents, making it hard to predict maximum output power. The graphs in Figure 1 through Figure 4 show the typical maximum output power possible for the output voltages 5V, 12V, 24V and 48V. The maximum output power curve is the calculated output power if the switch voltage is 510V during the switch-off time. 120V of margin is left for the leakage inductance voltage spike. To achieve this power level at a given input, a winding ratio must be calculated to stress the switch to 510V, resulting in some odd ratio values. The curves below the maximum output power curve are examples of common winding ratio values and the amount of output power at given input voltages. 16 14 THEORETICAL MAXIMUM 12 MAXIMUM OUTPUT POWER (W) MAXIMUM OUTPUT POWER (W) 16 NPS = 40 10 NPS = 20 8 6 NPS = 10 4 NPS = 5 2 0 0 100 200 300 VIN (V) 400 NPS = 20 NPS = 10 8 6 NPS = 5 4 NPS = 2 2 0 100 8315 F01 200 300 VIN (V) 400 500 8315 F02 Figure 2. Maximum Power, VOUT = 12V 16 16 14 THEORETICAL MAXIMUM 12 10 NPS = 10 MAXIMUM OUTPUT POWER (W) MAXIMUM OUTPUT POWER (W) 10 0 500 Figure 1. Maximum Power, VOUT = 5V NPS = 5 8 6 NPS = 2 4 NPS = 1 2 0 14 THEORETICAL MAXIMUM 12 0 100 200 300 VIN (V) 400 500 14 THEORETICAL MAXIMUM 12 10 8 NPS = 2 6 NPS = 1 4 NPS = 0.5 2 0 0 8315 F03 Figure 3. Maximum Power, VOUT = 24V NPS = 5 100 200 300 VIN (V) 400 500 8315 F04 Figure 4. Maximum Power, VOUT = 48V 8315fa For more information www.linear.com/LT8315 13 LT8315 APPLICATIONS INFORMATION Table 1. Predesigned Transformers -- Typical Specifications TRANSFORMER PART NUMBER LPRI (mH) NP:NS:NT ISOLATION VENDOR TARGET APPLICATIONS PS16-077 4 24:1:4 Reinforced Sumida 140V-380V to 5V/1.5A PS16-051 4 10:1:2 Reinforced Sumida 140V-380V to 12V/0.6A PS15-195 4 3:1:1 Reinforced Sumida 100V-500V to 12V/0.2A PS16-078 4 5:1:1 Reinforced Sumida 140V-380V to 24V/0.3A 750316022 3.3 24:1:4 Functional Wurth 140V-380V to 5V/1.5A 7508111324 2.75 10:1:1 Reinforced Wurth 140V-380V to 12V/0.6A 7508111518 2.4 2.5:1:0.25 Reinforced Wurth 140V-380V to 48V/0.15A The following equation calculates output power: POUT = 0.5 * * VIN * D * ISW(MAX) Flyback Transformer Modeling A flyback transformer can be thought of as an ideal transformer with a parallel magnetizing inductance and series leakage inductances, as shown in Figure 5. where = Efficiency 80% LLEAK(PRI) ( VOUT + VF ) * NPS D ( VOUT + VF ) * NPS + VIN NPS :1 LLEAK(SEC) LPRI ISW(MAX) = Max. switch current limit = 100mV/RSNS The calculated power is approximate, and does not take into account timing variations caused by circuit parasitics. The actual output power must be evaluated on the bench. Example: Consider a 12V output converter with a VIN(MIN) of 250V and a VIN(MAX) of 390V. With a ten-to-one primaryto-secondary winding ratio NPS = 10 and a sense resistor RSNS = 330m, the maximum power output is 11W at VIN(MAX) = 390V but lowers to 10W at VIN(MIN) = 250V. Selecting a Transformer Transformer specification and design is possibly the most critical part of successfully applying the LT8315. In addition to the usual list of guidelines dealing with high-frequency isolated power supply transformer design, the following information should be carefully considered. Linear Technology has worked with several leading magnetic component manufacturers to produce pre-designed flyback transformers for use with the LT8315. Table 1 shows the details of these transformers. 8315 F05 IDEAL Figure 5. Transformer Model The magnetizing inductance, which is the mutual inductance shared by both primary and secondary windings, is essential for absorbing energy and delivering it to the load. It stores energy in magnetic flux lines that pass through both primary and secondary windings. If the leakage inductances are small, the magnetizing inductance can be measured by leaving the secondary open-circuited and measuring the inductance of the primary, resulting in an inductance LPRI. The magnetizing inductance can also be measured from the secondary by leaving the primary open-circuited and measuring the secondary inductance LSEC. The relationship between the primary-referred magnetizing inductance and secondaryreferred magnetizing inductance is given by the primaryto-secondary turns ratio NPS as: LPRI = LSEC * NPS2 The transformer also has leakage inductances, which are parasitic inductances associated with each winding. These 8315fa 14 For more information www.linear.com/LT8315 LT8315 APPLICATIONS INFORMATION inductances store energy in magnetic flux lines which "leak" out of the magnetic core and do not pass through both windings, and therefore represent self-inductances whose energy cannot be transferred through the transformer. As such, they contribute to energy loss and reduced converter efficiency. If the leakage inductances are small, the combined leakage inductance can be measured by short-circuiting the secondary and measuring the primary inductance. This results in a primary-referred inductance, LLEAK = LLEAK(PRI) + LLEAK(SEC) * NPS2 The leakage inductance and magnetizing inductance are related by the coupling coefficient k according to the relation: k= L PRI L PRI + L LEAK / 2 Coupling coefficients of k=99% are common, and are a function of transformer construction and materials. Increased voltage isolation between primary and secondary is often desired for safety purposes, but generally reduces the coupling coefficient and increases leakage inductance. Bifilar windings maximize the coupling coefficient, but are often undesirable because of their minimal isolation and increased primary-to-secondary capacitance. In the end, a reasonable trade-off between isolation and coupling coefficient must be made. Magnetizing Inductance Requirement The appropriate magnetizing inductance depends on the LT8315's minimum switch-on time, its minimum switchoff time, and output power. The conduction of secondary current reflects the output voltage onto the tertiary winding during the flyback pulse. The LT8315 obtains output voltage information from the reflected output voltage on the FB pin. The sample-andhold error amplifier needs a minimum of 800ns to settle and sample the reflected output voltage. In order to ensure proper sampling, the secondary winding needs to conduct current for at least 800ns. The minimum value for primary-side magnetizing inductance is given by: LPRI tOFF(MIN) * NPS * ( VOUT + VF ) ISW(MIN) where tOFF(MIN) = Minimum switch-off time = 800ns ISW(MIN) = Minimum switch current limit = 20mV/RSNS The LT8315 has a minimum switch-on time that prevents the chip from turning on the power switch for a period shorter than 250ns in order to blank the initial switch turn-on current spike. If the inductor current exceeds the minimum switch current limit during that time, the minimum load current will increase. Therefore, the following equation must also be observed: LPRI tON(MIN) * VIN(MAX) ISW(MIN) where tON(MIN) = Minimum Switch-On Time = 250ns Additionally, the magnetizing inductance must be large enough to provide sufficient power to the output when the LT8315 operates at maximum frequency. This creates a third requirement for magnetizing inductance: LPRI 2 * (VOUT + VF ) *IOUT(MAX) *ISW(MAX)2 * fSW(MAX) where ISW(MAX) = Maximum switch current = 100mV/RSNS IOUT(MAX) = Maximum load current fSW(MAX) = Maximum switching frequency = 140kHz = Efficiency 80% In general, choose a transformer with its primary magnetizing inductance about 20% to 50% larger than the minimum values calculated above. 8315fa For more information www.linear.com/LT8315 15 LT8315 APPLICATIONS INFORMATION Example: For a 12V/750mA output converter with VIN(MAX) = 390V, VF = 300mV, NPS = 10, and RSNS = 330m, the first equation requires LPRI 1.64mH, the second equation requires LPRI 1.61mH, and the third equation requires LPRI 1.83mH. A reasonable standard value for primary inductance is LPRI = 2.2mH. Saturation Current The current in the transformer windings should not exceed its rated saturation current. Beyond its saturation value, the inductance drops and the current rises to an uncontrolled value, causing extra power dissipation and possible failure. Choose a transformer whose primary saturation current is at least 30% greater than ISW(MAX), which is 100mV/RSNS. Turns Ratios Typically, choose the transformer primary-to-secondary turns ratio NPS to maximize available output power. For low output voltages, a larger NPS ratio can be used to maximize the transformer's current gain. However, remember that the DRAIN pin sees a voltage that is equal to VIN plus the output voltage multiplied by NPS. Additionally, leakage inductance will cause a voltage spike (VLEAKAGE) that adds to this reflected voltage. This total quantity needs to remain below the 630V absolute maximum rating of the DRAIN pin to prevent breakdown of the internal power switch. Together these conditions place an upper limit on the turns ratio NPS for a given application. Choose a turns ratio low enough to ensure: NPS < 630V - VIN(MAX) - VLEAKAGE VOUT + VF For producing high output voltages, a low ratio NPS may be used. However, the multiplied capacitance presented to the DRAIN node may cause ringing that exceeds the 250ns tON(MIN), causing light-load instability. Fully evaluate these applications before use with the LT8315. During operation, the LT8315 derives its power from a tertiary winding through its BIAS pin. BIAS must be maintained between 10V and 40V for proper operation. This dictates a tertiary-to-secondary turns ratio NTS of: 10V VOUT < N TS < 40V VOUT Example: For VOUT = 12V, NTS must lie between 0.83 and 3.33, or a 5:6 and 10:3 tertiary-to-secondary ratio respectively. Because the output voltage is measured through the voltage appearing on the third winding, NTS directly affects the output voltage regulation accuracy. For best results, make sure the transformer is manufactured with a precise turns ratio specified within 1%. Leakage Inductance and Snubbers Any leakage inductance on either the primary or secondary windings causes a voltage spike to appear on the primary after the power switch turns off. This spike is increasingly prominent at higher load currents where more energy is stored in the leakage inductance. This energy cannot be delivered to the load, and must be dissipated as heat. It is thus very important to minimize transformer leakage inductance. When designing an application, adequate margin should be kept for the worst-case leakage voltage spikes even under overload conditions. In most cases, the reflected output voltage on the primary plus VIN should be kept below 510V, as shown in Figure 6. This leaves 120V margin for the leakage spike across line and load conditions. A larger voltage margin will be required for poorly wound transformers with excessive leakage inductance. In addition to the voltage spikes, the leakage inductance also causes the DRAIN pin to ring for a while after the power switch turns off. To prevent the voltage ringing from falsely triggering the boundary mode detector, the LT8315 internally blanks the boundary mode detector for 800ns. Any ringing after 800ns may trigger the power switch to turn back on again before the secondary current falls to zero, so the leakage inductance spike and associated ringing should be limited to less than 800ns. 8315fa 16 For more information www.linear.com/LT8315 LT8315 APPLICATIONS INFORMATION VSW VSW <630V VSW <630V <630V VLEAKAGE VLEAKAGE <510V VLEAKAGE <510V <510V tOFF > 800ns tOFF > 800ns tOFF > 800ns TIME TIME No Snubber TIME with DZ Snubber with RC Snubber 8315 F06 Figure 6. Maximum Voltages for SW Pin Flyback Waveform L L * Z D * C * * R 8315 F06a 8315 F06b DZ Snubber RC Snubber Figure 7. Snubber Circuits A snubber circuit is recommended for most applications. Figure 7 shows two types of snubber circuits that can protect the internal power switch: the DZ (diode-Zener) snubber and the RC (resistor-capacitor) snubber. The DZ snubber ensures a well-defined and consistent clamping voltage and has slightly higher power efficiency, while the RC snubber quickly damps the voltage spike ringing and provides better load regulation and EMI performance. Figure 6 shows the flyback waveforms with the DZ and RC snubbers. For the DZ snubber, proper care must be taken when choosing both the diode and the Zener diode. Choose a fast-recovery diode that has a reverse-voltage rating higher than the maximum DRAIN pin voltage. The Zener diode breakdown voltage should be chosen to balance power loss and switch voltage protection. The best compromise is to choose the largest voltage breakdown. Use the following equation to make the proper choice: VZENER(MAX) 630V - VIN(MAX) Multiple Zener diodes may be placed in series to attain the required voltage and power dissipation. The Zener diode must be rated to absorb the power loss in the clamp, which is due to energy storage in the leakage inductance and the primary-to-secondary commutation time, which decreases with higher clamp voltage. A 500mW Zener is typically recommended. 8315fa For more information www.linear.com/LT8315 17 LT8315 APPLICATIONS INFORMATION For the RC snubber, the recommended design approach is to power up at low voltage to avoid overvoltage stress, measure the period of the ringing on the DRAIN pin when the power switch turns off without the snubber (TRING), and then add capacitance CSNUBBER (starting with 100pF) until the period of the ringing is 1.5 to 2 times longer (TRING(SNUBBED)). The change in period will determine the value of the parasitic capacitance CDRAIN, from which the parasitic inductance LLEAK can also be determined, according to the equations: CDRAIN = CSNUBBER TRING(SNUBBED) TRING LLEAK = TRING 2 2 * 2 1 1 Secondary Leakage Inductance Leakage inductance on the secondary forms an inductive divider that effectively reduces the size of the tertiaryreferred flyback pulse used for voltage feedback. This will increase the output voltage by a similar percentage. Note that, unlike leakage spike behavior, this phenomenon is load independent. To the extent that the secondary leakage inductance is a constant percentage of mutual inductance (over manufacturing variations), this can be accommodated by adjusting the RFB2/RFB1 resistor ratio. Winding Resistance CDRAIN With the value of the DRAIN node capacitance and leakage inductance known, a resistor can be added in series with the snubber capacitor to dissipate power and critically dampen the ringing. The equation for deriving the optimal series resistance is: R SNUBBER = expected reverse voltage. A snubber or clamp may be implemented to reduce the voltage spike if it is desired to use a lower reverse voltage diode. L LEAK CDRAIN Energy absorbed by the RC snubber will be converted to heat and will not be delivered to the load. In high power applications, the snubber resistor may need to be sized for thermal dissipation. Note that the DRAIN capacitance is often dominated by transformer interwinding capacitance. Also note that oscilloscope probes present considerable loading capacitance. Use of low-capacitance, high-voltage 100x probes is recommended. Leakage Inductance and Output Diode Stress The output diode may also see increased reverse voltage stresses from leakage inductance. While it nominally sees a reverse voltage of the input voltage divided by NPS plus the output voltage when the MOSFET power switch turns on, the capacitance on the output diode and the leakage inductance form an LC tank which may ring beyond that Resistance in either the primary or secondary will reduce conversion efficiency. Good output voltage regulation will be maintained despite winding resistance due to the boundary/discontinuous conduction mode operation of the LT8315. Boundary Mode Detection Boundary mode is a variable frequency switching scheme that always returns the secondary current to zero with every cycle. The DCM pin uses a fast, current-input comparator in combination with a small capacitor CDCM to detect when the flyback waveform's dV/dt is negative, indicating that the secondary diode has turned off and the flyback pulse on the tertiary winding is falling. To avoid false tripping due to leakage inductance ringing, a blanking time of 800ns is applied after the switch turns off. The detector triggers when CDCM draws 170A of current out of the DCM pin. This information is used to set the timing of the FB sample-and-hold and estimate the output current. This is not the best time to turn the switch on because the DRAIN voltage is still nearly VIN + (VOUT * NPS), and turning the switch on would waste all the energy stored in the parasitic capacitance on the switching node. When the secondary current reaches zero, discontinuous ringing begins and the energy in the parasitic capacitance on the 8315fa 18 For more information www.linear.com/LT8315 LT8315 APPLICATIONS INFORMATION switch node resonates with the transformer's magnetizing inductance, delivering this energy back to VIN. The minimum voltage of the DRAIN node during this discontinuous ring is VIN - (VOUT * NPS). This is the optimal moment to turn the switch back on, and the LT8315 does this by sensing when current drawn out of DCM falls to 85A. This switching technique increases efficiency by up to 5%. to zero, the LT8315 reduces its minimum switching frequency by a factor of 16 from 3.5kHz to 220Hz. Typical CDCM values range from 10pF to 100pF. A good starting value is 47pF. If the LT8315 is observed not to run in boundary mode, then increasing this capacitor will help. An unnecessarily large CDCM value can cause premature switch turn-on and increased power loss. Output Current Regulation and Soft-Start Excessive current delivered to the DCM pin can cause erratic behavior. To avoid this, a resistor RDCM can be added in series with CDCM to limit the current. Typical values range from 5k to 50k. This reduces the minimum load current by a factor of 16, at the cost of slower transient response. Because the output voltage is sampled only once every 4.6ms, the LT8315 will be unable to respond to load steps for up to this period. Using duty cycle information and the current limit set by the VC pin, the LT8315 estimates the output current and regulates it to a setpoint determined by the voltage on the IREG/SS pin. The output current is regulated according to the equation: N *V IOUT = PS IREG/SS 25 * R SNS where Operation Under Light Output Loads The LT8315 detects the output voltage from the flyback pulse appearing on the tertiary winding, which requires delivering power to the output. Thus, the LT8315 delivers a minimum amount of energy even during light load conditions to ensure accurate output voltage information. The minimum operating frequency at minimum load is approximately 3.5kHz. The minimum delivery of energy creates a minimum load requirement on the output of approximately 1% of the maximum load power. A Zener diode sufficiently rated to handle the minimum load power can be used to provide a minimum load without decreasing efficiency in normal operation. In selecting a Zener diode for this purpose, the Zener voltage should be high enough that the diode does not become the load path during transient conditions but the voltage must still be low enough that the MOSFET and output voltage ratings are not exceeded when the Zener functions as the minimum load. Standby Mode Operation For extremely low no-load power dissipation, the LT8315 features a standby mode which is enabled by tying the SMODE pin to INTVCC. When the load current has dropped VIREG/SS = Voltage on IREG/SS pin. A trimmed 10A current flows out of the IREG/SS pin, so that a resistor tied from this pin to GND programs the output current according to the equation: RIREG/SS = 2.5M *IOUT * RSNS NPS Example: For an application with RSNS = 330m, NPS = 10, and a desired regulated output current IOUT = 0.5A, an IREG/SS resistor is selected RIREG/SS = 41.2k. Circuit parasitics, especially transformer capacitance, will influence the accuracy of output current regulation due to energy delivery to the parasitics. Although this effect is usually small, some iteration may be necessary if accuracy better than 5% is required. In this case, RIREG/SS can be implemented with a rheostat and adjusted until the desired output current is realized, and then replaced with a fixed-value resistor for production. When the rheostat is present, a small bypass capacitor is helpful to attenuate switching interference pickup by the rheostat. Additionally, an RC snubber placed across the secondary rectifier can improve current regulation accuracy. 8315fa For more information www.linear.com/LT8315 19 LT8315 APPLICATIONS INFORMATION Soft-start functionality can also be implemented by connecting a capacitor from the IREG/SS pin to GND. The 10A current will act to charge the external soft-start capacitor. At startup, the regulated output current will rise monotonically until reaching voltage regulation. The softstart capacitor then charges entirely and the output current regulation loop will not interfere with voltage regulation. In order to avoid an undervoltage condition which causes the chip to shut down, the combined capacitance on the INTVCC and BIAS pins must be sufficient to power the LT8315 until the output achieves regulation. If power at VIN is removed, over-temperature protection is engaged, undervoltage lockout trips, or overcurrent in the sense resistor is detected, a 20 pull-down switch to GND discharges any capacitance on the IREG/SS pin for the duration of the fault plus 150s. Protection from Shorted Output Conditions During a shorted output condition, the LT8315 operates at the minimum operating frequency. In normal operation, the tertiary winding provides power to the IC, but the tertiary winding voltage collapses during a shorted condition. This causes the part's INTVCC UVLO of 8.2V to shutdown switching and charge through the depletion startup current VIN 100V TO 500V To protect the output diode from excessive power dissipation during overload conditions, it is advised to program the regulated output current with a resistor RIREG/SS. For voltage regulators, the programmed current should be 120% to 150% of the maximum load current to ensure current regulation does not interfere with voltage regulation. Loop Compensation The LT8315 is compensated using an external resistorcapacitor network on the VC pin. Typical values are in the range of RC = 100k and CC = 47nF. If too large an RC value is used, the part will be more susceptible to high frequency noise and jitter. If too small of an RC value is used, the transient performance will suffer. The value choice for CC is somewhat the inverse of the RC choice: if too small a CC value is used, the loop may be unstable and if too large a CC value is used, the transient performance will suffer. Transient response may be evaluated with a load step box and adjusted with an adjustable RC compensation network. Stability should be confirmed over the full range of load current and input voltage. D2 200 0.1F source. The part starts switching again when INTVCC has reached its turn-on voltage of 12V. T1 3:1:1 20k D1 VOUT+ 12V 10mA TO 200mA 500H 88.7k 47pF 10F BIAS EN/UVLO 191k 10k 120F VOUT- 4.5mH D4 DRAIN VC 22nF TC LT8315 SMODE 110k D3 FB INTVCC 10F 500H DCM SOURCE IREG/SS GND 390m 0.33F 88.7k T1: SUMIDA PS15-195 D1, D2: CENTRAL CMMR1U-04 D3: DIODES SMAJ75A D4: CENTRAL CMMR1U-08 8315 F08 Figure 8. 12V High Input Voltage Isolated Flyback Converter 8315fa 20 For more information www.linear.com/LT8315 LT8315 APPLICATIONS INFORMATION VIN 19V TO 400V D1 D1: CENTRAL CMDSH2-3 D2: CENTRAL CMMR1U-06 D3: C3D1P7060Q D4: MICRO COMMERCIAL SMBJ5350B-TP 11k D2 4.99k 10pF BIAS DCM EN/UVLO 10F FB INTVCC TC LT8315 10F SMODE DRAIN VC 22.1k 1.24k 10nF SOURCE IREG/SS 330nF GND 330m 100nF 1mH VOUT 12V 3mA TO 120mA 44F D3 D4 8315 F09 Figure 9. Wide Input Range Nonisolated 12V Buck Converter VIN 250V TO 350V D2 200 0.1F T1 20:1:4 20k 158k 47pF 10F BIAS 100k 22pF 10k 1.5mF VOUT- 4mH D4 DRAIN VC 47nF TC LT8315 SMODE 100k D3 FB INTVCC 10F 10H DCM EN/UVLO VOUT+ 5V 20mA TO 1.5A D1 160H SOURCE IREG/SS GND 330m T1: SUMIDA PS16-077 D1: DIODES SBR8U60P5 D2: DIODES BAV20WS-7-F D3: DIODES SMAJ200A D4: CENTRAL CMMR1U-08 69.8k 8315 F10 Figure 10. 83% Efficiency 5V/1.5A Isolated Flyback Converter 8315fa For more information www.linear.com/LT8315 21 LT8315 PACKAGE DESCRIPTION Please refer to http://www.linear.com/product/LT8315#packaging for the most recent package drawings. FE Package Variation: FE20(16) 20-Lead Plastic TSSOP (4.4mm) (Reference LTC DWG # 05-08-1990 Rev O) Exposed Pad Variation BB 6.40 - 6.60* (.252 - .260) 5.68 (.224) REF 5.68 (.224) REF 20 19 18 17 16 15 14 13 12 11 6.60 0.10 1.78 (.070) REF SEE NOTE 4 5.00 0.10 0.42 0.48 (.019) REF 0.45 0.05 1.78 6.40 (.070) (.252) BSC 0.2 0.42 (.016) REF 0.22 1.78 (.070) 0.80 0.10 0.65 BSC 1 2 3 4.30 - 4.50* (.169 - .177) 0.09 - 0.20 (.0035 - .0079) 0.25 REF NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE 1.20 (.047) MAX 0 - 8 0.65 (.0256) BSC 0.50 - 0.75 (.020 - .030) 8 9 10 4.83 (.1902) RECOMMENDED SOLDER PAD LAYOUT 0.195 - 0.30 (.0077 - .0118) TYP 0.05 - 0.15 (.002 - .006) FE20(16) (BB) TSSOP REV O 1014 4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE 8315fa 22 For more information www.linear.com/LT8315 LT8315 REVISION HISTORY REV DATE DESCRIPTION A 12/17 Changed Electrical Characteristics Table Symbols, VSENSE(MIN), VSENSE(MAX), VSENSE(ILIM), to VSOURCE(MIN), VSOURCE(MAX), VSOURCE(ILIM) PAGE NUMBER 3 Corrected graph title EN/UVLO Threshold 5 Corrected TCF equations: Inverted sign of the denominator and added negative sign to VOUT/T formula 12 Corrected Table 1 bottom row transformer part number 14 Changed Figure 9 VIN range from 20V to 560V, to 19V to 400V and added new D3 diode part number and corrected D3 diode symbol 21 Changed Typical Application resistor value of RC network on DCM pin to 20k 24 8315fa Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications Forismore information www.linear.com/LT8315 subject to change without notice. No license granted by implication or otherwise under any patent or patent rights of Analog Devices. 23 LT8315 TYPICAL APPLICATION 85% Efficient Universal Input Offline Power Supply 90VAC TO 265VAC F1 L 330H D2 200 T1 10:1:2 D1 20k 10F 10F 95.3k 47pF 10H 10F BIAS RT1 N 102k T1: SUMIDA PS16-051 D1: DIODES DFLS2100 D2: DIODES BAV20WS D3: DIODES SMAJ200A D4: CENTRAL CMMR1U-08 D5: CENTRAL CMHZ5243B 5.05k D4 SOURCE IREG/SS 82nF 22pF DRAIN VC 100k D5 4mH TC LT8315 SMODE 300F D3 FB INTVCC 10F 40H DCM EN/UVLO VOUT 12V / 0.55A 160H GND 330m Y1 59k 2.2nF 8315 TA02 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT8304/LT8304-1 100VIN Micropower Isolated Flyback Converter with 150V/2A Low IQ Monolithic No-Opto Flyback, SO-8 Package Switch LT8304-1 Is Recommended for High Output Voltages LT8300 100VIN Micropower Isolated Flyback Converter with 150V/260mA Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8303 100VIN Micropower Isolated Flyback Converter with 150V/0.45A Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8301 42VIN Micropower Isolated Flyback Converter with 65V/1.2A Switch Low IQ Monolithic No-Opto Flyback, 5-Lead TSOT-23 LT8302 42VIN Micropower Isolated Flyback Converter with 65V/3.6A Switch Low IQ Monolithic No-Opto Flyback, SO-8 Package LT8309 Secondary-Side Synchronous Rectifier Driver 4.5V VCC 40V, Fast Turn-On and Turn-Off, 5-Lead TSOT-23 LT3573/LT3574/ LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch, MSOP-16(12) LT3748 100V Isolated Flyback Controller 5V VIN 100V, No-Opto Flyback, MSOP-16(12) LT3798 Off-Line Isolated No-Opto Flyback Controller with Active PFC VIN and VOUT Limited Only by External Components LT8312 Boost Controller with Power Factor Correction VIN and VOUT Limited Only by External Components 8315fa 24 LT 1217 REV A * PRINTED IN USA www.linear.com/LT8315 For more information www.linear.com/LT8315 ANALOG DEVICES, INC. 2017