1
LT1229/LT1230
TYPICAL APPLICATIO
U
APPLICATIO S
U
DESCRIPTIO
U
FEATURES
Dual and Quad 100MHz
Current Feedback Amplifiers
The LT
®
1229/LT1230 dual and quad 100MHz current
feedback amplifiers are designed for maximum perfor-
mance in small packages. Using industry standard pinouts,
the dual is available in the 8-pin miniDIP and the 8-pin SO
package while the quad is in the 14-pin DIP and 14-pin SO.
The amplifiers are designed to operate on almost any
available supply voltage from 4V (±2V) to 30V (±15V).
These current feedback amplifiers have very high input
impedance and make excellent buffer amplifiers. They
maintain their wide bandwidth for almost all closed-loop
voltage gains. The amplifiers drive over 30mA of output
current and are optimized to drive low impedance loads,
such as cables, with excellent linearity at high frequencies.
The LT1229/LT1230 are manufactured on Linear
Technology’s proprietary complementary bipolar process.
For a single amplifier like these see the LT1227 and for
better DC accuracy see the LT1223.
■
100MHz Bandwidth
■
1000V/µs Slew Rate
■
Low Cost
■
30mA Output Drive Current
■
0.04% Differential Gain
■
0.1° Differential Phase
■
High Input Impedance: 25MΩ, 3pF
■
Wide Supply Range: ±2V to ±15V
■
Low Supply Current: 6mA Per Amplifier
■
Inputs Common Mode to Within 1.5V of Supplies
■
Outputs Swing Within 0.8V of Supplies
■
Video Instrumentation Amplifiers
■
Cable Drivers
■
RGB Amplifiers
■
Test Equipment Amplifiers
–
+
V
OUT
LT1229 • TA01
12.1k
R
F2
750Ω
1% RESISTORS
WORST CASE CMRR = 22dB
TYPICALLY = 38dB
V
OUT
= G (V
IN+
– V
IN–
)
R
F1
= R
F2
R
G1
= (G – 1) R
F2
R
G2
=
TRIM CMRR WITH R
G1
HIGH INPUT RESISTANCE DOES NOT LOAD CABLE EVEN
WHEN POWER IS OFF
1/2
LT1229
R
F2
G – 1
R
G2
187Ω
R
F1
750Ω
R
G1
3.01k
–
+
1/2
LT1229 3.01k3.01k
12.1k
V
IN–
V
IN+
BNC INPUTS
Video Loop Through Amplifier Loop Through Amplifier Frequency
Response
FREQUENCY (Hz)
10
–60
GAIN (dB)
–50
–40
–30
–20
–10
10
100 1k 10k 100M
LT1229 • TA02
100k 1M 10M
0
COMMON MODE SIGNAL
NORMAL SIGNAL
, LTC and LT are registered trademarks of Linear Technology Corporation.
LT1229/LT1230
2
A
U
G
W
A
W
U
W
ARBSOLUTEXI T
IS
Supply Voltage ...................................................... ±18V
Input Current ......................................................±15mA
Output Short Circuit Duration (Note 2) .........Continuous
Operating Temperature Range
LT1229C, LT1230C ...............................0°C to 70°C
LT1229M, LT1230M (OBSOLETE).. –55°C to 125°C
Storage Temperature Range ..................–65°C to 150°C
Junction Temperature
Plastic Package ..............................................150°C
Ceramic Package (OBSOLETE) ................ 175°C
Lead Temperature (Soldering, 10 sec.).................300°C
ORDER PART
NUMBER
LT1230CN
LT1230CS
ORDER PART
NUMBER
S8 PART MARKING
LT1229CN8
LT1229CS8
1229
WU
U
PACKAGE/ORDER I FOR ATIO
S PACKAGE
14-LEAD PLASTIC SOIC
N PACKAGE
14-LEAD PLASTIC DIP
+
V
D
14
13
12
11
10
9
87
6
5
4
3
2
1OUT A
–IN A
+IN A
+IN B
–IN B
OUT B OUT C
V
–
–IN D
OUT D
TOP VIEW
A+IN D
+IN C
–IN C
C
B
8
7
6
54
3
2
1
+
–IN A
+IN A
V
TOP VIEW
N8 PACKAGE
8-LEAD PLASTIC DIP
OUT A
OUT B
–
V
–IN B
+IN B
A
B
S8 PACKAGE
8-LEAD PLASTIC SOIC
T
J MAX
= 150°C, θ
JA
= 100°C/W (N8)
T
J MAX
= 150°C, θ
JA
= 150°C/W (S8)
(Note 1)
T
J MAX
= 175°C, θ
JA
= 100°C/W (J8)
OBSOLETE PACKAGE
Consult LTC Marketing for parts specified with wider operating temperature ranges.
LT1229MJ8
LT1229CJ8
Consider the N Package for Alternate Source
LT1230MJ
LT1230CJ
T
J MAX
= 150°C, θ
JA
= 70°C/W (N)
T
J MAX
= 150°C, θ
JA
= 110°C/W (S)
OBSOLETE PACKAGE
Consider the N Package for Alternate Source
ORDER PART
NUMBER
J8 PACKAGE
8-LEAD CERAMIC DIP
T
J MAX
= 175°C, θ
JA
= 80°C/W (J)
J PACKAGE
14-LEAD CERAMIC DIP ORDER PART
NUMBER
3
LT1229/LT1230
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
V
OS
Input Offset Voltage T
A
= 25°C±3±10 mV
●±15 mV
Input Offset Voltage Drift ●10 µV/°C
I
IN+
Noninverting Input Current T
A
= 25°C±0.3 ±3µA
●±10 µA
I
IN–
Inverting Input Current T
A
= 25°C±10 ±50 µA
●±100 µA
e
n
Input Noise Voltage Density f = 1kHz, R
F
= 1k, R
G
= 10Ω, R
S
= 0Ω3.2 nV/√Hz
+i
n
Noninverting Input Noise Current Density f = 1kHz, R
F
= 1k, R
G
= 10Ω, R
S
= 10k 1.4 pA/√Hz
–in Inverting Input Noise Current Density f = 1kHz 32 pA/√Hz
R
IN
Input Resistance V
IN
= ±13V, V
S
= ±15V ●225 MΩ
V
IN
= ±3V, V
S
= ±5V ●225 MΩ
C
IN
Input Capacitance 3pF
Input Voltage Range V
S
= ±15V, T
A
= 25°C±13 ±13.5 V
●±12 V
V
S
= ±5V, T
A
= 25°C±3±3.5 V
●±2V
CMRR Common Mode Rejection Ratio V
S
= ±15V, V
CM
= ±13V, T
A
= 25°C5569dB
V
S
= ±15V, V
CM
= ±12V ●55 dB
V
S
= ±5V, V
CM
= ±3V, T
A
= 25°C5569dB
V
S
= ±5V, V
CM
= ±2V ●55 dB
Inverting Input Current V
S
= ±15V, V
CM
= ±13V, T
A
= 25°C 2.5 10 µA/V
Common Mode Rejection V
S
= ±15V, V
CM
= ±12V ●10 µA/V
V
S
= ±5V, V
CM
= ±3V, T
A
= 25°C 2.5 10 µA/V
V
S
= ±5V, V
CM
= ±2V ●10 µA/V
PSRR Power Supply Rejection Ratio V
S
= ±2V to ±15V, T
A
= 25°C6080dB
V
S
= ±3V to ±15V ●60 dB
Noninverting Input Current V
S
= ±2V to ±15V, T
A
= 25°C1050nA/V
Power Supply Rejection V
S
= ±3V to ±15V ●50 nA/V
Inverting Input Current V
S
= ±2V to ±15V, T
A
= 25°C 0.1 5 µA/V
Power Supply Rejection V
S
= ±3V to ±15V ●5µA/V
ELECTRICAL C CHARA TERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Each Amplifier, VCM = 0V, ±5V ≤ VS = ±15V, pulse tested unless
otherwise noted.
LT1229/LT1230
4
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. Each Amplifier, VCM = 0V, ±5V ≤ VS = ±15V, pulse tested unless
otherwise noted.
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
A
V
Large-Signal Voltage Gain, (Note 3) V
S
= ±15V, V
OUT
= ±10V, R
L
= 1k ●55 65 dB
V
S
= ±5V, V
OUT
= ±2V, R
L
= 150Ω●55 65 dB
R
OL
Transresistance, ∆V
OUT
/∆I
IN–
, (Note 3) V
S
= ±15V, V
OUT
= ±10V, R
L
= 1k ●100 200 kΩ
V
S
= ±5V, V
OUT
= ±2V, R
L
= 150Ω●100 200 kΩ
V
OUT
Maximum Output Voltage Swing, (Note 3) V
S
= ±15V, R
L
= 400Ω, T
A
= 25°C±12 ±13.5 V
●±10 V
V
S
= ±5V, R
L
= 150Ω, T
A
= 25°C±3±3.7 V
●±2.5 V
I
OUT
Maximum Output Current R
L
= 0Ω, T
A
= 25°C 30 65 125 mA
I
S
Supply Current, (Note 4) V
OUT
= 0V, Each Amplifier, T
A
= 25°C 6 9.5 mA
●11 mA
SR Slew Rate, (Notes 5 and 7) T
A
= 25°C 300 700 V/µs
SR Slew Rate V
S
= ±15V, R
F
= 750Ω, R
G
= 750Ω, R
L
= 400Ω2500 V/µs
t
r
Rise Time, (Notes 6 and 7) T
A
= 25°C1020ns
BW Small-Signal Bandwidth V
S
= ±15V, R
F
= 750Ω, R
G
= 750Ω, R
L
= 100Ω100 MHz
t
r
Small-Signal Rise Time V
S
= ±15V, R
F
= 750Ω, R
G
= 750Ω, R
L
= 100Ω3.5 ns
Propagation Delay V
S
= ±15V, R
F
= 750Ω, R
G
= 750Ω, R
L
= 100Ω3.5 ns
Small-Signal Overshoot V
S
= ±15V, R
F
= 750Ω, R
G
= 750Ω, R
L
= 100Ω15 %
t
s
Settling Time 0.1%, V
OUT
= 10V, R
F
=1k, R
G
= 1k, R
L
=1k 45 ns
Differential Gain, (Note 8) V
S
= ±15V, R
F
= 750Ω, R
G
= 750Ω, R
L
= 1k 0.01 %
Differential Phase, (Note 8) V
S
= ±15V, R
F
= 750Ω, R
G
= 750Ω, R
L
= 1k 0.01 Deg
Differential Gain, (Note 8) V
S
= ±15V, R
F
= 750Ω, R
G
= 750Ω, R
L
= 150Ω0.04 %
Differential Phase, (Note 8) V
S
= ±15V, R
F
= 750Ω, R
G
= 750Ω, R
L
= 150Ω0.1 Deg
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: A heat sink may be required depending on the power supply
voltage and how many amplifiers are shorted.
Note 3: The power tests done on ±15V supplies are done on only one
amplifier at a time to prevent excessive junction temperatures when testing
at maximum operating temperature.
Note 4: The supply current of the LT1229/LT1230 has a negative
temperature coefficient. For more information see the application
information section.
Note 5: Slew rate is measured at ±5V on a ±10V output signal while
operating on ±15V supplies with R
F
= 1k, R
G
= 110Ω and R
L
= 400Ω. The
slew rate is much higher when the input is overdriven and when the
amplifier is operated inverting, see the applications section.
Note 6: Rise time is measured from 10% to 90% on a ±500mV output
signal while operating on ±15V supplies with R
F
= 1k, R
G
= 110Ω and R
L
=
100Ω. This condition is not the fastest possible, however, it does
guarantee the internal capacitances are correct and it makes automatic
testing practical.
Note 7: AC parameters are 100% tested on the ceramic and plastic DIP
packaged parts (J and N suffix) and are sample tested on every lot of the
SO packaged parts (S suffix).
Note 8: NTSC composite video with an output level of 2V
P
.
ELECTRICAL C CHARA TERISTICS
5
LT1229/LT1230
CCHARA TERISTICS
UW
ATYPICALPER
FORCE
Voltage Gain and Phase vs –3dB Bandwidth vs Supply –3dB Bandwidth vs Supply
Frequency, Gain = 40dB Voltage, Gain = 100, RL = 100ΩVoltage, Gain = 100, RL = 1kΩ
Voltage Gain and Phase vs –3dB Bandwidth vs Supply –3dB Bandwidth vs Supply
Frequency, Gain = 20dB Voltage, Gain = 10, RL = 100ΩVoltage, Gain = 10, RL = 1k
Voltage Gain and Phase vs –3dB Bandwidth vs Supply –3dB Bandwidth vs Supply
Frequency, Gain = 6dB Voltage, Gain = 2, RL = 100ΩVoltage, Gain = 2, RL = 1k
SUPPLY VOLTAGE (±V)
2
–3dB BANDWIDTH (MHz)
40
100
120
12 16
LT1229 • TPC05
4068101418
0
20
60
140
160
180
R
F
= 500Ω
80
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
R
F
= 750Ω
R
F
= 1k
R
F
= 2k
R
F
= 250Ω
SUPPLY VOLTAGE (±V)
2
–3dB BANDWIDTH (MHz)
40
100
120
12 16
LT1229 • TPC06
4068101418
0
20
60
140
160
180
R
F
= 500Ω
80
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
R
F
= 750Ω
R
F
= 1k
R
F
= 2k
R
F
= 250Ω
SUPPLY VOLTAGE (±V)
2
–3dB BANDWIDTH (MHz)
4
10
12
12 16
LT1229 • TPC08
4068101418
0
2
6
14
16
18
R
F
= 500Ω
8R
F
= 1k
R
F
= 2k
SUPPLY VOLTAGE (±V)
2
–3dB BANDWIDTH (MHz)
4
10
12
12 16
LT1229 • TPC09
4068101418
0
2
6
14
16
18
R
F
= 500Ω
8
R
F
= 1k
R
F
= 2k
FREQUENCY (MHz)
0
VOLTAGE GAIN (dB)
2
4
6
8
0.1 10 100
LT1229 • TPC01
–2 1
7
5
3
1
–1
PHASE SHIFT (DEG)
180
90
0
45
135
225
PHASE
GAIN
V
S
= ±15V
R
L
= 100Ω
R
F
= 750Ω
SUPPLY VOLTAGE (±V)
2
–3dB BANDWIDTH (MHz)
40
100
120
12 16
LT1229 • TPC02
4068101418
0
20
60
140
160
180
R
F
= 500Ω
80
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
R
F
= 750Ω
R
F
= 1k
R
F
= 2k
SUPPLY VOLTAGE (±V)
2
–3dB BANDWIDTH (MHz)
40
100
120
12 16
LT1229 • TPC03
4068101418
0
20
60
140
160
180
80
PEAKING ≤ 0.5dB
PEAKING ≤ 5dB
R
F
= 750Ω
R
F
= 1k R
F
= 2k
R
F
= 500Ω
FREQUENCY (MHz)
14
VOLTAGE GAIN (dB)
16
18
20
22
0.1 10 100
LT1229 • TPC04
12 1
21
19
17
15
13
PHASE SHIFT (DEG)
180
90
0
45
135
225
PHASE
GAIN
V
S
= ±15V
R
L
= 100Ω
R
F
= 750Ω
FREQUENCY (MHz)
34
VOLTAGE GAIN (dB)
36
38
40
42
0.1 10 100
LT1229 • TPC07
32 1
41
39
37
35
33
PHASE SHIFT (DEG)
180
90
0
45
135
225
PHASE
GAIN
V
S
= ±15V
R
L
= 100Ω
R
F
= 750Ω
LT1229/LT1230
6
CCHARA TERISTICS
UW
ATYPICALPER
FORCE
Input Common Mode Limit vs Output Saturation Voltage vs Output Short-Circuit Current vs
Temperature Temperature Junction Temperature
Maximum Capacitance Load vs Total Harmonic Distortion vs 2nd and 3rd Harmonic
Feedback Resistor Frequency Distortion vs Frequency
FREQUENCY (Hz)
TOTAL HARMONIC DISTORTION (%)
0.01
0.10
10 1k 10k 100k
LT1229 • TPC11
0.001 100
VS = ±15V
RL = 400Ω
RF = RG = 750Ω
VO = 7VRMS
VO = 1VRMS
TEMPERATURE (°C)
–25
OUTPUT SHORT CIRCUIT CURRENT (mA)
40
60
100 150
LT1229 • TPC15
0–50 25 50 75 125 175
30
70
50
FREQUENCY (Hz)
10
1
10
100
1k 100k
LT1229 • TPC16
100 10k
SPOT NOISE (nV/√Hz OR pA/√Hz)
–in
en
+in
FREQUENCY (Hz)
OUTPUT IMPEDANCE (Ω)
0.1
100
10k 1M 10M 100M
LT1229 • TPC18
0.001 100k
0.01
10
VS = ±15V
1.0 RF = RG = 2k
RF = RG = 750Ω
FEEDBACK RESISTOR (kΩ)
10
CAPACITIVE LOAD (pF)
100
1000
10000
023
LT1229 • TPC10
11
V
S
= ±5V
V
S
= ±15V
R
L
= 1k
PEAKING ≤ 5dB
GAIN = 2
FREQUENCY (MHz)
1
–70
DISTORTION (dBc)
–60
–50
–40
–30
–20
10 100
LT1229 • TPC12
V
S
= ±15V
V
O
= 2V
P-P
R
L
= 100Ω
R
F
= 750Ω
A
V
= 10dB 2ND
3RD
TEMPERATURE (°C)
COMMON MODE RANGE (V)
2.0
V
+
–50 25 75 125
LT1229 • TPC13
V
–
0
1.0
–1.0
–2.0
–0.5
–1.5
1.5
0.5
–25 50 100
V
+
= 2V TO 18V
V
–
= –2V TO –18V
TEMPERATURE (°C)
OUTPUT SATURATION VOLTAGE (V)
V
+
–50 25 75 125
LT1229 • TPC14
V
–
0
1.0
–1.0
–0.5
0.5
–25 50 100
R
L
=
∞
±2V ≤ V
S
≤ ±18V
Spot Noise Voltage and Current vs Power Supply Rejection vs Output Impedance vs
Frequency Frequency Frequency
FREQUENCY (Hz)
POWER SUPPLY REJECTION (dB)
40
80
10k 1M 10M 100M
LT1229 • TPC17
0100k
V
S
= ±15V
R
L
= 100Ω
R
F
= R
G
= 750Ω
NEGATIVE
20
60
POSITIVE
7
LT1229/LT1230
Settling Time to 10mV vs Settling Time to 1mV vs
Output Step Output Step Supply Current vs Supply Voltage
W
I
SPL
II
FED S
W
A
CHETIC
CCHARA TERISTICS
UW
ATYPICALPER
FORCE
One Amplifier
SUPPLY VOLTAGE (±V)
SUPPLY CURRENT (mA)
12
LT1229 • TPC21
40816
0
10
5
1
2
3
4
6
7
8
9
2 6 10 14 18
–55°C
25°C
125°C
175°C
SETTLING TIME (ns)
OUTPUT STEP (V)
60
LT1229 • TPC19
200 40 80 100
–10
10
0
–8
–6
–4
–2
2
4
6
8NONINVERTING
INVERTING
V
S
= ±15V
R
F
= R
G
= 1k
INVERTING
NONINVERTING
SETTLING TIME (µs)
OUTPUT STEP (V)
12
LT1229 • TPC20
40 8 16 20
–10
10
0
–8
–6
–4
–2
2
4
6
8NONINVERTING
INVERTING
V
S
= ±15V
R
F
= R
G
= 1k
NONINVERTING
INVERTING
LT1229 • TA03
+IN –IN VOUT
V+
V–
LT1229/LT1230
8
limited by the gain bandwidth product of about 1GHz. The
curves show that the bandwidth at a closed-loop gain of
100 is 10MHz, only one tenth what it is at a gain of two.
Capacitance on the Inverting Input
Current feedback amplifiers want resistive feedback from
the output to the inverting input for stable operation. Take
care to minimize the stray capacitance between the output
and the inverting input. Capacitance on the inverting input
to ground will cause peaking in the frequency response
(and overshoot in the transient response), but it does not
degrade the stability of the amplifier. The amount of
capacitance that is necessary to cause peaking is a func-
tion of the closed-loop gain taken. The higher the gain, the
more capacitance is required to cause peaking. We can
add capacitance from the inverting input to ground to
increase the bandwidth in high gain applications. For
example, in this gain of 100 application, the bandwidth can
be increased from 10MHz to 17MHz by adding a 2200pF
capacitor.
LT1229 • TA05
–
+
CGRG
5.1Ω
RF
510Ω
VOUT
1/2
LT1229
VIN
Boosting Bandwidth of High Gain Amplifier with
Capacitance on Inverting Input
FREQUENCY (MHz)
1
19
GAIN (dB)
22
25
28
31
46
49
10 100
LT1229 • TA06
34
37
40
43 C
G
= 4700pF
C
G
= 2200pF
C
G
= 0
U
S
A
O
PPLICATI
WU
U
I FOR ATIO
The LT1229/LT1230 are very fast dual and quad current
feedback amplifiers. Because they are current feedback
amplifiers, they maintain their wide bandwidth over a wide
range of voltage gains. These amplifiers are designed to
drive low impedance loads such as cables with excellent
linearity at high frequencies.
Feedback Resistor Selection
The small-signal bandwidth of the LT1229/LT1230 is set
by the external feedback resistors and the internal junction
capacitors. As a result, the bandwidth is a function of the
supply voltage, the value of the feedback resistor, the
closed-loop gain and load resistor. The characteristic
curves of Bandwidth versus Supply Voltage are done with
a heavy load (100Ω) and a light load (1k) to show the effect
of loading. These graphs also show the family of curves
that result from various values of the feedback resistor.
These curves use a solid line when the response has less
than 0.5dB of peaking and a dashed line when the re-
sponse has 0.5dB to 5dB of peaking. The curves stop
where the response has more than 5dB of peaking.
Small-Signal Rise Time with
RF = RG = 750Ω, VS = ±15V, and RL = 100Ω
LT1229 • TA04
At a gain of two, on ±15V supplies with a 750Ω feedback
resistor, the bandwidth into a light load is over 160MHz
without peaking, but into a heavy load the bandwidth
reduces to 100MHz. The loading has so much effect
because there is a mild resonance in the output stage that
enhances the bandwidth at light loads but has its Q
reduced by the heavy load. This enhancement is only
useful at low gain settings; at a gain of ten it does not boost
the bandwidth. At unity gain, the enhancement is so
effective the value of the feedback resistor has very little
effect. At very high closed-loop gains, the bandwidth is
9
LT1229/LT1230
amplifier at 150°C is less than 7mA and typically is only
4.5mA. The power in the IC due to the load is a function of
the output voltage, the supply voltage and load resistance.
The worst case occurs when the output voltage is at half
supply, if it can go that far, or its maximum value if it
cannot reach half supply.
For example, let’s calculate the worst case power dissipa-
tion in a video cable driver operating on ±12V supplies that
delivers a maximum of 2V into 150Ω.
Capacitive Loads
The LT1229/LT1230 can drive capacitive loads directly
when the proper value of feedback resistor is used. The
graph Maximum Capacitive Load vs Feedback Resistor
should be used to select the appropriate value. The value
shown is for 5dB peaking when driving a 1k load at a gain
of 2. This is a worst case condition; the amplifier is more
stable at higher gains and driving heavier loads. Alterna-
tively, a small resistor (10Ω to 20Ω) can be put in series
with the output to isolate the capacitive load from the
amplifier output. This has the advantage that the amplifier
bandwidth is only reduced when the capacitive load is
present, and the disadvantage that the gain is a function of
the load resistance.
Power Supplies
The LT1229/LT1230 amplifiers will operate from single or
split supplies from ±2V (4V total) to ±15V (30V total). It is
not necessary to use equal value split supplies, however,
the offset voltage and inverting input bias current will
change. The offset voltage changes about 350µV per volt
of supply mismatch, the inverting bias current changes
about 2.5µA per volt of supply mismatch.
Power Dissipation
The LT1229/LT1230 amplifiers combine high speed and
large output current drive into very small packages. Be-
cause these amplifiers work over a very wide supply range,
it is possible to exceed the maximum junction temperature
under certain conditions. To ensure that the LT1229 and
LT1230 remain within their absolute maximum ratings,
we must calculate the worst case power dissipation,
define the maximum ambient temperature, select the
appropriate package and then calculate the maximum
junction temperature.
The worst case amplifier power dissipation is the total of
the quiescent current times the total power supply voltage
plus the power in the IC due to the load. The quiescent
supply current of the LT1229/LT1230 has a strong nega-
tive temperature coefficient. The supply current of each
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Now if that is the dual LT1229, the total power in the
package is twice that, or 0.602W. We now must calcu-
late how much the die temperature will rise above the
ambient. The total power dissipation times the thermal
resistance of the package gives the amount of tempera-
ture rise. For the above example, if we use the SO8
surface mount package, the thermal resistance is
150°C/W junction to ambient in still air.
Temperature Rise = P
d (MAX)
R
θJA
= 0.602W •
150°C/W = 90.3°C
The maximum junction temperature allowed in the plastic
package is 150°C. Therefore, the maximum ambient al-
lowed is the maximum junction temperature less the
temperature rise.
Maximum Ambient = 150°C – 90.3°C = 59.7°C
Note that this is less than the maximum of 70°C that is
specified in the absolute maximum data listing. If we must
use this package at the maximum ambient we must lower
the supply voltage or reduce the output swing.
As a guideline to help in the selection of the LT1229/
LT1230 the following table describes the maximum sup-
ply voltage that can be used with each part in cable driving
applications.
PVI VV
V
R
PVmAVV
V
W per Amp
d MAX SS MAX SO MAX
O MAX
L
d MAX
() () () ()
()
=+
=+
()
=+=
2
2 12 7 12 2 2
150
0 168 0 133 0 301
–
•• –•
... Ω
LT1229/LT1230
10
Assumptions:
1. The maximum ambient is 70°C for the commercial
parts (C suffix) and 125°C for the full temperature
parts (M suffix).
2. The load is a double-terminated video cable, 150Ω.
3. The maximum output voltage is 2V (peak or DC).
4. The thermal resistance of each package:
J8 is 100°C/W J is 80°/W
N8 is 100°C/W N is 70°/W
S8 is 150°C/W S is 110°/W
Maximum Supply Voltage for 75Ω Cable Driving Applications at
Maximum Ambient Temperature
PART PACKAGE MAX POWER AT T
A
MAX SUPPLY
LT1229MJ8 Ceramic DIP 0.500W at 125°CV
S
< ±10.1
LT1229CJ8 Ceramic DIP 1.050W at 70°CV
S
< ±18.0
LT1229CN8 Plastic DIP 0.800W at 70°CV
S
< ±15.6
LT1229CS8 Plastic SO8 0.533W at 70°CV
S
< ±10.6
LT1230MJ Ceramic DIP 0.625W at 125°CV
S
< ±6.6
LT1230CJ Ceramic DIP 1.313W at 70°CV
S
< ±13.0
LT1230CN Plastic DIP 1.143W at 70°CV
S
< ±11.4
LT1230CS Plastic SO14 0.727W at 70°CV
S
< ±7.6
Slew Rate
The slew rate of a current feedback amplifier is not
independent of the amplifier gain the way it is in a tradi-
tional op amp. This is because the input stage and the
output stage both have slew rate limitations. The input
stage of the LT1229/LT1230 amplifiers slew at about
100V/µs before they become nonlinear. Faster input sig-
nals will turn on the normally reverse-biased emitters on
the input transistors and enhance the slew rate signifi-
cantly. This enhanced slew rate can be as much as
2500V/µs.
The output slew rate is set by the value of the feedback
resistors and the internal capacitance. At a gain of ten with
a 1k feedback resistor and ±15V supplies, the output slew
rate is typically 700V/µs and –1000V/µs. There
is no input stage enhancement because of the high gain.
Large-Signal Response, AV = 2, RF = RG = 750Ω
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LT1229 • TA07
Settling Time
The characteristic curves show that the LT1229/LT1230
amplifiers settle to within 10mV of final value in 40ns to
55ns for any output step up to 10V. The curve of settling
to 1mV of final value shows that there is a slower thermal
contribution up to 20µs. The thermal settling component
comes from the output and the input stage. The output
contributes just under 1mV per volt of output change and
the input contributes 300µV per volt of input change.
Fortunately, the input thermal tends to cancel the output
thermal. For this reason the noninverting gain of two
configurations settles faster than the inverting gain of one.
Larger feedback resistors will reduce the slew rate as will
lower supply voltages, similar to the way the bandwidth is
reduced.
Large-Signal Response, AV = 10, RF = 1k, RG = 110Ω
LT1229 • TA08
11
LT1229/LT1230
Crosstalk and Cascaded Amplifiers
The amplifiers in the LT1229/LT1230 do not share any
common circuitry. The only thing the amplifiers share is
the supplies. As a result, the crosstalk between amplifiers
is very low. In a good breadboard or with a good PC board
layout the crosstalk from the output of one amplifier to the
input of another will be over 100dB down, up to 100kHz
and 65dB down at 10MHz. The following curve shows
the crosstalk from the output of one amplifier to the
input of another.
Amplifier Crosstalk vs Frequency
FREQUENCY (Hz)
10
50
OUTPUT TO INPUT CROSSTALK (dB)
60
70
80
90
100
120
100 1k 10k 100M
LT1229 • TA12
100k 1M 10M
110
V
S
= ±15V
A
V
= 10
R
S
= 50Ω
R
L
= 100Ω
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The high frequency crosstalk between amplifiers is
caused by magnetic coupling between the internal wire
bonds that connect the IC chip to the package lead frame.
The amount of crosstalk is inversely proportional to the
load resistor the amplifier is driving, with no load (just
the feedback resistor) the crosstalk improves 18dB. The
curve shows the crosstalk of the LT1229 amplifier B
output (Pin 7) to the input of amplifier A. The crosstalk
from amplifier A’s output (Pin 1) to amplifier B is about
10dB better. The crosstalk between all of the LT1230
amplifiers is as shown. The LT1230 amplifiers that are
separated by the supplies are a few dB better.
When cascading amplifiers the crosstalk will limit the
amount of high frequency gain that is available because
the crosstalk signal is out of phase with the input signal.
This will often show up as unusual frequency response.
For example: cascading the two amplifiers in the LT1229,
each set up with 20dB of gain and a –3dB bandwidth of
65MHz into 100Ω will result in 40dB of gain, BUT the
response will start to drop at about 10MHz and then flatten
out from 20MHz to 30MHz at about 0.5dB down. This is
due to the crosstalk back to the input of the first amplifier.
For best results when cascading amplifiers use the LT1229
and drive amplifier B and follow it with amplifier A.
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Single 5V Supply Cable Driver for Composite Video
This circuit amplifies standard 1V peak composite video
input (1.4V
P-P
) by two and drives an AC coupled, doubly
terminated cable. In order for the output to swing
2.8V
P-P
on a single 5V supply, it must be biased accu-
rately. The average DC level of the composite input is a
function of the luminance signal. This will cause problems
if we AC couple the input signal into the amplifier because
a rapid change in luminance will drive the output into the
rails. To prevent this we must establish the DC level at the
input and operate the amplifier with DC gain.
The transistor’s base is biased by R1 and R2 at 2V. The
emitter of the transistor clamps the noninverting input of
the amplifier to 1.4V at the most negative part of the input
(the sync pulses). R4, R5 and R6 set the amplifier up with
a gain of two and bias the output so the bottom of the sync
pulses are at 1.1V. The maximum input then drives the
output to 3.9V.
LT1229 • TA11
–
+
1/2
LT1229
V
OUT
R3
150k
R2
2k
V
IN
R5
750Ω
C1
1µF
R8
10k
R1
3k
C2
1µF
R4
1.5k
2N3904
5V
C3
47µF
R6
510Ω
R7
75Ω
C4
1000µF
+
+
LT1229/LT1230
12
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PACKAGE DESCRIPTIO
J8 Package
8-Lead CERDIP (Narrow .300 Inch, Hermetic)
(Reference LTC DWG # 05-08-1110)
J Package
14-Lead CERDIP (Narrow .300 Inch, Hermetic)
(Reference LTC DWG # 05-08-1110)
OBSOLETE PACKAGES
J8 1298
0.014 – 0.026
(0.360 – 0.660)
0.200
(5.080)
MAX
0.015 – 0.060
(0.381 – 1.524)
0.125
3.175
MIN
0.100
(2.54)
BSC
0.300 BSC
(0.762 BSC)
0.008 – 0.018
(0.203 – 0.457) 0° – 15°
0.005
(0.127)
MIN
0.405
(10.287)
MAX
0.220 – 0.310
(5.588 – 7.874)
1234
8765
0.025
(0.635)
RAD TYP
0.045 – 0.068
(1.143 – 1.727)
FULL LEAD
OPTION
0.023 – 0.045
(0.584 – 1.143)
HALF LEAD
OPTION
CORNER LEADS OPTION
(4 PLCS)
0.045 – 0.065
(1.143 – 1.651)
NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP/PLATE
OR TIN PLATE LEADS
J14 1298
0.045 – 0.065
(1.143 – 1.651)
0.100
(2.54)
BSC
0.014 – 0.026
(0.360 – 0.660)
0.200
(5.080)
MAX
0.015 – 0.060
(0.381 – 1.524)
0.125
(3.175)
MIN
0.300 BSC
(0.762 BSC)
0.008 – 0.018
(0.203 – 0.457) 0° – 15°
1234567
0.220 – 0.310
(5.588 – 7.874)
0.785
(19.939)
MAX
0.005
(0.127)
MIN 14 11 891013 12
0.025
(0.635)
RAD TYP
NOTE: LEAD DIMENSIONS APPLY TO SOLDER DIP/PLATE
OR TIN PLATE LEADS
13
LT1229/LT1230
N8 Package
8-Lead PDIP (Narrow .300 Inch)
(Reference LTC DWG # 05-08-1510)
S8 Package
8-Lead Plastic Small Outline (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1610)
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PACKAGE DESCRIPTIO
N8 1098
0.100
(2.54)
BSC
0.065
(1.651)
TYP
0.045 – 0.065
(1.143 – 1.651)
0.130 ± 0.005
(3.302 ± 0.127)
0.020
(0.508)
MIN
0.018 ± 0.003
(0.457 ± 0.076)
0.125
(3.175)
MIN
12 34
8765
0.255 ± 0.015*
(6.477 ± 0.381)
0.400*
(10.160)
MAX
0.009 – 0.015
(0.229 – 0.381)
0.300 – 0.325
(7.620 – 8.255)
0.325 +0.035
–0.015
+0.889
–0.381
8.255
()
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
0.016 – 0.050
(0.406 – 1.270)
0.010 – 0.020
(0.254 – 0.508)× 45°
0°– 8° TYP
0.008 – 0.010
(0.203 – 0.254)
SO8 1298
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
TYP
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
1234
0.150 – 0.157**
(3.810 – 3.988)
8765
0.189 – 0.197*
(4.801 – 5.004)
0.228 – 0.244
(5.791 – 6.197)
DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
*
**
LT1229/LT1230
14
S Package
14-Lead Plastic Small Outline (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1610)
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PACKAGE DESCRIPTIO
1234
0.150 – 0.157**
(3.810 – 3.988)
14 13
0.337 – 0.344*
(8.560 – 8.738)
0.228 – 0.244
(5.791 – 6.197)
12 11 10 9
567
8
0.016 – 0.050
(0.406 – 1.270)
0.010 – 0.020
(0.254 – 0.508)× 45°
0° – 8° TYP
0.008 – 0.010
(0.203 – 0.254)
S14 1298
0.053 – 0.069
(1.346 – 1.752)
0.014 – 0.019
(0.355 – 0.483)
TYP
0.004 – 0.010
(0.101 – 0.254)
0.050
(1.270)
BSC
DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
*
**
15
LT1229/LT1230
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
N Package
14-Lead PDIP (Narrow .300 Inch)
(Reference LTC DWG # 05-08-1510)
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PACKAGE DESCRIPTIO
N14 1098
0.020
(0.508)
MIN
0.125
(3.175)
MIN
0.130 ± 0.005
(3.302 ± 0.127)
0.045 – 0.065
(1.143 – 1.651)
0.065
(1.651)
TYP
0.018 ± 0.003
(0.457 ± 0.076)
0.100
(2.54)
BSC
0.005
(0.125)
MIN
0.255 ± 0.015*
(6.477 ± 0.381)
0.770*
(19.558)
MAX
31 24567
8910
11
1213
14
0.009 – 0.015
(0.229 – 0.381)
0.300 – 0.325
(7.620 – 8.255)
0.325 +0.035
–0.015
+0.889
–0.381
8.255
()
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
LT1229/LT1230
16 LINEAR TECHNOLOGY CORPORATION 1992
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507
●
www.linear.com
122930fb LT/CP 0801 1.5K REV B • PRINTED IN USA
RELATED PARTS
PART NUMBER DESCRIPTION COMMENTS
LT1227 Single 140MHz CFA Single Version of the LT1229
LT1395/LT1396/LT1397 Single/Dual/Quad 400MHz CFA SOT-23, MSOP-8 and SSOP-16 Packaging
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Single Supply AC Coupled Amplifiers
Noninverting Inverting
LT1229 • TA10
–
+
+
1/2
LT1229
5V 4.7µF
0.1µF
510Ω
V
OUT
10kΩ
10kΩ
V
IN
A
V
= 10
BW = 600Hz TO 50MHz
51Ω
R
S
+
510Ω
R
S
+ 51Ω
4.7µF
LT1229 • TA09
–
+
+
1/2
LT1229
5V 4.7µF
0.1µF
510Ω
51Ω
V
OUT
10k
10k
V
IN
A
V
= 11
BW = 600Hz TO 50MHz
+
4.7µF
