SS/ON1
SS/ON2
LM2642
PGOOD1
H
H
L
L
UV_Delay VOUT1
1.3V-27V
VOUT2
1.3V-27V
VIN
4.5V-30V
LM2642
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SNVS203I –MAY 2002–REVISED APRIL 2013
LM2642 Two-Phase Synchronous Step-Down Switching Controller
Check for Samples: LM2642
1FEATURES DESCRIPTION
The LM2642 consists of two current mode
2• Two Synchronous Buck Regulators synchronous buck regulator controllers with a
• 180° Out of Phase Operation switching frequency of 300kHz.
• 4.5V to 30V Input Range The two switching regulator controllers operate 180°
• Power Good Function Monitors Ch.1 out of phase. This feature reduces the input ripple
• 37µA Shutdown Current RMS current, thereby significantly reducing the
required input capacitance. The two switching
• 0.04% (typical) Line and Load Regulation Error regulator outputs can also be paralleled to operate as
• Current Mode Control With or Without a Sense a dual-phase single output regulator.
Resistor The output of each channel can be independently
• Independent Enable/Soft-start Pins Allow adjusted from 1.3 to VIN• maximum duty cycle. An
Simple Sequential Startup Configuration. internal 5V rail is also available externally for driving
• Configurable for Single Output Parallel bootstrap circuitry.
Operation. (See Figure 3). Current-mode feedback control assures excellent line
• Adjustable Cycle-by-Cycle Current Limit and load regulation and a wide loop bandwidth for
• Input Under-voltage Lockout excellent response to fast load transients. Current is
sensed across either the Vds of the top FET or
• Output Over-voltage Latch Protection across an external current-sense resistor connected
• Output Under-voltage Protection with Delay in series with the drain of the top FET. Current limit is
• Thermal Shutdown independently adjustable for each channel.
• Self Discharge of Output Capacitors When the The LM2642 features analog soft-start circuitry that is
Regulator is OFF independent of the output load and output
capacitance. This makes the soft-start behavior more
• TSSOP package predictable and controllable than traditional soft-start
circuits.
APPLICATIONS A PGOOD1 pin is provided to monitor the dc output
• Embedded Computer Systems of channel 1. Over-voltage protection is available for
• High End Gaming Systems both outputs. A UV-Delay pin is also available to
• Set-top Boxes allow delayed shut off time for the IC during an output
under-voltage event.
• WebPAD
BLOCK DIAGRAM
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2002–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
RSNS1
SW1
HDRV1
CBOOT1
VDD1
LDRV1
VIN
PGND
LDRV2
VDD2
CBOOT2
HDRV2
SW2
RSNS2KS2
ILIM2
COMP2
FB2
ON/SS2
ON/SS1
SGND
VLIN5
UVDELAY
PGOOD1
FB1
COMP1
ILIM1
KS1 28
27
26
25
24
23
22
21
20
19
18
17
16
1514
13
12
11
10
9
8
7
6
5
4
3
2
1
LM2642
SNVS203I –MAY 2002–REVISED APRIL 2013
www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
CONNECTION DIAGRAM
TOP VIEW
Figure 1. 28-Lead TSSOP
PIN DESCRIPTIONS
KS1 (Pin 1) The positive (+) Kelvin sense for the internal current sense amplifier of Channel 1. Use a separate
trace to connect this pin to the current sense point. It should be connected to VIN as close as possible to
the node of the current sense resistor. When no current-sense resistor is used, connect as close as
possible to the drain node of the upper MOSFET.
ILIM1 (Pin 2) Current limit threshold setting for Channel 1. It sinks a constant current of 10 µA, which is
converted to a voltage across a resistor connected from this pin to VIN. The voltage across the resistor is
compared with either the VDS of the top MOSFET or the voltage across the external current sense
resistor to determine if an over-current condition has occurred in Channel 1.
COMP1 (Pin 3)Compensation pin for Channel 1. This is the output of the internal transconductance amplifier.
The compensation network should be connected between this pin and the signal ground, SGND (Pin 8).
FB1 (Pin 4) Feedback input for channel 1. Connect to VOUT through a voltage divider to set the channel 1
output voltage.
PGOOD1 (Pin 5) An open-drain power-good output for Channel 1. It is 'LOW' (low impedance to ground)
whenever the output voltage of Channel 1 falls outside of a +15% to -9% window. PGOOD1 stays latched
in a 'LOW' state during OVP or UVP on either channel. It will recover to a 'HIGH' state (high impedance to
ground) after a Channel 1 output under-voltage event (<91%) when the output returns to within 6% of its
nominal value. See Operation Descriptions for details.
UV_DELAY (Pin 6) A capacitor from this pin to ground sets the delay time for UVP. The capacitor is charged
from a 5µA current source. When UV_DELAY charges to 2.3V (typical), the system immediately latches
off. Connecting this pin to ground will disable the output under-voltage protection.
VLIN5 (Pin 7)The output of an internal 5V LDO regulator derived from VIN. It supplies the internal bias for the
chip and supplies the bootstrap circuitry for gate drive. Bypass this pin to signal ground with a minimum of
4.7µF capacitor.
SGND (Pin 8)The ground connection for the signal-level circuitry. It should be connected to the ground rail of the
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system.
ON/SS1 (Pin 9) Channel 1 enable pin. This pin is internally pulled up to one diode drop above VLIN5. Pulling
this pin below 1.2V (open-collector type) turns off Channel 1. If both ON/SS1 and ON/SS2 pins are pulled
below 1.2V, the whole chip goes into shut down mode. Adding a capacitor to this pin provides a soft-start
feature that minimizes inrush current and output voltage overshoot.
ON/SS2 (Pin 10) Channel 2 enable pin. See the description for Pin 9, ON/SS1. May be connected to ON/SS1 for
simultaneous startup or for parallel operation.
FB2 (Pin 11)Feedback input for channel 2. Connect to VOUT through a voltage divider to set the Channel 2
output voltage.
COMP2 (Pin 12) Compensation pin for Channel 2. This is the output of the internal transconductance amplifier.
The compensation network should be connected between this pin and the signal ground SGND (Pin 8).
ILIM2 (Pin 13) Current limit threshold setting for Channel 2. See ILIM1 (Pin 2).
KS2 (Pin 14)The positive (+) Kelvin sense for the internal current sense amplifier of Channel 2. See KS1 (Pin 1).
RSNS2 (Pin 15) The negative (-) Kelvin sense for the internal current sense amplifier of Channel 2. Connect this
pin to the low side of the current sense resistor that is placed between VIN and the drain of the top
MOSFET. When the Rds of the top MOSFET is used for current sensing, connect this pin to the source of
the top MOSFET. Always use a separate trace to form a Kelvin connection to this pin.
SW2 (Pin 16)Switch-node connection for Channel 2, which is connected to the source of the top MOSFET of
Channel 2. It serves as the negative supply rail for the top-side gate driver, HDRV2.
HDRV2 (Pin 17)Top-side gate-drive output for Channel 2. HDRV is a floating drive output that rides on the
corresponding switching-node voltage.
CBOOT2 (Pin 18) Bootstrap capacitor connection. It serves as the positive supply rail for the Channel 2 top-side
gate drive. Connect this pin to VDD2 (Pin 19) through a diode, and connect the low side of the bootstrap
capacitor to SW2 (Pin16).
VDD2 (Pin 19) The supply rail for the Channel 2 low-side gate drive. Connected to VLIN5 (Pin 7) through a 4.7Ω
resistor and bypassed to power ground with a ceramic capacitor of at least 1µF. Tie this pin to VDD1 (Pin
24).
LDRV2 (Pin 20) Low-side gate-drive output for Channel 2.
PGND (Pin 21) The power ground connection for both channels. Connect to the ground rail of the system.
VIN (Pin 22) The power input pin for the chip. Connect to the positive (+) input rail of the system. This pin must
be connected to the same voltage rail as the top FET drain (or the current sense resistor when used).
LDRV1 (Pin 23) Low-side gate-drive output for Channel 1.
VDD1 (Pin 24)The supply rail for Channel 1 low-side gate drive. Tie this pin to VDD2 (Pin 19).
CBOOT1 (Pin 25)Bootstrap capacitor connection. It serves as the positive supply rail for Channel 1 top-side gate
drive. See CBOOT2 (Pin 18).
HDRV1 (Pin 26)Top-side gate-drive output for Channel 1. See HDRV2 (Pin 17).
SW1 (Pin 27) Switch-node connection for Channel 1. See SW2 (Pin16).
RSNS1 (Pin 28)The negative (-) Kelvin sense for the internal current sense amplifier of Channel 1. See RSNS2
(Pin 15).
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ABSOLUTE MAXIMUM RATINGS (1)
Voltages from the indicated pins to SGND/PGND:
VIN, ILIM1, ILIM2, KS1, KS2 −0.3V to 32V
SW1, SW2, RSNS1, RSNS2 −0.3 to (VIN + 0.3)V
FB1, FB2, VDD1, VDD2 −0.3V to 6V
PGOOD, COMP1, COMP2, UV Delay −0.3V to (VLIN5 +0.3)V
ON/SS1, ON/SS2 (2) −0.3V to (VLIN5 +0.6)V
CBOOT1 to SW1, CBOOT2 to SW2 −0.3V to 7V
LDRV1, LDRV2 −0.3V to (VDD+0.3)V
HDRV1 to SW1, HDRV2 to SW2 −0.3V
HDRV1 to CBOOT1, HDRV2 to CBOOT2 +0.3V
Power Dissipation (TA= 25°C),
(3) 1.1W
Ambient Storage Temperature Range −65°C to +150°C
Soldering Dwell Time, Temperature (4)
Wave 4 sec, 260°C
Infrared 10sec, 240°C
Vapor Phase 75sec, 219°C
ESD Rating (5) 2kV
(1) Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Range indicates conditions for
which the device is intended to be functional, but does not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics. The specifications apply only for the test conditions. Some performance characteristics
may degrade when the device is not operated under the listed test conditions.
(2) ON/SS1 and ON/SS2 are internally pulled up to one diode drop above VLIN5. Do not apply an external pull-up voltage to these pins. It
may cause damage to the IC.
(3) The maximum allowable power dissipation is calculated by using PDMAX = (TJMAX - TA)/θJA, where TJMAX is the maximum junction
temperature, TAis the ambient temperature and θJA is the junction-to-ambient thermal resistance of the specified package. The 1.1W
rating results from using 125°C, 25°C, and 90.6°C/W for TJMAX, TA, and θJA respectively. A θJA of 90.6°C/W represents the worst-case
condition of no heat sinking of the 28-pin TSSOP. A thermal shutdown will occur if the temperature exceeds the maximum junction
temperature of the device.
(4) For detailed information on soldering plastic small-outline packages, see the TI website at www.ti.com/packaging.
(5) For testing purposes, ESD was applied using the human-body model, a 100pF capacitor discharged through a 1.5kΩresistor.
OPERATING RATINGS (1)
VIN (VLIN5 tied to VIN) 4.5V to 5.5V
VIN (VIN and VLIN5 separate) 5.5V to 30V
Junction Temperature −40°C to +125°C
(1) Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Range indicates conditions for
which the device is intended to be functional, but does not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics. The specifications apply only for the test conditions. Some performance characteristics
may degrade when the device is not operated under the listed test conditions.
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ELECTRICAL CHARACTERISTICS
Unless otherwise specified, VIN = 15V, GND = PGND = 0V, VLIN5 = VDD1 = VDD2. Limits appearing in boldface type apply
over the specified operating junction temperature range, (-20°C to +125°C, if not otherwise specified). Specifications
appearing in plain type are measured using low duty cycle pulse testing with TA= 25°C (1),(2). Min/Max limits are specified by
design, test, or statistical analysis.
Symbol Parameter Conditions Min Typ Max Units
System
ΔVOUT/VOUT Load Regulation VIN = 15V, Vcompx = 0.5V to 1.5V 0.04 %
ΔVOUT/VOUT Line Regulation 5.5V ≤VIN ≤30V, Vcompx =1.25V 0.04 %
VFB1_FI2 Feedback Voltage 5.5V ≤VIN ≤30V 1.215 1.238 1.260
0°C to 125°C 1.217 1.259 V
-40°C to 125°C 1.212 1.261
IVIN Input Supply Current VON_SSx > 2V 1.0 mA
5.5V ≤VIN ≤30V 2.0
Shutdown (3) 37 µA
VON_SS1 = VON_SS2= 0V 110
VLIN5 VLIN5 Output Voltage (4) IVLIN5 = 0 to 25mA, 4.70 55.30
5.5V ≤VIN ≤30V V
-40°C to 125°C 4.68 5.30
VCLos Current Limit Comparator ±2 ±7.0 mV
Offset (VILIMX −VRSNSX)
ICL Current Limit Sink Current 910 11 µA
-40°C to 125°C 8.67 11
Iss_SC1, Soft-Start Source Current VON_ss1 = VON_ss2 = 1.5V (on) 0.5 25.0 µA
Iss_SC2
Iss_SK1, Soft-Start Sink Current VON_ss1 = VON_ss2 = 2V 25.2 10 µA
Iss_SK2
VON_SS1, Soft-Start On Threshold 0.7 1.12 1.4 V
VON_SS2
VSSTO Soft-Start Timeout (5) 3.3 V
Threshold
Isc_uvdelay UV_DELAY Source Current UV-DELAY = 2V 259µA
Isk_uvdelay UV_DELAY Sink Current UV-DELAY = 0.4V 0.2 0.48 1.2 mA
VUVDelay UV_DELAY Threshold 2.3 V
Voltage
VUVP FB1, FB2, Under Voltage As a percentage of nominal output voltage 75 80 86 %
Protection Latch Threshold (falling edge)
Hysteresis 4 %
VOVP VOUT Overvoltage As a percentage measured at VFB1, VFB2 107 113 122 %
Shutdown Latch Threshold
Vpwrbad Regulator Window Detector As a percentage of output voltage
Thresholds (PGOOD1 from 86.5 90.3 94.5 %
High to Low)
Vpwrgd Regulator Window Detector
Thresholds (PGOOD1 from 91.5 94 97.0 %
Low to High)
Swx_R SW1, SW2 ON-Resistance VSW1 = VSW2 = 2V 420 480 535 Ω
Gate Drive
ICBOOT CBOOTx Leakage Current VCBOOT1 = VCBOOT2 = 7V 10 nA
(1) A typical is the center of characterization data measured with low duty cycle pulse tsting at TA= 25°C. Typicals are not ensured.
(2) All limits are specified. All electrical characteristics having room-temperature limits are tested during production with TA= TJ= 25°C. All
hot and cold limits are specified by correlating the electrical characteristics to process and temperature variations and applying statistical
process control.
(3) Both switching controllers are off. The linear regulator VLIN5 remains on.
(4) The output voltage at the VLIN5 pin may be as high as 5.9V in shutdown mode (ON/SS1 = ON/SS2 = 0V).
(5) When SS1 and SS2 pins are charged above this voltage and either of the output voltages at Vout1 or Vout2 is still below the regulation
limit, the under voltage protection feature is initialized.
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ELECTRICAL CHARACTERISTICS (continued)
Unless otherwise specified, VIN = 15V, GND = PGND = 0V, VLIN5 = VDD1 = VDD2. Limits appearing in boldface type apply
over the specified operating junction temperature range, (-20°C to +125°C, if not otherwise specified). Specifications
appearing in plain type are measured using low duty cycle pulse testing with TA= 25°C (1),(2). Min/Max limits are specified by
design, test, or statistical analysis.
Symbol Parameter Conditions Min Typ Max Units
ISC_DRV HDRVx and LDRVx Source VCBOOT1 = VCBOOT2 = 5V, VSWx=0V, 0.5 A
Current HDRVx=LDRVx=2.5V
Isk_HDRV HDRVx Sink Current VCBOOTx = VDDx = 5V, VSWx = 0V, HDRVX 0.8 A
= 2.5V
Isk_LDRV LDRVx Sink Current VCBOOTx = VDDx = 5V, VSWx = 0V, LDRVX 1.1 A
= 2.5V
RHDRV HDRV1 & 2 Source On- VCBOOT1 = VCBOOT2 = 5V, 3.1 Ω
Resistance VSW1 = VSW2 = 0V
HDRV1 & 2 Sink On- 1.5 Ω
Resistance
RLDRV LDRV1 & 2 Source On- VCBOOT1 = VCBOOT2 = 5V, 3.1 Ω
Resistance VSW1 = VSW2 = 0V
VDD1 = VDD1 = 5V
LDRV1 & 2 Sink On- 1.1 Ω
Resistance
Oscillator
Fosc Oscillator Frequency 260 300 340 kHz
-40°C to 125°C 257.5 340
Don_max Maximum On-Duty Cycle VFB1 = VFB2 = 1V, Measured at pins 96 98
HDRV1 and HDRV2 %
-40°C to 125°C 95.64
Ton_min Minimum On-Time 166 ns
SSOT_delta HDRV1 and HDRV2 Delta ON/SS1 = ON/SS2 = 2V 20 150 ns
On Time
Error Amplifier
IFB1, IFB2 Feedback Input Bias VFB1_FIX = 1.5V, VFB2_FIX = 1.5V 65 ±200 nA
Current
Icomp1_SC, COMP Output Source VFB1_FIX = VFB2_FIX = 1V, 18 113
Icomp2_SC Current VCOMP1 = VCOMP2 = 1V µA
0°C to 125°C 32
-40°C to 125°C 6
Icomp1_SK, COMP Output Sink Current VFB1_FIX = VFB2_FIX = 1.5V and 18 108
Icomp2_SK VCOMP1 = VCOMP2 = 0.5V µA
0°C to 125°C 32
-40°C to 125°C 6
gm1, gm2 Transconductance 650 µmho
GISNS1, Current Sense Amplifier VCOMPx = 1.25V 4.2 5.2 7.5
GISNS2 (1&2) Gain
Voltage References and Linear Voltage Regulators
UVLO VLIN5 Under-voltage ON/SS1, ON/SS2 transition
Lockout from low to high 3.6 4.0 4.4 V
Threshold Rising
Logic Outputs
IOL PGOOD Low Sink Current VPGOOD = 0.4V 0.60 0.95 mA
IOH PGOOD High Leakage VPGOOD = 5V 5200 nA
Current
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Product Folder Links: LM2642
C16
22uF
C7
0.1uF
C20
1nF
R1
13k
Q1
FDS6690A
Q2
FDS6690A
L1
8.2uH
D3A
BAW56 D4
MBRS140T3
C8
100uF
R27
4R7
C5
10uF
GND
C2 10nF
VLIN5
R32
4R7 VDD
C25
0.1uF
R13
13k
Q3
FDS6690A
Q4
FDS6690A
L2
6uH C22
100uF
D3B
BAW56
D5 MBRS140T3
C17
10uF
C13 10nF
R33
4R7 VDD
VIN
VLIN5
R28
220k
FB1 4
ILIM1 2
SGND
8
COMP1
3
ON/SS1
9
COMP2
12
RSNS1 28
KS1 1
SW1 27
HDRV1 26
CBOOT1 25
VDD1
24
LDRV1 23
PGND 21
VIN
22
VLIN5
7
PGOOD1
5
VDD2
19
UV_DELAY
6
FB2 11
ILIM2 13
RSNS2 15
KS2 14
SW2 16
HDRV2 17
CBOOT2 18
LDRV2 20
IC1
LM2642
VDD
5V/3A
PGOOD1
R10
60k4
R11
20k
R19
33k2
R20
20k
3.3V/3A
C1
1uF
C34
0.1uF
ON/SS2
10
C12
10nF
C11
10nF
C6
22uF
C26
4.7uF
C27
1uF
C19
1nF
R24
20k
R23
20k
Vo1
Vo2
+
+
+
+
+
VIN 6V-30V
LM2642
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SNVS203I –MAY 2002–REVISED APRIL 2013
Figure 2. Typical 2 Channel Application Circuit
Copyright © 2002–2013, Texas Instruments Incorporated Submit Documentation Feedback 7
Product Folder Links: LM2642
VIN+ 4.5V-20V
GND
UV_DELAY
PGOOD1
ON/SS1
ON/SS2
COMP1
COMP2
VLIN5
VDD1
VDD2
SGND FB2
LDRV2
CBOOT2
SW2
HDRV2
RSNS2
KS2
ILIM2
FB1
PGND
LDRV1
CBOOT1
SW1
HDRV1
RSNS1
KS1
ILIM1
VIN
IC1
LM2642
+ +
++
VDD
VDD
VIN
VLIN5 Vout
1.8V/14A
GND
C8
C23
220uF
220uF
220uF
C22 220uF
C9
C25
0.1uF D3B
L2
3u6H
4R7
R33
D5
Q4
Q3
R16
100R
100R
R14
R13
11K
C15
100pF
C17 C16
10uF 22uF
10nF 100pFC13 C14
C2 C3
C5 C6
C7
R1 R2
R7
R6
C4
R32 L1
D4
Q2 R10
R11
D3A
3u6H
9k
20k
0.1uF
4R7
Q1
100R
100R11k
12m
R15
12m
10nF 100pF
100pF
22uF
10uF
2.2nF
470pF
20k
4R7
1uF
4.7uF
R27
C27
C26
R23
C19
C18
C11
22nF
S1
R28
220K
C34
0.1uF
C1
1uF
PGOOD1
ON1/2
LM2642
SNVS203I –MAY 2002–REVISED APRIL 2013
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Figure 3. Typical Single Channel Application Circuit
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-
Vref
+
-
ILIMx
KSx
10uA
RSNSx
+
VDDx
LDRVx
PGNDx
Input Power
Supply
Shoot through
protection
sequencer
Q
Q
R
S
Shifter
& latch
+
-
+
0.50V
+
-
Corrective
ramp
SGND
OSC
300 kHz
0 180
To Ch2
PGOOD
Q
Q
R
S
Reset by
POR or SD
UVP
OVP
ON/SSx
ON/OFF
&
S/S
control
PWM comp
5V LDO
(Allways ON)
fault FAULT
TSD
UVLO
OVP
UVPG1
comparator
UVP
COMPx
CHx
output
error amp
BG
Ch1 and Ch2 are identical
HDRVx
CHx
Output
S/S level Cycle
Skip
comp
SWx
Q
Q
R
S
UV_DELAY
+
-
+
-
ILIM
Comp
ISENSE
amp
+
-CBOOTx
-
+
BG
reference
BG
IREF Current
bias
Voltage
and
Current
generator
From another Ch.
From
another
CH.
Bias
Generator
VIN
Normal:
ON
SS:
ON
FBx
2uA
VLIN5
7uA
SD Disable
Active
discharge
Rdson=
500 Ohm
5uA
UV
PWM logic
control
LM2642
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SNVS203I –MAY 2002–REVISED APRIL 2013
BLOCK DIAGRAM
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100ms/div
VOUT1
2V/div
ON/SS1=ON/SS2
5V/div
VOUT2
2V/div
10ms/div
ILOAD
10A/div
VOUT
1V/div
ON/SS
2V/div UV_DELAY
2V/div
10ms/div
ILOAD
10A/div
VOUT1
2V/div
VOUT2
1V/div
UV_DELAY
2ms/div
VOUT1
2V/div
VOUT2
2V/div
VOUT1
2V/div
VOUT2
2V/div
VIN
22V/div
VIN
7V/div
ON/SS1=ON/SS2
2V/div
2ms/div
PGOOD1
5V/div
VOUT1
2V/div
ON/SS1=ON/SS2
1V/div
VIN
10V/div
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TYPICAL PERFORMANCE CHARACTERISTICS
Softstart Waveforms Power On and PGOOD1 Waveforms
(ILOAD1 = ILOAD2 = 0A) (ILOAD1 = ILOAD2 = 0A)
Figure 4. Figure 5.
Over-Current and UVP Shutdown
UVP Startup Waveforms (ILOAD2 = 0A)
Figure 6. Figure 7.
Shutdown Waveforms Ch.1 Load Transient Response
(ILOAD1 = ILOAD2 = 0A) 5VOUT, 12VIN
Figure 8. Figure 9.
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-25 -5 15 35 55 75 95 115 135
TEMPERATURE (oC)
5.01
5.02
5.03
5.04
5.05
5.06
5.07
VLIN5 (V)
VIN=5.5V
VIN=30V
5.5 10.5 15.5 20.5 25.5 30
4.98
4.99
5
5.01
5.02
5.03
5.04
5.05
5.06
VLIN5 (V)
VIN (V)
-25 -5 15 35 55 75 95 115 135
TEMPERATURE (oC)
32
34
36
38
40
42
44
Iq (PA)
35
VIN (V)
Iq (uA)
37
39
41
43
45
47
49
5.5 10.5 15.5 30
20.5 25.5
LM2642
www.ti.com
SNVS203I –MAY 2002–REVISED APRIL 2013
Ch.2 Load Transient Response Load Transient Response
3.3VOUT, 12VIN Parallel Operation 1.8VOUT, 12VIN
Figure 10. Figure 11.
Input Supply Current Input Supply Current
vs vs
Temperature VIN
(Shutdown Mode VIN = 15V) Shutdown Mode (25°C)
Figure 12. Figure 13.
VLIN5 VLIN5
vs vs
Temperature VIN (25°C)
Figure 14. Figure 15.
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40
1.E-02 1.E-01 1.E+00 1.E+01
LOAD CURRENT(A)
EFFICIENCY (%)
VIN=7V
VIN=22V
VIN=12V
50
60
70
80
90
100
40
1.E-02 1.E-01 1.E+00 1.E+01
LOAD CURRENT(A)
EFFICIENCY (%)
VIN=7V
VIN=22V
VIN=12V
50
60
70
80
90
100
700.0
JUNCTION TEMPEARTURE (oC)
EA gm (umho)
650.0
600.0
550.0
500.0
450.0
400.0
-40 -20 0 20 40 60 80 100 120 140
40
1.E-02 1.E-01 1.E+00 1.E+01
LOAD CURRENT(A)
EFFICIENCY (%)
VIN=7V
VIN=22V
VIN=12V
50
60
70
80
90
100
-40 -20 0 20 40 60 80 100 120
TEMPERATURE (oC)
1.234
1.236
1.238
1.240
1.242
1.244
VREF (V)
1.232
FREQUENCY (kHz)
320
315
310
305
300
295
290
-40 -20 0 20 40 60 80 100 120
285
280
275
270
TEMPERATURE (oC)
LM2642
SNVS203I –MAY 2002–REVISED APRIL 2013
www.ti.com
FB Reference Voltage Operating Frequency
vs vs
Temperature Temperature
Figure 16. Figure 17.
Efficiency
Error Amplifier Gain vs
vs Load Current
Temperature Ch.1 = 5V, Ch.2 = Off
Figure 18. Figure 19.
Efficiency Efficiency
vs vs
Load Current Load Current
Ch.2 = 2.5V, Ch.1 = Off Ch.2 = 3.3V, Ch.1 = Off
Figure 20. Figure 21.
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Product Folder Links: LM2642
+
-
+
-SS:0.55V
OP:2V
0.45V
high clamp
low clamp
COMPx
+
-
2uA
7uA
+
-
Q
Q
R
S
S>R
1.2V/
1.05V ON/OFF
comparator
+
-S/S level
S/S buffer
ON: 2uA source
Fault: 5uA sink
ON/SSx ONx
fault
disable
LM2642
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APPLICATION INFORMATION
OPERATION DESCRIPTIONS
SOFT START
The ON/SS1 pin has dual functionality as both channel enable and soft start control. The soft start block diagram
is shown in Figure 22.
The LM2642 will remain in shutdown mode while both soft start pins are grounded.In a normal application (with a
soft start capacitor connected between the ON/SS1 pin and SGND) soft start functions as follows. As the input
voltage rises (note: Iss starts to flow when VIN ≥2.2V), the internal 5V LDO starts up, and an internal 2µA
current charges the soft start capacitor. During soft start phase, the error amplifier output voltage at the COMPx
pin is clamped at 0.55V and the duty cycle is controlled only by the soft start voltage. As the SSx pin voltage
ramps up, the duty cycle increases proportional to the soft start ramp, causing the output voltage to ramp up. The
rate at which the duty cycle increases depends on the capacitance of the soft start capacitor. The higher the
capacitance, the slower the output voltage ramps up. When the corresponding output voltage exceeds 98%
(typical) of the set target voltage, the regulator switches from soft start to normal operating mode. At this time,
the 0.55V clamp at the output of the error amplifier releases and peak current feedback control takes over. Once
in peak current feedback control mode, the output of the error amplifier will travel within the 0.5V and 2V window
to achieve PWM control. See Figure 23.
During soft start, over-voltage protection and current limit remain in effect. The under voltage protection feature is
activated when the ON/SS pin exceeds the timeout threshold (3.3V typical). If the ON/SSx capacitor is too small,
the duty cycle may increase too rapidly, causing the device to latch off due to output voltage overshoot above the
OVP threshold. This becomes more likely in applications requiring low output voltage, high input voltage and light
load. A capacitance of 10nF is recommended at each soft start pin to provide a smooth monotonic output ramp.
Figure 22. Soft Start and ON/OFF
Figure 23. Voltage Clamp at COMPx Pin
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FBx
1.13BG
in: 0.94BG
out: 0.91BG
in: 0.84BG
out: 0.80BG
OFF 1
UVPG1
UVP ½ (latched)
OVP ½ (latched)
ONx
SS Timeout
SD
power on
reset shutdown
latch UVP
HDRV: off
LDRV: off
shutdown
latch OVP
HDRV: off
LDRV: on
TSD
UVLO
PGOOD1
OVP 1/2
UV_DELAY
5 µA
fault
from other CH.
OVPx
UVPGx
UVPx
from other CH.
_
+
_
+
_
+
OVP & PG
UVPG
UVP
PGOOD Protection
Comparators
LM2642
SNVS203I –MAY 2002–REVISED APRIL 2013
www.ti.com
SEQUENTIAL STARTUP
Sequential startup can be implemented by simply connecting PGOOD1 to SS/ON2. Once channel 1 has reached
94% of nominal, PGOOD1 will go high, thus enabling SS/ON2. In this mode of operation, channel 2 will be
controlled by the state of channel 1. If channel 1 falls out of the PGOOD1 window, channel 2 will be switched off
immediately.
Figure 24. PGOOD, OVP and UVP
OVER VOLTAGE PROTECTION (OVP)
If the output voltage on either channel rises above 113% of nominal, over voltage protection activates. Both
channels will latch off, and the PGOOD1 pin will go low. When the OVP latch is set, the high side FET driver,
HDRVx, is immediately turned off and the low side FET driver, LDRVx, is turned on to discharge the output
capacitor through the inductor. To reset the OVP latch, either the input voltage must be cycled, or both channels
must be switched off.
UNDER VOLTAGE PROTECTION (UVP) AND UV DELAY
If the output voltage on either channel falls below 80% of nominal, under voltage protection activates. As shown
in Figure 24, an under-voltage event will shut off the UV_DELAY MOSFET, which will allow the UV_DELAY
capacitor to charge at 5uA (typical). At the UV_DELAY threshold (2.3V typical) both channels will latch off. Also,
UV_DELAY will be disabled and the UV_DELAY pin will return to 0V. During UVP, both the high side and low
side FET drivers will be turned off. If no capacitor is connected to the UV_DELAY pin, the UVP latch will be
activated immediately. To reset the UVP latch, either the input voltage must be cycled, or both ON/SS pins must
be pulled low. The UVP function can be disabled by connecting the UV_DELAY pin to ground.
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POWER GOOD
A power good pin (PGOOD1) is available to monitor the output status of Channel 1. As shown in Figure 24, the
pin connects to the output of an open drain MOSFET, which will remain open while Channel 1 is within operating
range. PGOOD1 will go low (low impedance to ground) under the following four conditions:
1. Channel 1 is turned off
2. Channel 1 output falls below 90.3% of nominal (UVPG1)
3. OVP on either channel
4. UVP on either channel
When on, the PGOOD1 pin is capable of sinking 0.95mA (typical). If an OVP or UVP condition occurs, both
channels will latch off, and the PGOOD1 pin will be latched low. During a UVPG1 condition, however, PGOOD1
will not latch off. The pin will stay low until Channel 1 output voltage returns to 94% (typical) of nominal. See
Vpwrgd in the Electrical Characteristics table.
OUTPUT CAPACITOR DISCHARGE
Each channel has an embedded 480ΩMOSFET with the drain connected to the SWx pin. This MOSFET will
discharge the output capacitor of its channel if its channel is off, or the IC enters a fault state caused by one of
the following conditions:
1. UVP
2. UVLO
3. Thermal shut-down (TSD)
If an output over voltage event occurs, the HDRVx will be turned off and LDRVx will be turned on immediately to
discharge the output capacitor of both channels through the inductor.
BOOTSTRAP DIODE SELECTION
The bootstrap diode and capacitor form a supply that floats above the switch node voltage. VLIN5 powers this
supply, creating approximately 5V (minus the diode drop) which is used to power the high side FET drivers and
driver logic. When selecting a bootstrap diode, Schottky diodes are preferred due to their low forward voltage
drop, but care must be taken for circuits that operate at high ambient temperature. The reverse leakage of some
Schottky diodes can increase by more than 1000x at high temperature, and this leakage path can deplete the
charge on the bootstrap capacitor, starving the driver and logic. Standard PN junction diodes and fast rectifier
diodes can also be used, and these types maintain tighter control over reverse leakage current across
temperature.
SWITCHING NOISE REDUCTION
Power MOSFETs are very fast switching devices. In synchronous rectifier converters, the rapid increase of drain
current in the top FET coupled with parasitic inductance will generate unwanted Ldi/dt noise spikes at the source
node of the FET (SWx node) and also at the VIN node. The magnitude of this noise will increase as the output
current increases. This parasitic spike noise may turn into electromagnetic interference (EMI), and can also
cause problems in device performance. Therefore, it must be suppressed using one of the following methods.
It is strongly recommended to add R-C filters to the current sense amplifier inputs as shown in Figure 26. This
will reduce the susceptibility to switching noise, especially during heavy load transients and short on time
conditions. The filter components should be connected as close as possible to the IC. Note that these filters
should be used when a current sense resistor is used.
As shown in Figure 25, adding a resistor in series with the SWx pin will slow down the gate drive (HDRVx), thus
slowing the rise and fall time of the top FET, yielding a longer drain current transition time.
Usually a 3.3Ωto 4.7Ωresistor is sufficient to suppress the noise. Top FET switching losses will increase with
higher resistance values.
Small resistors (1-5 ohms) can also be placed in series with the HDRVx pin or the CBOOTx pin to effectively
reduce switch node ringing. A CBOOT resistor will slow the rise time of the FET, whereas a resistor at HDRV will
reduce both rise and fall times.
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Product Folder Links: LM2642
+
-
+
-
10uA
comp
ISENSE
amp
POWER
SUPPLY
100
100
100pF 100pF
LIMx
KSx
RSNSx
10nF
20m
13k
LIMx
Rsw
CBOOTx
4R7
HDRVx
SWx 0.1uF
LM2642
SNVS203I –MAY 2002–REVISED APRIL 2013
www.ti.com
Figure 25. SW Series Resistor
CURRENT SENSING AND LIMITING
As shown in Figure 26, the KSx and RSNSx pins are the inputs of the current sense amplifier. Current sensing is
accomplished either by sensing the Vds of the top FET or by sensing the voltage across a current sense resistor
connected from VIN to the drain of the top FET. The advantage of sensing current across the top FET are
reduced parts count, cost and power loss, whereas using a current sense resistor improves the current sense
accuracy. Keeping the differential current-sense voltage below 200mV ensures linear operation of the current
sense amplifier. Therefore, the Rdson of the top FET or the current sense resistor must be small enough so that
the current sense voltage does not exceed 200mV when the top FET is on. There is a leading edge blanking
circuit that forces the top FET on for at least 166ns. Beyond this minimum on time, the output of the PWM
comparator is used to turn off the top FET. Additionally, a minimum voltage of at least 50mV across Rsns is
recommended to ensure a high SNR at the current sense amplifier.
Assuming a maximum of 200mV across Rsns, the current sense resistor can be calculated as follows:
(1)
where Imax is the maximum expected load current, including overload multiplier (ie:120%), and Irip is the
inductor ripple current (See equation 7). The above equation gives the maximum allowable value for Rsns.
Switching losses will increase with Rsns, thus lowering efficiency.
The peak current limit is set by an external resistor connected between the ILIMx pin and the KSx pin. An
internal 10µA current sink on the ILIMx pin produces a voltage across the resistor to set the current limit
threshold which is compared to the current sense voltage. A 10nF capacitor across this resistor is required to
filter unwanted noise that could improperly trip the current limit comparator.
Figure 26. Current Sense and Current Limit
Current limit is activated when the inductor current is high enough to cause the voltage at the RSNSx pin to be
lower than that of the ILIMx pin. This toggles the comparator, thus turning off the top FET immediately. The
comparator is disabled either when the top FET is turned off or during the leading edge blanking time. The
equation for current limit resistor, Rlim, is as follows:
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(2)
Where Ilim is the load current at which the current limit comparator will be tripped.
When sensing current across the top FET, replace Rsns with the Rdson of the FET. This calculated Rlim value
specifies that the minimum current limit will not be less than Imax. It is recommended that a 1% tolerance resistor
be used.
When sensing across the top FET, Rdson will show more variation than a current sense resistor, largely due to
temperature. Rdson will increase proportional to temperature according to a specific temperature coefficient.
Refer to the manufacturer's datasheet to determine the range of Rdson values over operating temperature or see
the Component Selection section (equation 12) for a calculation of maximum Rdson. This will prevent Rdson
variations from prematurely setting off the current limit comparator as the operating temperature increases.
To ensure accurate current sensing, special attention in board layout is required. The KSx and RSNSx pins
require separate traces to form a Kelvin connection to the corresponding current sense nodes.
INPUT UNDER VOLTAGE LOCKOUT (UVLO)
The input under-voltage lock out threshold, which is sensed via the VLIN5 internal LDO output, is 4.0V (typical).
Below this threshold, both HDRVx and LDRVx will be turned off and the internal 480ΩMOSFETs will be turned
on to discharge the output capacitors through the SWx pins. During UVLO, the ON/SS pins will sink 5mA to
discharge the soft start capacitors and turn off both channels. As the input voltage increases again above 4.0V,
UVLO will be de-activated, and the device will restart again from soft start phase. If the voltage at VLIN5 remains
below 4.5V, but above the 4.0V UVLO threshold, the device cannot be ensured to operate within specification.
If the input voltage is between 4.0V and 5.2V, the VLIN5 pin will not regulate, but will follow approximately
200mV below the input voltage.
DUAL-PHASE PARALLEL OPERATION
In applications with high output current demand, the two switching channels can be configured to operate as a
two-180° out of phase converter to provide a single output voltage with current sharing between the two
switching channels. This approach greatly reduces the stress and heat on the output stage components while
lowering input ripple current. The sum of inductor ripple current is also reduced which results in lowering output
ripple voltage. Figure 3 shows an example of a typical two-phase circuit. Because precision current sense is the
primary design criteria to ensure accurate current sharing between the two channels, both channels must use
external sense resistors for current sensing. To minimize the error between the error amplifiers of the two
channels, tie the feedback pins FB1 and FB2 together and connect to a single voltage divider for output voltage
sensing. Also, tie the COMP1 and COMP2 together and connect to the compensation network. ON/SS1 and
ON/SS2 must be tied together to enable and disable both channels simultaneously.
COMPONENT SELECTION
OUTPUT VOLTAGE SETTING
The output voltage for each channel is set by the ratio of a voltage divider as shown in Figure 27. The resistor
values can be determined by the following equation:
(3)
Where Vfb=1.238V. Although increasing the value of R1 and R2 will increase efficiency, this will also decrease
accuracy. Therefore, a maximum value is recommended for R2 in order to keep the output within .3% of Vnom.
This maximum R2 value should be calculated first with the following equation:
(4)
Where 200nA is the maximum current drawn by FBx pin.
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VIN
VOUT
0 2 4 6 8 10 1214 16 18 2022 24 26 28 30
0
5
10
15
20
25
30
Vout
R2
FBx
GND R1
LM2642
SNVS203I –MAY 2002–REVISED APRIL 2013
www.ti.com
Figure 27. Output Voltage Setting
Example: Vnom=5V, Vfb=1.238V, Ifbmax=200nA.
(5)
Choose 60K
(6)
The output voltage is limited by the maximum duty cycle as well as the minimum on time. Figure 28 shows the
limits for input and output voltages. The recommended maximum output voltage is approximately 1V less than
the nominal input voltage. At 30V input, the minimum output is approximately 2.3V and the maximum is
approximately 27V.
For input voltages below 5.5V, VLIN5 must be connected to Vin through a small resistor (approximately 4.7
ohm). This will ensure that VLIN5 does not fall below the UVLO threshold.
Figure 28. Available Output Voltage Range
OUTPUT CAPACITOR SELECTION
In applications that exhibit large and fast load current swings, the slew rate of such a load current transient may
be beyond the response speed of the regulator. Therefore, to meet voltage transient requirements during worst-
case load transients, special consideration should be given to output capacitor selection. The total combined
ESR of the output capacitors must be lower than a certain value, while the total capacitance must be greater
than a certain value. Also, in applications where the specification of output voltage regulation is tight and ripple
voltage must be low, starting from the required output voltage ripple will often result in fewer design iterations.
ALLOWED TRANSIENT VOLTAGE EXCURSION
The allowed output voltage excursion during a load transient (ΔVc_s) is:
(7)
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Where ±δ% is the output voltage regulation window and ±ε% is the output voltage initial accuracy.
Example: Vnom = 5V, δ% = 7%, ε% = 3.4%, Vrip = 40mV peak to peak.
(8)
Since the ripple voltage is included in the calculation of ΔVc_s, the inductor ripple current should not be included
in the worst-case load current excursion. That is, the worst-case load current excursion should be simply
maximum load current change specification, ΔIc_s.
MAXIMUM ESR CALCULATION
Unless the rise and fall times of a load transient are slower than the response speed of the control loop, if the
total combined ESR (Re) is too high, the load transient requirement will not be met, no matter how large the
capacitance.
The maximum allowed total combined ESR is:
(9)
Example: ΔVc_s = 160mV, ΔIc_s = 3A. Then Re_max = 53.3mΩ.
Maximum ESR criterion can be used when the associated capacitance is high enough, otherwise more
capacitors than the number determined by this criterion should be used in parallel.
MINIMUM CAPACITANCE CALCULATION
In a switch mode power supply, the minimum output capacitance is typically dictated by the load transient
requirement. If there is not enough capacitance, the output voltage excursion will exceed the maximum allowed
value even if the maximum ESR requirement is met. The worst-case load transient is an unloading transient that
happens when the input voltage is the highest and when the present switching cycle has just finished. The
corresponding minimum capacitance is calculated as follows:
(10)
Notice it is already assumed the total ESR, Re, is no greater than Re_max, otherwise the term under the square
root will be a negative value. Also, it is assumed that L has already been selected, therefore the minimum L
value should be calculated before Cmin and after Re (see Inductor Selection below). Example: Re = 20mΩ,
Vnom = 5V, ΔVc_s = 160mV, ΔIc_s = 3A, L = 8µH
(11)
Generally speaking, Cmin decreases with decreasing Re, ΔIc_s, and L, but with increasing Vnom and ΔVc_s.
INDUCTOR SELECTION
The size of the output inductor can be determined from the desired output ripple voltage, Vrip, and the
impedance of the output capacitors at the switching frequency. The equation to determine the minimum
inductance value is as follows:
(12)
In the above equation, Re is used in place of the impedance of the output capacitors. This is because in most
cases, the impedance of the output capacitors at the switching frequency is very close to Re. In the case of
ceramic capacitors, replace Re with the true impedance.
Example: Vin (max)= 30V, Vnom = 5.0V, Vrip = 40mV, Re =20mΩ, f = 300kHz
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(13)
Lmin = 7µH
The actual selection process usually involves several iterations of all of the above steps, from ripple voltage
selection, to capacitor selection, to inductance calculations. Both the highest and the lowest input and output
voltages and load transient requirements should be considered. If an inductance value larger than Lmin is
selected, make sure that the Cmin requirement is not violated.
Priority should be given to parameters that are not flexible or more costly. For example, if there are very few
types of capacitors to choose from, it may be a good idea to adjust the inductance value so that a requirement of
3.2 capacitors can be reduced to 3 capacitors.
Since inductor ripple current is often the criterion for selecting an output inductor, it is a good idea to double-
check this value. The equation is:
(14)
Where D is the duty cycle, defined by Vnom/Vin.
Also important is the ripple content, which is defined by Irip /Inom. Generally speaking, a ripple content of less
than 50% is ok. Larger ripple content will cause too much loss in the inductor.
Example: Vin = 12V, Vnom = 5.0V, f = 300kHz, L = 8µH
(15)
Given a maximum load current of 3A, the ripple content is 1.2A / 3A = 40%.
When choosing the inductor, the saturation current should be higher than the maximum peak inductor current
and the RMS current rating should be higher than the maximum load current.
INPUT CAPACITOR SELECTION
The fact that the two switching channels of the LM2642 are 180° out of phase will reduce the RMS value of the
ripple current seen by the input capacitors. This will help extend input capacitor life span and result in a more
efficient system. Input capacitors must be selected that can handle both the maximum ripple RMS current at
highest ambient temperature as well as the maximum input voltage. In applications in which output voltages are
less than half of the input voltage, the corresponding duty cycles will be less than 50%. This means there will be
no overlap between the two channels' input current pulses. The equation for calculating the maximum total input
ripple RMS current for duty cycles under 50% is:
(16)
where I1 is maximum load current of Channel 1, I2 is the maximum load current of Channel 2, D1 is the duty
cycle of Channel 1, and D2 is the duty cycle of Channel 2.
Example: Imax_1 = 3.6A, Imax_2 = 3.6A, D1 = 0.42, and D2 = 0.275
(17)
Choose input capacitors that can handle 1.66A ripple RMS current at highest ambient temperature. In
applications where output voltages are greater than half the input voltage, the corresponding duty cycles will be
greater than 50%, and there will be overlapping input current pulses. Input ripple current will be highest under
these circumstances. The input RMS current in this case is given by:
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(18)
Where, again, I1 and I2 are the maximum load currents of channel 1 and 2, and D1 and D2 are the duty cycles.
This equation should be used when both duty cycles are expected to be higher than 50%.
Input capacitors must meet the minimum requirements of voltage and ripple current capacity. The size of the
capacitor should then be selected based on hold up time requirements. Bench testing for individual applications
is still the best way to determine a reliable input capacitor value. The input capacitor should always be placed as
close as possible to the current sense resistor or the drain of the top FET.
MOSFET SELECTION
BOTTOM FET SELECTION
During normal operation, the bottom FET is switching on and off at almost zero voltage. Therefore, only
conduction losses are present in the bottom FET. The most important parameter when selecting the bottom FET
is the on resistance (Rdson). The lower the on resistance, the lower the power loss. The bottom FET power loss
peaks at maximum input voltage and load current. The equation for the maximum allowed on resistance at room
temperature for a given FET package, is:
(19)
where Tj_max is the maximum allowed junction temperature in the FET, Ta_max is the maximum ambient
temperature, Rθja is the junction-to-ambient thermal resistance of the FET, and TC is the temperature coefficient
of the on resistance which is typically in the range of 10,000ppm/°C.
If the calculated Rdson_max is smaller than the lowest value available, multiple FETs can be used in parallel.
This effectively reduces the Imax term in the above equation, thus reducing Rdson. When using two FETs in
parallel, multiply the calculated Rdson_max by 4 to obtain the Rdson_max for each FET. In the case of three
FETs, multiply by 9.
(20)
If the selected FET has an Rds value higher than 35.3Ω, then two FETs with an Rdson less than 141mΩ(4 x
35.3mΩ) can be used in parallel. In this case, the temperature rise on each FET will not go to Tj_max because
each FET is now dissipating only half of the total power.
TOP FET SELECTION
The top FET has two types of losses: switching loss and conduction loss. The switching losses mainly consist of
crossover loss and bottom diode reverse recovery loss. Since it is rather difficult to estimate the switching loss, a
general starting point is to allot 60% of the top FET thermal capacity to switching losses. The best way to
precisely determine switching losses is through bench testing. The equation for calculating the on resistance of
the top FET is thus:
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10 100 1
k10
k100
k1M
FREQUENCY
(Hz)
-60
-40
-20
0
20
GAIN
(dB)
0
-45
-90
-135
-180
PHASE (°)
Asymptoti
c
Gain
Phas
e
LM2642
SNVS203I –MAY 2002–REVISED APRIL 2013
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(21)
Example: Tj_max = 100°C, Ta_max = 60°C, Rqja = 60°C/W, Vin_min = 5.5V, Vnom = 5V, and Iload_max = 3.6A.
(22)
When using FETs in parallel, the same guidelines apply to the top FET as apply to the bottom FET.
LOOP COMPENSATION
The general purpose of loop compensation is to meet static and dynamic performance requirements while
maintaining stability. Loop gain is what is usually checked to determine small-signal performance. Loop gain is
equal to the product of control-output transfer function and the output-control transfer function (the compensation
network transfer function). Generally speaking it is a good idea to have a loop gain slope that is -20dB /decade
from a very low frequency to well beyond the crossover frequency. The crossover frequency should not exceed
one-fifth of the switching frequency, i.e. 60kHz in the case of LM2642. The higher the bandwidth is, the faster the
load transient response speed will potentially be. However, if the duty cycle saturates during a load transient,
further increasing the small signal bandwidth will not help. Since the control-output transfer function usually has
very limited low frequency gain, it is a good idea to place a pole in the compensation at zero frequency, so that
the low frequency gain will be relatively large. A large DC gain means high DC regulation accuracy (i.e. DC
voltage changes little with load or line variations). The rest of the compensation scheme depends highly on the
shape of the control-output plot.
Figure 29. Control-Output Transfer Function
As shown in Figure 29, the control-output transfer function consists of one pole (fp), one zero (fz), and a double
pole at fn (half the switching frequency). The following can be done to create a -20dB /decade roll-off of the loop
gain: Place the first pole at 0Hz, the first zero at fp, the second pole at fz, and the second zero at fn. The
resulting output-control transfer function is shown in Figure 30.
22 Submit Documentation Feedback Copyright © 2002–2013, Texas Instruments Incorporated
Product Folder Links: LM2642
fp max =
2S 300k 8P 100PF
x
x
x
.5
1
2S 1.7: 100PF
x
x
+
= 1.27kHz
-20dB/dec
(fp1 is at zero frequency)
-20dB/dec
FREQUENCY
GAIN (dB)
B
fz1 fz2
fp2
LM2642
www.ti.com
SNVS203I –MAY 2002–REVISED APRIL 2013
Figure 30. Output-Control Transfer Function
The control-output corner frequencies, and thus the desired compensation corner frequencies, can be
determined approximately by the following equations:
(23)
(24)
Since fp is determined by the output network, it will shift with loading (Ro) and duty cycle. First determine the
range of frequencies (fpmin/max) of the pole across the expected load range, then place the first compensation
zero within that range.
Example: Re= 20mΩ, Co= 100µF, Romax = 5V/100mA = 50Ω, Romin = 5V/3A = 1.7Ω:
(25)
(26)
(27)
Once the fp range is determined, Rc1 should be calculated using:
(28)
Where B is the desired gain in V/V at fp (fz1), gm is the transconductance of the error amplifier, and R1 and R2
are the feedback resistors. A gain value around 10dB (3.3v/v) is generally a good starting point.
Example: B = 3.3 v/v, gm=650 m, R1 = 20 KΩ, R2 = 60.4 KΩ:
(29)
Bandwidth will vary proportional to the value of Rc1. Next, Cc1 can be determined with the following equation:
(30)
Example: fpmin = 363 Hz, Rc1=20 KΩ:
(31)
Copyright © 2002–2013, Texas Instruments Incorporated Submit Documentation Feedback 23
Product Folder Links: LM2642
Vo
Vc
R2
R1
gm
compensation
network
CC1 CC2
RC1 RC2
LM2642
SNVS203I –MAY 2002–REVISED APRIL 2013
www.ti.com
The value of Cc1 should be within the range determined by Fpmin/max. A higher value will generally provide a
more stable loop, but too high a value will slow the transient response time.
The compensation network (Figure 31) will also introduce a low frequency pole which will be close to 0Hz.
A second pole should also be placed at fz. This pole can be created with a single capacitor Cc2 and a shorted
Rc2 (see Figure 31). The minimum value for this capacitor can be calculated by:
(32)
Cc2 may not be necessary, however it does create a more stable control loop. This is especially important with
high load currents and in current sharing mode.
Example: fz = 80 kHz, Rc1 = 20 KΩ:
(33)
A second zero can also be added with a resistor in series with Cc2. If used, this zero should be placed at fn,
where the control to output gain rolls off at -40dB/dec. Generally, fn will be well below the 0dB level and thus will
have little effect on stability. Rc2 can be calculated with the following equation:
(34)
Figure 31. Compensation Network
24 Submit Documentation Feedback Copyright © 2002–2013, Texas Instruments Incorporated
Product Folder Links: LM2642
LM2642
www.ti.com
SNVS203I –MAY 2002–REVISED APRIL 2013
REVISION HISTORY
Changes from Revision H (April 2013) to Revision I Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 24
Copyright © 2002–2013, Texas Instruments Incorporated Submit Documentation Feedback 25
Product Folder Links: LM2642
PACKAGE OPTION ADDENDUM
www.ti.com 7-Nov-2017
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM2642MTC/NOPB ACTIVE TSSOP PW 28 48 Green (RoHS
& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 LM2642MTC
LM2642MTCX/NOPB ACTIVE TSSOP PW 28 2500 Green (RoHS
& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 LM2642MTC
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
PACKAGE OPTION ADDENDUM
www.ti.com 7-Nov-2017
Addendum-Page 2
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM2642MTCX/NOPB TSSOP PW 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 24-Aug-2017
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM2642MTCX/NOPB TSSOP PW 28 2500 367.0 367.0 38.0
PACKAGE MATERIALS INFORMATION
www.ti.com 24-Aug-2017
Pack Materials-Page 2
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