LTC6948IUFD-2#PBF - ANALOG DEVICES

LTC6948

6948fa

For more information www.linear.com/LTC6948

Typical applicaTion

FeaTures DescripTion

Ultralow Noise 0.37GHz to

6.39GHz Fractional-N Synthesizer

with Integrated VCO

The LT C

6948 is a high performance, low noise, 6.39GHz

phase-locked loop (PLL) with a fully integrated VCO,

including a reference divider, phase-frequency detector

(PFD), ultralow noise charge pump, fractional feedback

divider, and VCO output divider.

The fractional divider uses an advanced, 4th order Δ∑

modulator which provides exceptionally low spurious

levels. This allows wide loop bandwidths, producing

extremely low integrated phase noise values.

The programmable VCO output divider, with a range of 1

through 6, extends the output frequency range.

Output Frequency Options

LTC6948-1 LTC6948-2 LTC6948-3 LTC6948-4

O_DIV = 1 2.240 to 3.740 3.080 to 4.910 3.840 to 5.790 4.200 to 6.390

O_DIV = 2 1.120 to 1.870 1.540 to 2.455 1.920 to 2.895 2.100 to 3.195

O_DIV = 3 0.747 to 1.247 1.027 to 1.637 1.280 to 1.930 1.400 to 2.130

O_DIV = 4 0.560 to 0.935 0.770 to 1.228 0.960 to 1.448 1.050 to 1.598

O_DIV = 5 0.448 to 0.748 0.616 to 0.982 0.768 to 1.158 0.840 to 1.278

O_DIV = 6 0.373 to 0.623 0.513 to 0.818 0.640 to 0.965 0.700 to 1.065

6.3GHz Wideband Receiver

applicaTions

n Low Noise Fractional-N PLL with Integrated VCO

n No ∆-∑ Modulator Spurs

n 18-Bit Fractional Denominator

n –226 dBc/Hz Normalized In-Band Phase Noise Floor

n –274 dBc/Hz Normalized In-Band 1/f Noise

n –157 dBc/Hz Wideband Output Phase Noise Floor

n Excellent Integer Boundary Spurious Performance

n Output Divider (1 to 6, 50% Duty Cycle)

n Output Buffer Muting

n Reference Input Frequency Up to 425MHz

n Fast Frequency Switching

n FracNWizard™ Software Design Tool Support

n Wireless Basestations (LTE, WiMAX, W-CDMA, PCS)

n Microwave Data Links

n Military and Secure Radio

n Test and Measurement

L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and

FracNWizard is a trademark of Analog Devices, Inc. All other trademarks are the property of

their respective owners.

UNUSED OUTPUT

AVAILABLE FOR

OTHER USE

50Ω

0.1µF

R = 2, fPFD

= 50MHz

N = 84 TO 127.8

LBW = 186kHz

O = 1

0.01µF

1µF

0.01µF

3.3V

0.1µF

68nH

0.01µF

3.3V

76.8Ω

100pF

1.6nF

3.3V

fLO = 4200MHz TO 6390MHz IN 190.7Hz STEPS

LTC6948-4

STAT

TUNE

MUTE

SDI

VD+

LDO

RF+

RF–

GND

SCLK

GND

VRF+

REF+

SDO

CMC

CMB

CMA

GND

BVCO

VVCO+

GND

VCP+

VREF+

REF–

GND

0.1µF

100MHz

51.1Ω

1µF

3.3V

1µF

0.1µF

3.3V

1µF

2.2µF

15Ω

1µF

6948 TA01a

SPI BUS

10µH

RF INPUT SIGNAL

IF OUTPUT

IFLO

76.8Ω

56nF

2.4nF

LTC6948-4 Phase Noise,

fRF = 6236MHz

OFFSET FREQUENCY (Hz)

–140

PHASE NOISE (dBc/Hz)

–130

–110

–90

100 10k 100k 10M

40M

6946 TA01b

–150

1k 1M

–100

–120

–160

RMS NOISE = 0.412°

RMS JITTER = 183fs

fPFD = 50MHz

LOOP BW = 186kHz

INTN = 0

CPLE = 1

LTC6948

6948fa

For more information www.linear.com/LTC6948

pin conFiguraTionabsoluTe MaxiMuM raTings

Supply Voltages

V+ (VREF+, VRF+, VD+) to GND ............................... 3.6V

VCP+, VVCO+ to GND .............................................5.5V

Voltage on CP Pin .................GND – 0.3V to VCP+ + 0.3V

Voltage on all other Pins ...........GND – 0.3V to V+ + 0.3V

Operating Junction Temperature Range, TJ (Note 2)

LTC6948I ........................................... –40°C to 105°C

Junction Temperature, TJMAX ................................ 125°C

Storage Temperature Range .................. –65°C to 150°C

(Note 1)

9 10

TOP VIEW

GND

UFD PACKAGE

28-LEAD (4mm × 5mm) PLASTIC QFN

11 12 13

28 27 26 25 24

REF+

STAT

SCLK

SDI

SDO

LDO

VD+

BVCO

GND

CMA

CMB

CMC

GND

TUNE

REF–

VREF+

VCP+

GND

VVCO+

MUTE

GND

RF–

RF+

VRF+

815

TJMAX = 125°C, θJCbottom = 7°C/W

EXPOSED PAD (PIN 29) IS GND, MUST BE SOLDERED TO PCB

orDer inForMaTion

available opTions

LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION JUNCTION TEMPERATURE RANGE

LTC6948IUFD-1#PBF LTC6948IUFD-1#TRPBF 69481 28-Lead (4mm × 5mm) Plastic QFN –40°C to 105°C

LTC6948IUFD-2#PBF LTC6948IUFD-2#TRPBF 69482 28-Lead (4mm × 5mm) Plastic QFN –40°C to 105°C

LTC6948IUFD-3#PBF LTC6948IUFD-3#TRPBF 69483 28-Lead (4mm × 5mm) Plastic QFN –40°C to 105°C

LTC6948IUFD-4#PBF LTC6948IUFD-4#TRPBF 69484 28-Lead (4mm × 5mm) Plastic QFN –40°C to 105°C

Consult LTC Marketing for parts specified with wider operating temperature ranges.

Consult LTC Marketing for information on nonstandard lead based finish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/

VCO FREQUENCY

RANGE (GHz)

PACKAGE STYLE OUTPUT FREQUENCY RANGE vs OUTPUT DIVIDER SETTING (GHz)

QFN-28 (UFD28) 0 DIV = 6 0 DIV = 5 0 DIV = 4 0 DIV = 3 0 DIV = 2 0 DIV = 1

2.240 to 3.740 LTC6948IUFD-1 0.373 to 0.623 0.448 to 0.748 0.560 to 0.935 0.747 to 1.247 1.120 to 1.870 2.240 to 3.740

3.080 to 4.910 LTC6948IUFD-2 0.513 to 0.818 0.616 to 0.982 0.770 to 1.228 1.027 to 1.637 1.540 to 2.455 3.080 to 4.910

3.840 to 5.790 LTC6948IUFD-3 0.640 to 0.965 0.768 to 1.158 0.960 to 1.448 1.280 to 1.930 1.920 to 2.895 3.840 to 5.790

4.200 to 6.390 LTC6948IUFD-4 0.700 to 1.065 0.840 to 1.278 1.050 to 1.598 1.400 to 2.130 2.100 to 3.195 4.200 to 6.390

Overlapping Frequency Bands

http://www.linear.com/product/LTC6948#orderinfo

LTC6948

6948fa

For more information www.linear.com/LTC6948

elecTrical characTerisTics

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

Reference Inputs (REF+, REF–)

fREF Input Frequency l10 425 MHz

VREF Input Signal Level Single-Ended, 1µF AC-Coupling Capacitors l0.5 2 2.7 VP-P

Input Slew Rate l20 V/µs

Input Duty Cycle 50 %

Self-Bias Voltage l1.65 1.85 2.25 V

Input Resistance Differential l5.8 8.4 11.6 kΩ

Input Capacitance Differential 14 pF

VCO

fVCO Frequency Range LTC6948-1 (Note 3)

LTC6948-2 (Note 3)

LTC6948-3 (Note 3)

LTC6948-4 (Note 3)

2.24

3.08

3.84

4.20

3.74

4.91

5.79

6.39

GHz

KVCO Tuning Sensitivity LTC6948-1 (Notes 3, 4)

LTC6948-2 (Notes 3, 4)

LTC6948-3 (Notes 3, 4)

LTC6948-4 (Notes 3, 4)

4.7 to 7.2

4.7 to 7.0

4.0 to 6.0

4.5 to 6.5

%Hz/V

RF Output (RF+, RF–)

fRF Output Frequency l0.373 6.39 GHz

O Output Divider Range All Integers Included l1 6

Output Duty Cycle 50 %

Output Resistance Single-Ended, Each Output to VRF+l100 136 175 Ω

PRF-SE Output Power, Single-Ended, fRF = 900MHz RFO[1:0] = 0, RZ = 50Ω, LC Match

RFO[1:0] = 1, RZ = 50Ω, LC Match

RFO[1:0] = 2, RZ = 50Ω, LC Match

RFO[1:0] = 3, RZ = 50Ω, LC Match

–9

–6.1

–2.9

0.1

–7.3

–4.5

–1.4

1.5

–5.5

–2.8

0.2

dBm

Output Power, Muted, fRF = 900MHz RZ = 50Ω, Single-Ended, O = 2 to 6 l–80 dBm

Mute Enable Time l110 ns

Mute Disable Time l170 ns

Phase/Frequency Detector

fPFD Input Frequency Integer mode

Fractional mode

LDOEN = 0

LDOV = 3, LDOEN = 1

LDOV = 2, LDOEN = 1

LDOV = 1, LDOEN = 1

LDOV = 0, LDOEN = 1

100

76.1

66.3

56.1

45.9

34.3

MHz

Charge Pump

ICP Output Current Range 8 Settings (See Table 6) 1 11.2 mA

Output Current Source/Sink Accuracy All Settings, V(CP) = VCP+/2 ±6 %

Output Current Source/Sink Matching ICP = 1.0mA to 2.8mA, V(CP) = VCP+/2

ICP = 4.0mA to 11.2mA, V(CP) = VCP+/2

±3.5

±2

Output Current vs Output Voltage Sensitivity (Note 5) l0.2 1 %/V

Output Current vs Temperature V(CP) = VCP+/2 l170 ppm/°C

Output Hi-Z Leakage Current ICP = 11.2mA, CPCLO = CPCHI = 0 (Note 5) 0.03 nA

The l denotes the specifications which apply over the full operating

junction temperature range, otherwise specifications are at TA = 25°C. VREF+ = VD+ = VRF+ = 3.3V, VCP+ = VVCO+ = 5V unless otherwise

specified (Note 2). All voltages are with respect to GND.

LTC6948

6948fa

For more information www.linear.com/LTC6948

elecTrical characTerisTics

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

VCLMP-LO Low Clamp Voltage CPCLO = 1 0.84 V

VCLMP-HI High Clamp Voltage CPCHI = 1, Referred to VCP+–0.96 V

VMID Mid-Supply Output Bias Ratio Referred to (VCP+ – GND) 0.48 V/V

Reference (R) Divider

R Divide Range All Integers Included l1 31 Counts

VCO (N) Divider

N Divide Range All Integers Included, Integer Mode

All Integers Included, Fractional Mode

1023

1019

Counts

Fractional ∆∑ Modulator

Numerator Range All Integers Included l1 262143 Counts

Modulator LDO

Output Voltage LDO Enabled, Four Values

LDO Disabled

1.7 to 2.6

VD+V

External Pin Capacitance Required for LDO Stability l0.047 0.1 1 µF

Digital Pin Specifications

VIH High Level Input Voltage MUTE, CS, SDI, SCLK l1.55 V

VIL Low Level Input Voltage MUTE, CS, SDI, SCLK l0.8 V

VIHYS Input Voltage Hysteresis MUTE, CS, SDI, SCLK 250 mV

Input Current MUTE, CS, SDI, SCLK l±1 µA

IOH High Level Output Current SDO and STAT, VOH = VD+ – 400mV l–3.3 –1.9 mA

IOL Low Level Output Current SDO and STAT, VOL = 400mV l2.0 3.4 mA

SDO Hi-Z Current l±1 µA

Digital Timing Specifications (See Figure 6 and Figure 7)

tCKH SCLK High Time l25 ns

tCKL SCLK Low Time l25 ns

tCSS CS Setup Time l10 ns

tCSH CS High Time l10 ns

tCS SDI to SCLK Setup Time l6 ns

tCH SDI to SCLK Hold Time l6 ns

tDO SCLK to SDO Time To VIH/VIL/Hi-Z with 30pF Load l16 ns

Power Supply Voltages

VREF+ Supply Range l3.15 3.3 3.45 V

VD+ Supply Range l3.15 3.3 3.45 V

VRF+ Supply Range l3.15 3.3 3.45 V

VVCO+ Supply Range l4.75 5.0 5.25 V

VCP+ Supply Range l4.0 5.25 V

Power Supply Currents

IDD VD+ Supply Current Digital Inputs at Supply Levels, Integer Mode

Digital Inputs at Supply Levels, Fractional

Mode, fPFD = 66.3MHz, LDOV[1:0] = 3

18.2

1500

µA

ICC(5V) Sum VCP+, VVCO+ Supply Currents ICP = 11.2mA

ICP = 1.0mA

PDALL = 1

450

1000

µA

The l denotes the specifications which apply over the full operating

junction temperature range, otherwise specifications are at TA = 25°C. VREF+ = VD+ = VRF+ = 3.3V, VCP+ = VVCO+ = 5V unless otherwise

specified (Note 2). All voltages are with respect to GND.

LTC6948

6948fa

For more information www.linear.com/LTC6948

elecTrical characTerisTics

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 2: The LTC6948I is guaranteed to meet specified performance limits

over the full operating junction temperature range of –40°C to 105°C.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

ICC(3.3V) Sum VREF+, VRF+ Supply Currents RF Muted, OD[2:0] = 1

RF Enabled, RFO[1:0] = 0, OD[2:0] = 1

RF Enabled, RFO[1:0] = 3, OD[2:0] = 1

RF Enabled, RFO[1:0] = 3, OD[2:0] = 2

RF Enabled, RFO[1:0] = 3, OD[2:0] = 3

RF Enabled, RFO[1:0] = 3, OD[2:0] = 4 to 6

PDALL = 1

70.4

81.1

91.3

109.2

114.8

119.6

105

125

135

140

250

µA

Phase Noise and Spurious

LVCO VCO Phase Noise (LTC6948-1, fVCO = 3.0GHz,

fRF = 3.0GHz, OD[2:0] = 1 (Note 6))

10kHz Offset

1MHz Offset

40MHz Offset

–80

–130

–157

dBc/Hz

VCO Phase Noise (LTC6948-2, fVCO = 4.0GHz,

fRF = 4.0GHz, OD[2:0] = 1 (Note 6))

10kHz Offset

1MHz Offset

40MHz Offset

–77

–127

–156

dBc/Hz

VCO Phase Noise (LTC6948-3, fVCO = 5.0GHz,

fRF = 5.0GHz, OD[2:0] = 1 (Note 6))

10kHz Offset

1MHz Offset

40MHz Offset

–75

–126

–155

dBc/Hz

VCO Phase Noise (LTC6948-4, fVCO = 6.0GHz,

fRF = 6.0GHz, OD[2:0] = 1 (Note 6))

10kHz Offset

1MHz Offset

40MHz Offset

–73

–123

–154

dBc/Hz

VCO Phase Noise (LTC6948-3, fVCO = 5.0GHz,

fRF = 2.50GHz, OD[2:0] = 2 (Note 6))

10kHz Offset

1MHz Offset

40MHz Offset

–81

–132

–155

dBc/Hz

VCO Phase Noise (LTC6948-3, fVCO = 5.0GHz,

fRF = 1.667GHz, OD[2:0] = 3 (Note 6))

10kHz Offset

1MHz Offset

40MHz Offset

–84

–135

–156

dBc/Hz

VCO Phase Noise (LTC6948-3, fVCO = 5.0GHz,

fRF = 1.25GHz, OD[2:0] = 4 (Note 6))

10kHz Offset

1MHz Offset

40MHz Offset

–87

–138

–156

dBc/Hz

VCO Phase Noise (LTC6948-3, fVCO = 5.0GHz,

fRF = 1.00GHz, OD[2:0] = 5 (Note 6))

10kHz Offset

1MHz Offset

40MHz Offset

–89

–140

–157

dBc/Hz

VCO Phase Noise (LTC6948-3, fVCO = 5.0GHz,

fRF = 0.833GHz, OD[2:0] = 6 (Note 6))

10kHz Offset

1MHz Offset

40MHz Offset

–90

–141

–158

dBc/Hz

LNORM(INT) Integer Normalized In-Band Phase Noise

Floor

INTN = 1, ICP = 5.6mA (Notes 7, 8, 10) –226 dBc/Hz

LNORM(FRAC) Fractional Normalized In-Band Phase Noise

Floor

INTN = 0, CPLE = 1, ICP = 5.6mA

(Notes 7, 8, 10)

–225 dBc/Hz

L1/f Normalized In-Band 1/f Phase Noise ICP = 11.2mA (Notes 7, 11) –274 dBc/Hz

In-Band Phase Noise Floor Fractional Mode, CPLE = 1

(Notes 7, 9, 10, 12)

–113 dBc/Hz

Integrated Phase Noise from 100Hz to

40MHz

Fractional Mode, CPLE = 1 (Notes 9, 12) 0.14 °RMS

Spurious Fractional Mode, fOFFSET = fPFD, PLL locked

(Notes 8, 13, 14)

–98 dBc

The l denotes the specifications which apply over the full operating

junction temperature range, otherwise specifications are at TA = 25°C. VREF+ = VD+ = VRF+ = 3.3V, VCP+ = VVCO+ = 5V unless otherwise

specified (Note 2). All voltages are with respect to GND.

LTC6948

6948fa

For more information www.linear.com/LTC6948

TA = 25°C. VREF+ = VD+ = VRF+ = 3.3V, VCP+ = VVCO+ =

5V, INTN = 0, DITHEN = 1, CPLE = 1, RFO[1:0] = 3, unless otherwise noted.

Typical perForMance characTerisTics

elecTrical characTerisTics

Note 3: Valid for 1.60V ≤ V(TUNE) ≤ 2.85V with part calibrated after a

power cycle or software power-on reset (POR).

Note 4: Based on characterization.

Note 5: For 0.9V < V(CP) < (VCP+ – 0.9V).

Note 6: Measured outside the loop bandwidth, using a narrowband loop,

RFO[1:0] = 3.

Note 7: Measured inside the loop bandwidth with the loop locked.

Note 8: Reference frequency supplied by Wenzel 501-04516,

fREF = 100MHz, PREF = 10dBm.

Note 9: Reference frequency supplied by Wenzel 500-23571,

fREF = 61.44MHz, PREF = 10dBm.

Note 10: Output phase noise floor is calculated from normalized phase

noise floor by LOUT = LNORM + 10log10 (fPFD) + 20log10 (fRF/fPFD).

Note 11: Output 1/f noise is calculated from normalized 1/f phase noise by

LOUT(1/f) = L1/f + 20log10 (fRF) – 10log10 (fOFFSET).

Note 12: ICP = 5.6mA, fPFD = 61.44MHz, FILT[1:0] = 0, LoopBW=180kHz;

fRF = 2377.7MHz, fVCO = 4755.4MHz (LTC6948-3)

Note 13: ICP = 5.6mA, fPFD = 25MHz, FILT[1:0] = 0, LoopBW=73kHz;

fRF = 891.85MHz, fVCO = 2675Hz (LTC6948-1), fVCO = 4459MHz

(LTC6948-2), fVCO = 5351MHz (LTC6948-3, LTC6948-4).

Note 14: Measured using DC1959.

Charge Pump Sink Current Error

vs Voltage, Output Current

REF Input Sensitivity vs

Frequency

Charge Pump Sink Current Error

vs Voltage, Temperature

CP Hi-Z Current vs Voltage,

Temperature

Charge Pump Source Current

Error vs Voltage, Output Current

TUNE Current vs Voltage,

Temperature

FREQUENCY (MHz)

SENSITIVITY (dBm)

–15

–20

–25

–30

–35

–40

–45

–50

400

6948 G01

100 200 300

450

35050 150 250

BST = 1

FILT = 0

105°C

25°C

–40°C

OUTPUT VOLTAGE (V)

CURRENT (nA)

–1

–2

–3

–5

–4

4.0

6948 G02

1.0 2.0 3.0

5.0

3.50.5 1.5 2.5 4.5

ICP = 11.2mA

CPRST = 1

CPLE = 0

105°C

25°C

–40°C

VOLTAGE (V)

CURRENT (nA)

–1

–2

–3

–5

–4

6948 G03

1.0 2.0 3.00.5 1.5 2.5

105°C

25°C

–40°C

OUTPUT VOLTAGE (V)

ERROR (%)

–1

–2

–3

–5

–4

4.0

6948 G04

1.0 2.0 3.0

5.0

3.50.5 1.5 2.5 4.5

CPLE = 0

1mA

5.6mA

11.2mA

OUTPUT VOLTAGE (V)

ERROR (%)

–1

–2

–3

–5

–4

4.0

6948 G05

1.0 2.0 3.0

5.0

3.50.5 1.5 2.5 4.5

ICP = 11.2mA

CPLE = 0

105°C

25°C

–40°C

OUTPUT VOLTAGE (V)

ERROR (%)

–1

–2

–3

–5

–4

4.0

6948 G06

1.0 2.0 3.0 5.03.50.5 1.5 2.5 4.5

CPLE = 0

1mA

5.6mA

11.2mA

LTC6948

6948fa

For more information www.linear.com/LTC6948

TA = 25°C. VREF+ = VD+ = VRF+ = 3.3V, VCP+ = VVCO+ =

5V, INTN = 0, DITHEN = 1, CPLE = 1, RFO[1:0] = 3, unless otherwise noted.

Typical perForMance characTerisTics

RF Output HD3 vs Output Divide

(Single-Ended on RF+)

Charge Pump Source Current

Error vs Voltage, Temperature

MUTE Output Power vs fVCO and

Output Divide

(Single-Ended on RF+)

RF Output Power vs Frequency

(Single-Ended on RF+)

LTC6948-3 Frequency Step

Transient

RF Output HD2 vs Output Divide

(Single-Ended on RF+)

LTC6948-1 VCO Tuning Sensitivity LTC6948-2 VCO Tuning Sensitivity LTC6948-3 VCO Tuning Sensitivity

OUTPUT VOLTAGE (V)

ERROR (%)

4.0

6948 G07

–1

–3

–2

–4

–5 1.00.5 2.01.5 3.0 3.5 4.52.5

5.0

105°C

25°C

–40°C

ICP = 11.2mA

CPLE = 0

FREQUENCY (GHz)

0.5

OUT

(dBm)

–2.5

0.5

1.0

2.0

1.5

1.5 3.5 4.5

6948 G08

–0.5

–1.5

–3.0

–3.5

–1.0

–2.0

2.5 5.5

6.5

105°C

25°C

–40°C

LTC6948-4

LC = 68nH

CS = 100pF

fVCO (GHz)

4.2

–55

–50

HD2 (dBc)

–45

–40

–35

–30

5.0 5.8

6.6

6948 G09

–25

–20

4.6 5.4 6.2

LTC6948-4, LC = 68nH,

CS = 100pF , fRF = fVCO/O

O = 3

O = 2

O = 1

O = 6

O = 4

O = 5

4.2 5.0 5.8

6.6

4.6 5.4 6.2

fVCO (GHz)

–45

–40

HD3 (dBc)

–35

–30

–25

–20

6948 G10

–15

–10

–5

LTC6948-4

LC = 68nH

CS = 100pF

fRF = fVCO/O

O = 6

O = 3

O = 2

O = 1

O = 5

O = 4

4.2 5.0 5.8

6.6

4.6 5.4 6.2

fVCO (MHz)

OUT

AT f

VCO

/O (dBm)

–75

–65

–55

6948 G11

–85

–95

–45

LTC6948-4, LC = 68nH

CS = 100pF, fRF = fVCO/O

O = 1

O = 4

O = 6

O = 5

O = 2

O = 3

TIME (µs)

–5 0

2.2

FREQUENCY (GHz)

2.3

2.4

2.5

2.6

2.7

2.8

2.9

3.0

5 10 15 20

6948 G12

CALIBRATION TIME

126MHz STEP

fPFD = 61.44MHz

fCAL = 1.28MHz

LOOP BW = 180kHz

MTCAL = 0

O = 2

FREQUENCY (GHz)

2.2 2.4 2.6 2.8

3.0

VCO

(%/V)

3.5

4.5

5.0

5.5

8.0

6.5

6948 G13

4.0

7.0

7.5

6.0

3.8

3.63.43.23.0

FREQUENCY (GHz)

3.0 3.2 3.4 3.6 3.8

3.0

VCO

(%/V)

3.5

4.5

5.0

5.5

8.0

6.5

6948 G14

4.0

7.0

7.5

6.0

5.0

4.84.64.44.24.0

FREQUENCY (GHz)

3.8

2.5

VCO

(%/V)

3.0

4.0

4.5

5.0

7.0

6948 G15

3.5

4.2 5.4

4.6

5.8

5.5

6.0

6.5

LTC6948

6948fa

For more information www.linear.com/LTC6948

TA = 25°C. VREF+ = VD+ = VRF+ = 3.3V, VCP+ = VVCO+ =

5V, INTN = 0, DITHEN = 1, CPLE = 1, RFO[1:0] = 3, unless otherwise noted.

Typical perForMance characTerisTics

LTC6948-1 VCO Phase Noise LTC6948-2 VCO Phase Noise LTC6948-3 VCO Phase Noise

LTC6948-4 VCO Phase Noise

LTC6948-1 VCO Phase Noise vs

fVCO, Output Divide

(fOFFSET = 10kHz)

LTC6948-2 VCO Phase Noise vs

fVCO, Output Divide

(fOFFSET = 10kHz)

LTC6948-4 VCO Tuning Sensitivity

Normalized In-Band Phase Noise

Floor vs fVCO

Normalized In-Band Phase Noise

Floor vs CP Current

FREQUENCY (GHz)

4.3 4.7 5.1

3.0

VCO

(%/V)

3.5

4.5

5.0

5.5

7.5

6.5

6948 G16

4.0

7.0

6.0

6.7

6.35.95.5

OFFSET FREQUENCY (Hz)

–100

PHASE NOISE (dBc/Hz)

–90

–80

–70

–60

100k10k 1M 10M

40M

6948 G19

–110

–130

–150

–120

–140

–160

–50

–40

fRF = fVCO = 3GHz

OFFSET FREQUENCY (Hz)

–100

PHASE NOISE (dBc/Hz)

–90

–80

–70

–60

100k10k 1M 10M

40M

6948 G22

–110

–130

–150

–120

–140

–160

–50

–40

fRF = fVCO = 6GHz

fVCO (GHz)

2.2

–105

PHASE NOISE (dBc/Hz)

–100

–95

–90

–85

–75

2.4 2.6 2.8 3.0

6948 G23

3.2 3.4 3.6

3.8

–80

fRF = fVCO/O O = 1

O = 2

O = 3

O = 6

O = 5

O = 4

–100

–95

–90

–85

–80

–70

–75

fVCO (GHz)

3.0

PHASE NOISE (dBc/Hz)

3.3 3.6 3.9

6948 G24

4.2 4.5 4.8

fRF = fVCO/O

O = 1

O = 2

O = 3

O = 6

O = 4

O = 5

OFFSET FREQUENCY (Hz)

–100

PHASE NOISE (dBc/Hz)

–90

–80

–70

–60

100k10k 1M 10M

40M

6948 G20

–110

–130

–150

–120

–140

–160

–50

–40

fRF = fVCO = 4GHz

OFFSET FREQUENCY (Hz)

–100

PHASE NOISE (dBc/Hz)

–90

–80

–70

–60

100k10k 1M 10M

40M

6948 G21

–110

–130

–150

–120

–140

–160

–50

–40

fRF = fVCO = 5GHz

fVCO (GHz)

–227

PHASE NOISE FLOOR (dBc/Hz)

–226

–224

6948 G17

–222

–225

–223

5 6

ICP = 5.6mA

CPLE = 1

FRACTIONAL-N

INTEGER-N

ICP (mA)

–227

PHASE NOISE FLOOR (dBc/Hz)

–226

3 5

6948 G18

–224

–222

–225

–223

fVCO = 5GHz

CPLE = 1

FRACTIONAL-N

INTEGER-N

LTC6948

6948fa

For more information www.linear.com/LTC6948

TA = 25°C. VREF+ = VD+ = VRF+ = 3.3V, VCP+ = VVCO+ =

5V, INTN = 0, DITHEN = 1, CPLE = 1, RFO[1:0] = 3, unless otherwise noted.

Typical perForMance characTerisTics

LTC6948-2 VCO Phase Noise vs

fVCO, Output Divide

(fOFFSET = 1MHz)

LTC6948-3 VCO Phase Noise vs

fVCO, Output Divide

(fOFFSET = 1MHz)

LTC6948-4 VCO Phase Noise vs

fVCO, Output Divide

(fOFFSET = 1MHz)

Closed-Loop Phase Noise,

LTC6948-1 fRF = 891.857MHz

Closed-Loop Phase Noise,

LTC6948-3 fRF = 2377.728MHz

Closed-Loop Phase Noise,

LTC6948-2 fRF = 3646.464MHz

LTC6948-3 VCO Phase Noise vs

fVCO, Output Divide

(fOFFSET = 10kHz)

LTC6948-4 VCO Phase Noise vs

fVCO, Output Divide

(fOFFSET = 10kHz)

LTC6948-1 VCO Phase Noise vs

fVCO, Output Divide

(fOFFSET = 1MHz)

–150

–145

–140

–135

–125

–130

fVCO (GHz)

2.2

PHASE NOISE (dBc/Hz)

2.4 2.6 2.8 3.0

6948 G27

3.2 3.4 3.6

3.8

fRF = fVCO/O

O = 1

O = 2

O = 3

O = 6

O = 4

O = 5

–80

–75

–70

–85

–90

–95

fVCO (GHz)

3.8

PHASE NOISE (dBc/Hz)

4.2

6948 G25

4.6 5.0 5.4

5.8

fRF = fVCO/O

O = 1

O = 2

O = 3

O = 6

O = 4

O = 5

–130

–125

–135

–140

–145

fVCO (GHz)

3.0

PHASE NOISE (dBc/Hz)

3.3 3.6 3.9

6948 G28

4.2 4.5 4.8

fRF = fVCO/O

O = 1

O = 2

O = 3

O = 6

O = 4

O = 5

–130

–125

–120

–135

–140

–145

fVCO (GHz)

3.8

PHASE NOISE (dBc/Hz)

4.2

6948 G29

4.6 5.0 5.4

5.8

fRF = fVCO/O

O = 1

O = 2

O = 3

O = 6

O = 4

O = 5

4.3 4.7 5.1 5.5 5.9

6.7

6.3

–130

–125

–120

–135

–140

–145

fVCO (GHz)

PHASE NOISE (dBc/Hz)

6948 G30

fRF = fVCO/O

O = 1

O = 2

O = 3

O = 6

O = 4

O = 5

–80

–75

–70

–85

–90

–95

fVCO (GHz)

4.3

PHASE NOISE (dBc/Hz)

4.7

6948 G26

5.1 5.5 5.9

6.7

6.3

fRF = fVCO/O

O = 1

O = 2

O = 3

O = 6

O = 4

O = 5

OFFSET FREQUENCY (Hz)

–140

PHASE NOISE (dBc/Hz)

–130

–110

–90

100 10k 100k 10M

40M

6948 G31

–150

1k 1M

–100

–120

–160

RMS NOISE = 0.074°

fPFD = 25MHz

O = 3

INTN = 0

CPLE = 1

fIB-SPUR = 572kHz

LOOP BW = 73kHz

NOTES 8, 13

OFFSET FREQUENCY (Hz)

–140

PHASE NOISE (dBc/Hz)

–130

–110

–90

100 10k 100k 10M

40M

6948 G32

–150

1k 1M

–100

–120

–160

RMS NOISE = 0.141°

fPFD = 61.44MHz

O = 2

INTN = 0

CPLE = 1

LOOP BW = 180kHz

NOTES 9, 12

OFFSET FREQUENCY (Hz)

–140

PHASE NOISE (dBc/Hz)

–130

–110

–90

100 10k 100k 10M

40M

6948 G33

–150

1k 1M

–100

–120

–160

RMS NOISE = 0.224°

fPFD = 61.44MHz

O = 1

INTN = 0

CPLE = 1

LOOP BW = 178kHz

NOTE 9

LTC6948

6948fa

For more information www.linear.com/LTC6948

TA = 25°C. VREF+ = VD+ = VRF+ = 3.3V, VCP+ = VVCO+ =

5V, INTN = 0, DITHEN = 1, CPLE = 1, RFO[1:0] = 3, unless otherwise noted.

Typical perForMance characTerisTics

Integer Boundary Spur Power vs

Output Frequency, LTC6948-3 LTC6948-4 Supply Current

VD+ Supply Current vs LDOV, fPFD

(INTN = 0, PDFN = 0)

VD+ Supply Current vs LDOV,

Temperature

(INTN = 0, PDFN = 0, fPFD Noted)

LTC6948-1 Spurious Response

fRF = 891.85MHz, fREF = 100MHz,

fPFD = 25MHz, Loop BW = 74kHz

LTC6948-3 Spurious Response

fRF = 2377.73MHz, fREF = 61.44MHz,

fPFD = 61.44MHz, Loop BW = 180k

LTC6948-4 Spurious Response

fRF = 6236MHz, fREF = 100MHz,

fPFD = 50MHz, Loop BW = 152kHz

FREQUENCY OFFSET (MHz IN 10kHz SEGMENTS)

–100

–100dBc

–75

–140

OUT

(dBm)

–120

–100

–80

25 50 75

6948 G34

–50 –25 0 100

–60

–40

–20

RBW = 10Hz

VBW = 10Hz

INTN = 0

CPLE = 1

O = 3

NOTES 8, 14

–103dBc

OUTPUT FREQUENCY (GHz)

–80

IB SPUR LEVEL (dBc)

–75

–70

–65

–60

–40

0.5 1.5 2.5

6948 G37

3.5 4.51 2 3 4 5 5.5

–55

–50

–45

O = 1

O = 2

O = 3

O = 6

O = 5

O = 4

SPUR IN BAND

fPFD = 50MHz

CPLE = 1

fPFD (MHz)

5 15

SUPPLY CURRENT (mA)

25 35 45 55

6948 G39

65 75

LDOEN = 0

LDOV = 3

LDOV = 2

LDOV = 1

LDOV = 0

TC = 25°C

TJ (°C)

–40

3.3V CURRENT (mA)

5V CURRENT (mA)

–20 020 40

6948 G38

80 100

O = 1, MUTE = 0

RFO = 3, ICP = 5.6mA

EXCLUDES VD+

TJ (°C)

–40 –20

SUPPLY CURRENT (mA)

0 20 40 60

6948 G40

100

LDOEN = 0, 75MHz

LDOV = 3, 65MHz

LDOV = 2, 55MHz

LDOV = 1, 45MHz

LDOV = 0, 30MHz

FREQUENCY OFFSET (MHz IN 10kHz SEGMENTS)

–246

–96dBc

–78dBc–78dBc

–184

–140

OUT

(dBm)

–120

–100

–80

61.4 123 184

6948 G35

–123–61.4 0 246

–60

–40

–20

RBW = 100Hz

VBW = 100Hz

INTN = 0

CPLE = 1

O = 2

NOTES 9, 14

–91dBc

FREQUENCY OFFSET (MHz IN 10kHz SEGMENTS)

–200

–70dBc

–150

–140

OUT

(dBm)

–120

–100

–80

50 100 150

6948 G36

–100 –50 0 200

–60

–40

–20

RBW = 100Hz

VBW = 100Hz

INTN = 0

CPLE = 1

O = 1

NOTES 8, 14

–72dBc

LTC6948

6948fa

For more information www.linear.com/LTC6948

pin FuncTions

REF+, REF– (Pins 1, 28): Reference Input Signals. This

differential input is buffered with a low noise amplifier,

which feeds the reference divider. They are self-biased and

must be AC-coupled with 1µF capacitors. If used single-

ended with VREF+ ≤ 2.7VP-P, bypass REF– to GND with a

1µF capacitor. If used single-ended with VREF+ > 2.7VP-P,

bypass REF– to GND with a 47pF capacitor.

STAT (Pin 2): Status Output. This signal is a configurable

logical OR combination of the UNLOK, LOK, ALCHI, ALCLO,

THI, and TLO status bits, programmable via the STATUS

CS (Pin 3): Serial Port Chip Select. This CMOS input initi-

ates a serial port communication burst when driven low,

ending the burst when driven back high. See the Operation

section for more details.

SCLK (Pin 4): Serial Port Clock. This CMOS input clocks

serial port input data on its rising edge. See the Operation

section for more details.

SDI (Pin 5): Serial Port Data Input. The serial port uses

this CMOS input for data. See the Operation section for

more details.

SDO (Pin 6): Serial Port Data Output. This CMOS three-

state output presents data from the serial port during a

read communication burst. Optionally attach a resistor of

>200k to GND to prevent a floating output. See the Ap-

plications Information section for more details.

LDO (Pin 7): Δ∑ Modulator LDO Bypass Pin. This pin

should be bypassed directly to the ground plane using a

low ESR (<0.8Ω) 0.1µF ceramic capacitor as close to the

pin as possible.

VD+ (Pin 8): 3.15V to 3.45V Positive Supply Pin for Serial

Port and Δ∑ Modulator Circuitry. This pin should be by-

passed directly to the ground plane using a 0.1µF ceramic

capacitor as close to the pin as possible.

MUTE (Pin 9): RF Mute. The CMOS active-low input mutes

the RF± differential outputs while maintaining internal bias

levels for quick response to deassertion.

GND (Pins 10, 17, 21, Exposed Pad Pin 29): Negative

Power Supply (Ground). These pins should be tied directly

to the ground plane with multiple vias for each pin. The

package exposed pad must be soldered directly to the PCB

land. The PCB land pattern should have multiple thermal

vias to the ground plane for both low ground inductance

and also low thermal resistance.

RF–, RF+ (Pins 11, 12): RF Output Signals. The VCO

output divider is buffered and presented differentially on

these pins. The outputs are open-collector, with 136Ω

(typical) pull-up resistors tied to VRF+ to aid impedance

matching. If used single-ended, the unused output should

be terminated to 50Ω. See the Applications Information

section for more details on impedance matching.

VRF+ (Pin 13): 3.15V to 3.45V Positive Supply Pin for

RF circuitry. This pin should be bypassed directly to the

ground plane using a 0.01µF ceramic capacitor as close

to the pin as possible.

BB (Pin 14): RF Reference Bypass. This output has a 2.5k

resistance and must be bypassed with a 1µF ceramic ca-

pacitor to GND. Do not couple this pin to any other signal.

TUNE (Pin 15): VCO Tuning Input. This frequency control

pin is normally connected to the external loop filter. See

the Applications Information section for more details.

LTC6948

6948fa

For more information www.linear.com/LTC6948

TB (Pin 16): VCO Bypass. This output has a 2k resistance

and must be bypassed with a 2.2µF ceramic capacitor to

GND. It is normally connected to CMA, CMB, and CMC with

a short trace. Do not couple this pin to any other signal.

CMA, CMB, CMC (Pins 18, 19, 20): VCO Bias Inputs. These

inputs are normally connected to TB with a short trace

and bypassed with a 2.2µF ceramic capacitor to GND. Do

not couple these pins to any other signal. For best phase

noise performance, DO NOT place a trace between these

pads underneath the package.

BVCO (Pin 22): VCO Bypass Pin. This output must be

bypassed with a 1µF ceramic capacitor to GND. Do not

couple this pin to any other signal.

VVCO+ (Pin 23): 4.75V to 5.25V Positive Supply Pin for

VCO Circuitry. This pin should be bypassed directly to the

ground plane using a 0.01µF ceramic capacitor as close

to the pin as possible.

GND (24): Negative Power Supply (Ground). This pin is

attached directly to the die attach paddle (DAP) and should

be tied directly to the ground plane.

VCP+ (Pin 25): 4V to 5.25V Positive Supply Pin for Charge

Pump Circuitry. This pin should be bypassed directly to

the ground plane using a 0.1µF ceramic capacitor as close

to the pin as possible.

CP (Pin 26): Charge Pump Output. This bidirectional cur-

rent output is normally connected to the external loop filter.

See the Applications Information section for more details.

VREF+ (Pin 27): 3.15V to 3.45V Positive Supply Pin for

Reference Input Circuitry. This pin should be bypassed

directly to the ground plane using a 0.1µF ceramic capaci-

tor as close to the pin as possible.

pin FuncTions

LTC6948

6948fa

For more information www.linear.com/LTC6948

block DiagraM

RF–

GND

MUTE

929

RF+

VRF+

REF+

LDO

REGULATOR

EXPOSED

PAD

≤425MHz

≤100MHz

÷1 TO 31

÷1 TO 6, 50%

1.7V TO 2.6V

÷32 TO 1023

MUTE

REF–

VREF+

R_DIV

LOCK

PFD

O_DIV

N_DIV

∆∑

B_DIV

CAL, ALC

CONTROL

2.24GHz TO 3.74GHz (LTC6948-1) OR

3.08GHz TO 4.91GHz (LTC6948-2) OR

3.84GHz TO 5.79GHz (LTC6948-3) OR

4.20GHz TO 6.390GHz (LTC6948-4)

373MHz TO 6.39GHz TUNE 15

VVCO+

GND

CMA

CMB

CMC

GND

1mA TO

11.2mA 26

VCP+

GND

BVCO

6948 BD

SERIAL

PORT

STAT

6SDO

SDI

SCLK

8VD+

–

LTC6948

6948fa

For more information www.linear.com/LTC6948

operaTion

The LTC6948 is a high-performance fractional-N PLL

complete with a low noise VCO available in four different

frequency range options. The output frequency range

may be further extended by utilizing the output divider

(see Available Options table for more details). The device

is able to achieve superior integrated phase noise by the

combination of its extremely low in-band phase noise

performance and excellent VCO noise characteristics.

The fractional-N feedback divider uses an advanced Δ∑

modulator, resulting in virtually no discrete modulator

spurious tones. The modulator may be disabled if integer-N

feedback is required.

REFERENCE INPUT BUFFER

The PLL’s reference frequency is applied differentially on

pins REF+ and REF–. These high impedance inputs are

self-biased and must be AC-coupled with 1µF capacitors

(see Figure 1 for a simplified schematic). Alternatively, the

inputs may be used single-ended by applying the refer-

ence frequency at REF+ and bypassing REF– to GND with

a 1µF capacitor. If the single-ended signal is greater than

2.7VP-P, then use a 47pF capacitor for the GND bypass.

Additional options are available through serial port register

h0B to further refine the application. Bits FILT[1:0] control

the reference input buffer’s lowpass filter, and should be

set based upon fREF to limit the reference’s wideband

noise. The FILT[1:0] bits must be set correctly to reach

the LNORM normalized in-band phase noise floor. See Table

1 for recommended settings.

Table 1. FILT[1:0] Programming

FILT[1:0] fREF

3 <20MHz

2 NA

1 20MHz to 50MHz

0 >50MHz

The BST bit should be set based upon the input signal level

to prevent the reference input buffer from saturating. See

Table 2 for recommended settings and the Applications

Information section for programming examples.

Table 2. BST Programming

BST VREF

1 <2VP-P

0 ≥2VP-P

REFERENCE (R) DIVIDER

A 5-bit divider, R_DIV, is used to reduce the frequency

seen at the PFD. Its divide ratio R may be set to any inte-

ger from 1 to 31, inclusive. Use the RD[4:0] bits found in

registers h06 to directly program the R divide ratio. See

the Applications Information section for the relationship

between R and the fREF, fPFD, fVCO, and fRF frequencies.

PHASE/FREQUENCY DETECTOR (PFD)

The phase/frequency detector (PFD), in conjunction with

the charge pump, produces source and sink current pulses

proportional to the phase difference between the outputs

of the R and N dividers. This action provides the necessary

feedback to phase-lock the loop, forcing a phase align-

Figure 1. Simplified REF Interface Schematic

4.2k

REF+

REF–

4.2k

6948 F01

1.9V

BST

BIAS

VREF

LOWPASS

FILT[1:0]

A high quality signal must be applied to the REF± inputs

as they provide the frequency reference to the entire PLL.

To achieve the part’s in-band phase noise performance,

apply a CW signal of at least 6dBm into 50Ω, or a square

wave of at least 0.5VP-P with slew rate of at least 40V/µs.

LTC6948

6948fa

For more information www.linear.com/LTC6948

operaTion

Figure 2. Simplified PFD Schematic

LOCK INDICATOR

The lock indicator uses internal signals from the PFD to

measure phase coincidence between the R and N divider

output signals. It is enabled by programming LKCT[1:0]

in the serial port register h0C (see Table 5), and produces

both LOCK and UNLOCK status flags, available through

both the STAT output and serial port register h00.

The user sets the phase difference lock window time tLWW

for a valid LOCK condition with the LKWIN[2:0] bits. When

using the device as a fractional-N synthesizer (fractional

mode), the Δ∑ modulator changes the instantaneous phase

seen at the PFD on every R_DIV and N_DIV cycle. The

maximum allowable time difference in this case depends

upon both the VCO frequency fVCO and also the charge

pump linearization enable bit CPLE (see the Charge Pump

Linearizer section for an explanation of this function). Table

3 contains recommended settings for LKWIN[2:0] when

using the device in fractional mode. See the Applications

Information section for examples.

D Q

RST

N DIV

D Q

RST

CPRST

DOWN

6948 F02

DELAY

R DIV

Table 3. LKWIN[2:0] Fractional Mode Programming

LKWIN[2:0] tLWW fVCO (CPLE = 1) fVCO (CPLE = 0)

0 5.0ns ≥2.97GHz ≥1.35GHz

1 7.35ns ≥2.00GHz ≥919MHz

2 10.7ns ≥1.39GHz ≥632MHz

3 15.8ns ≥941MHz ≥428MHz

4 23.0ns ≥646MHz ≥294MHz

5 34.5ns ≥431MHz ≥196MHz

6 50.5ns ≥294MHz ≥134MHz

7 76.0ns ≥196MHz ≥89MHz

When using the device as an integer-N synthesizer (integer

mode), the phase difference seen at the PFD is minimized

by the feedback of the PLL and no longer depends upon

fVCO. Table 4 contains recommended settings for different

fPFD frequencies. See the Applications Information section

for examples.

Table 4. LKWIN[2:0] Integer Mode Programming

LKWIN[2:0] tLWW fPFD

0 5.0ns >6.8MHz

1 7.35ns ≤6.8MHz

2 10.7ns ≤4.7MHz

3 15.8ns ≤3.2MHz

4 23.0ns ≤2.2MHz

5 34.5ns ≤1.5MHz

6 50.5ns ≤1.0MHz

7 76.0ns ≤660kHz

The PFD phase difference must be less than tLWW for the

COUNTS number of successive counts before the lock

indicator asserts the LOCK flag. The LKCT[1:0] bits found

in register h0C are used to set COUNTS depending upon

the application. Set LKCT to 0 to disable the lock indicator.

See Table 5 for LKCT[1:0] programming and the Applica-

tions Information section for examples.

ment at the PFD’s inputs. The PFD may be disabled with

the CPRST bit which prevents UP and DOWN pulses from

being produced. See Figure 2 for a simplified schematic

of the PFD.

LTC6948

6948fa

For more information www.linear.com/LTC6948

operaTion

Figure 4. Simplified Charge Pump Schematic

Figure 3. UNLOCK and LOCK Timing

+tLWW

–tLWW

UNLOCK FLAG

LOCK FLAG t = COUNTS/fPFD

6948 F03

PHASE

DIFFERENCE

AT PFD

–

ICP

THI

0.9V

VCP

TLO

–

0.9V

6948 F04

–

VCP+/2

CP LINEARIZER

CONTROL

ENABLE

CPMID

CPUP

ILIN

CPDN

CP[2:0]

CPINV

DOWN

ND[9:0]

INTN

CPLE

VCP+

Table 5. LKCT[1:0] Programming

LKCT[1:0] COUNTS

0 Lock Indicator Disabled

1 32

2 256

3 2048

When the PFD phase difference is greater than tLWW, the

lock indicator immediately asserts the UNLOCK status

flag and clears the LOCK flag, indicating an out-of-lock

condition. The UNLOCK flag is immediately de-asserted

when the phase difference is less than tLWW. See Figure3

for more details.

Note that fREF must be present for the LOCK and UNLOCK

flags to properly assert and clear.

to the lower impedance of the loop filter components,

although currents larger than 5.6mA typically cause worse

spurious performance. See Table 6 for programming

specifics and the Applications Information section for

loop filter examples.

Table 6. CP[2:0] Programming

CP[2:0] ICP

0 1.0mA

1 1.4mA

2 2.0mA

3 2.8mA

4 4.0mA

5 5.6mA

6 8.0mA

7 11.2mA

The CPINV bit found in register h0D should be set for ap-

plications requiring signal inversion from the PFD, such as

for complex external loops using an inverting op amp. A

passive loop filter as shown in Figure 13 requires CPINV = 0.

CHARGE PUMP

The charge pump, controlled by the PFD, forces sink

(DOWN) or source (UP) current pulses onto the CP pin,

which should be connected to an appropriate loop filter.

See Figure 4 for a simplified schematic of the charge pump.

The output current magnitude ICP may be set from 1mA to

11.2mA using the CP[2:0] bits found in serial port register

h0C. A larger ICP can result in lower in-band noise due

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Charge Pump Functions

The charge pump contains additional features to aid in

system startup. See Table 7 below for a summary.

Table 7. Charge Pump Function Bit Descriptions

BIT DESCRIPTION

CPCHI Enable High Voltage Output Clamp

CPCLO Enable Low Voltage Output Clamp

CPDN Force Sink Current

CPINV Invert PFD Phase

CPLE Linearizer Enable

CPMID Enable Mid-Voltage Bias

CPRST Reset PFD

CPUP Force Source Current

CPWIDE Extend Current Pulse Width

THI High Voltage Clamp Flag

TLO Low Voltage Clamp Flag

The CPCHI and CPCLO bits found in register h0D enable

the high and low voltage clamps, respectively. When CPCHI

is enabled and the CP pin voltage exceeds approximately

VCP+ – 0.9V, the THI status flag is set, and the charge

pump sourcing current is disabled. Alternately, when

CPCLO is enabled and the CP pin voltage is less than

approximately 0.9V, the TLO status flag is set, and the

charge pump sinking current is disabled. See Figure4 for

a simplified schematic.

The CPMID bit also found in register h0D enables a resis-

tive VCP+/2 output bias which may be used to pre-bias

troublesome loop filters into a valid voltage range. When

using CPMID, it is recommended to also assert the CPRST

bit, forcing a PFD reset. Both CPMID and CPRST must be

set to 0 for normal operation.

The CPUP and CPDN bits force a constant ICP source or

sink current, respectively, on the CP pin. The CPRST bit

may also be used in conjunction with the CPUP and CPDN

bits, allowing a precharge of the loop to a known state,

if required. CPUP, CPDN, and CPRST must be set to 0 to

allow the loop to lock.

The CPWIDE bit extends the charge pump output current

pulse width by increasing the PFD reset path’s delay value

(see Figure 2). CPWIDE is normally set to 0.

Charge Pump Linearizer

When the LTC6948 is operated in fractional mode, the

charge pump’s current output versus its phase stimulus

(its gain linearity) must be extremely accurate. The CP

gain linearizer automatically adds a correction current ILIN

to minimize the charge pump’s impact on in-band phase

noise and spurious products during fractional operation.

The CP gain linearizer is enabled by setting CPLE = 1. It is

automatically disabled when in integer mode. CPLE should

be set to 0 if CPRST or CPMID are asserted to prevent the

linearizer from producing unintended currents.

VCO

The integrated VCO is available in one of four frequency

ranges. The output frequency range may be further ex-

tended by utilizing the output divider (see Available Options

table, for more details). The wide frequency range of the

VCO, coupled with the output divider capability, allows the

LTC6948 to cover an extremely wide range of continuously

selectable frequencies.

The BB and TB pins are used to bias internal VCO circuitry.

The BB pin has a 2k output resistance and should be

bypassed with a 1µF ceramic capacitor to GND, giving

a time constant of 2ms. The TB pin has a 2.5k output

resistance and should be bypassed with a 2.2µF ceramic

capacitor to GND, resulting in a time constant of 5.5ms.

Stable bias voltages are achieved after approximately 3

time constants following power-up.

VCO Calibration

The VCO must be calibrated each time its frequency is

changed by changing any of fREF, the R divider value, the

N divider value, or Δ∑ modulator fractional value, but not

the O divider (see the Applications Information section

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for the relationship between R, N, NUM, O, and the fREF,

fPFD, fVCO, and fRF frequencies). The output frequency is

then stable over the LTC6948’s entire temperature range,

regardless of the temperature at which it was calibrated,

until the part is reset due to a power cycle or software

power-on reset (POR).

The output of the B divider is used to clock digital calibra-

tion circuitry as shown in the Block Diagram. The B value,

programmed with bits BD[3:0], and fPFD determine the

calibration frequency fCAL. Calculate the B value using

Equation 1.

B≥

PFD

fCAL-MAX

(1)

The maximum calibration frequency fCAL-MAX for each part

option is shown in Table 8.

Table 8. Maximum Calibration Frequency

PART fCAL-MAX (MHz)

LTC6948-1 1.0

LTC6948-2 1.33

LTC6948-3 1.7

LTC6948-4 1.8

The relationship between bits BD(3:0) and the B value is

shown in Table 9.

Table 9. BD[3:0] Programming

BD[3:0] B DIVIDE VALUE

0 8

1 12

2 16

3 24

4 32

5 48

6 64

7 96

8 128

9 192

10 256

11 384

12 to 15 Invalid

Once the RD[4:0], ND[9:0], NUM[17:0], and BD[3:0] bits

are written and the reference frequency fREF is present and

stable at the REF± inputs, the VCO must be calibrated by

setting CAL = 1 (the bit is self-clearing). The calibration

cycle takes between 12 and 14 clocks of the B divider

output. The MTCAL bit may be set to mute the RF output

during the calibration.

Note that the fREF frequency and TB and BB voltages must

be stable for proper calibration. Stable bias voltages are

achieved after approximately 3 time constants following

power-up.

Setting AUTOCAL = 1 causes the CAL bit to be set auto-

matically whenever any of serial port registers h06 through

h0A is written. When AUTOCAL is enabled, there is no

need for a separate register write to set the CAL bit. See

Table 10 for a summary of the VCO bits.

Table 10. VCO Bit Descriptions

BIT DESCRIPTION

AUTOCAL Calibrate VCO Whenever Registers h06 to h0A Are Written

CAL Start VCO Calibration (Auto Clears)

MTCAL Mute RF Output During Calibration

VCO Automatic Level Control (ALC)

The VCO uses an internal automatic level control (ALC)

algorithm to maintain an optimal amplitude on the VCO

resonator, and thus optimal phase noise performance. The

user has several ALC configuration and status reporting

options as seen in Table 11.

Table 11. ALC Bit Descriptions

BIT DESCRIPTION

ALCCAL Auto Enable ALC during CAL operation

ALCEN Always Enable ALC (Overrides ALCCAL, ALCMON, and

ALCULOK)

ALCHI ALC too High Flag (Resonator Amplitude too High)

ALCLO ALC too Low Flag (Resonator Amplitude too Low)

ALCMON Enable ALC Monitoring for Status Flags Only; Does Not Enable

Amplitude Control

ALCULOK Auto Enable ALC when PLL Unlocked

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Changes in the internal ALC output can cause extremely

small jumps in the VCO frequency. These jumps may be

acceptable in some applications but not in others. Use the

above table to choose when the ALC is active. The ALCHI

and ALCLO flags, valid only when the ALC is active or the

ALCMON bit is set, may be used to monitor the resonator

amplitude.

The ALC must be allowed to operate during or after a

calibration cycle. At least one of the ALCCAL, ALCEN, or

ALCULOK bits must be set.

VCO (N) DIVIDER

The 10-bit N divider provides the feedback from the VCO to

the PFD. Its divide ratio N is restricted to any integer from

35 to 1019, inclusive, when in fractional mode. The divide

ratio may be programmed from 32 to 1023, inclusive, when

in integer mode. Use the ND[9:0] bits found in registers

h06 and h07 to directly program the N divide ratio. See

the Applications Information section for the relationship

between N and the fREF, fPFD, fVCO, and fRF frequencies.

∆∑ MODULATOR

The Δ∑ modulator changes the N divider’s ratio each PFD

cycle to achieve an average fractional divide ratio. The

fractional numerator NUM[17:0] is programmable from

1 to 262143, or 218 – 1. The fractional denominator is

fixed at 262144 (or 218), with the resulting fractional ratio

F given by Equation 4. See the Applications Information

section for the relationship between NUM, F, and the fREF,

fPFD, fVCO, and fRF frequencies.

The Δ∑ modulator uses digital signal processing (DSP)

techniques to achieve an average fractional divide ratio.

The modulator is clocked at the fPFD rate. This process

produces output modulation noise known as quantization

noise with a highpass frequency response. The external

lowpass loop filter is used to filter this quantization noise to

a level beneath the phase noise of the VCO. This prevents

the noise from contributing to the overall phase noise of

the system. The loop filter must be designed to adequately

filter the quantization noise.

The oversampling ratio OSR is defined as the ratio of the

Δ∑ modulator clock frequency fPFD to the loop bandwidth

BW of the PLL (see Equation 11). See the Applications

Information section for guidelines concerning the OSR

and the loop filter.

When the desired output frequency is such that the needed

NUM value is 0, the LTC6948 should be operated in integer

mode (INTN = 1). In integer mode, the modulator is placed

in standby, with all blocks still powered up, thus allowing

it to resume fractional operation immediately.

Enable numerator dither mode (DITHEN = 1) to further

reduce spurious produced by the modulator. Dither has no

measurable impact on in-band phase noise, and is enabled

by default. See Table 12 for a complete list of modulator

bit descriptions.

Modulator Reset

To achieve consistent spurious performance, the modulator

DSP circuitry should be re-initialized by setting RSTFN=1

whenever NUM[17:0] is changed. Setting AUTORST = 1

causes the RSTFN bit to be set automatically whenever

any of serial port registers h05 through h0A are written.

When AUTORST is enabled, there is no need for a sepa-

rate register write to set the RSTFN bit. See Table 12 for

a summary of the modulator bits.

Table 12. Fractional Modulator Bit Descriptions

BIT DESCRIPTION

AUTORST Automatically Reset Modulator when Registers h05 to h0A

Are Written

DITHEN Enable Fractional Numerator Dither

INTN Integer Mode; Fractional Modulator Placed in Standby

RSTFN Reset Modulator (Auto Clears)

SEED Seed Value for Pseudorandom Dither Algorithm

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LDO REGULATOR

The adjustable low dropout (LDO) regulator supplies power

to the Δ∑ modulator. The regulator requires a low ESR

ceramic capacitor (ESR < 0.8Ω) connected to the LDO

pin (pin 7) for stability. The capacitor value may range

from 0.047µF to 1µF.

The LDO voltage is set using the LDOV[1:0] bits, and should

be chosen based upon the fPFD frequency to minimize

power and spurious. The regulator is disabled by setting

the LDOEN bit to 0. When disabled by using either the

LDOEN or PDFN bits, the LDO pin is connected directly

to VD+ using a low impedance switch, and the regulator

is powered down. See Table 13 for programming details.

Table 13. LDOV[1:0] and LDOEN Programming

LDOV[1:0] LDOEN VLDO fPFD

0 1 1.7V ≤34.3MHz

1 1 2.0V ≤45.9MHz

2 1 2.3V ≤56.1MHz

3 1 2.6V ≤66.3MHz

X 0 VD+≤76.1MHz

OUTPUT (O) DIVIDER

The 3-bit O divider can reduce the frequency from the VCO

to extend the output frequency range. Its divide ratio O

may be set to any integer from 1 to 6, inclusive, outputting

a 50% duty cycle even with odd divide values. Use the

OD[2:0] bits found in register h0B to directly program the

O divide ratio. See the Applications Information section

for the relationship between O and the fREF, fPFD, fVCO,

and fRF frequencies.

RF OUTPUT BUFFER

The low noise, differential output buffer produces a dif-

ferential output power of –4.3dBm to +4.5dBm, settable

with bits RFO[1:0] according to Table 14. The outputs may

be combined externally, or used individually. Terminate any

unused output with a 50Ω resistor to VRF+.

Table 14. RFO[1:0] Programming

RFO[1:0} PRF (DIFFERENTIAL) PRF (SINGLE-ENDED)

0 –4.3dBm –7.3dBm

1 –1.5dBm –4.5dBm

2 1.6dBm –1.4dBm

3 4.5dBm 1.5dBm

Each output is open-collector with 136Ω pull-up resistors

to VRF+, easing impedance matching at high frequencies.

See Figure 5 for circuit details and the Applications Infor-

mation section for matching guidelines. The buffer may be

muted with either the OMUTE bit, found in register h02,

or by forcing the MUTE input low.

6948 F05

VRF

RF+

136Ω136Ω

RF–

MUTE

OMUTE

RFO[1:0]

9MUTE

Figure 5. Simplified RF Interface Schematic

SERIAL PORT

The SPI-compatible serial port provides control and

monitoring functionality. A configurable status output

STAT gives additional instant monitoring.

Communication Sequence

The serial bus is comprised of CS, SCLK, SDI, and SDO.

Data transfers to the part are accomplished by the se-

rial bus master device first taking CS low to enable the

LTC6948’s port. Input data applied on SDI is clocked on

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Figure 6. Serial Port Write Timing Diagram

Figure 8. Serial Port Write Sequence

Figure 7. Serial Port Read Timing Diagram

MASTER–CS

MASTER–SCLK

tCSS

tCS tCH

DATA DATA

6948 F06

tCKL tCKH

tCSS

tCSH

MASTER–SDI

MASTER–CS

MASTER–SCLK

LTC6948–SDO Hi-Z Hi-Z

6948 F07

8TH CLOCK

DATA DATA

tDO

tDO tDO

the rising edge of SCLK, with all transfers MSB first. The

communication burst is terminated by the serial bus master

returning CS high. See Figure 6 for details.

Data is read from the part during a communication burst

using SDO. Readback may be multidrop (more than one

LTC6948 connected in parallel on the serial bus), as SDO

is three-stated (Hi-Z) when CS = 1, or when data is not

being read from the part. If the LTC6948 is not used in

a multidrop configuration, or if the serial port master is

not capable of setting the SDO line level between read

sequences, it is recommended to attach a high value

resistor of greater than 200k between SDO and GND to

ensure the line returns to a known level during Hi-Z states.

See Figure 7 for details.

Single Byte Transfers

The serial port is arranged as a simple memory map, with

status and control available in 15 byte-wide registers. All

data bursts are comprised of at least two bytes. The seven

most significant bits of the first byte are the register address,

with an LSB of 1 indicating a read from the part, and LSB

of 0 indicating a write to the part. The subsequent byte,

or bytes, is data from/to the specified register address.

See Figure 8 for an example of a detailed write sequence,

and Figure 9 for a read sequence.

Figure 10 shows an example of two write communication

bursts. The first byte of the first burst sent from the serial

bus master on SDI contains the destination register address

A6 A5 A4 A3 A2

7-BIT REGISTER ADDRESS

Hi-Z

MASTER–CS

MASTER–SCLK

MASTER–SDI

LTC6948–SD0

A1 A0 0D7 D6 D5 D4 D3 D2 D1 D0

8 BITS OF DATA

0 = WRITE

6948 F08

16 CLOCKS

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(Addr0) and an LSB of 0 indicating a write. The next byte is

the data intended for the register at address Addr0. CS is

then taken high to terminate the transfer. The first byte of

the second burst contains the destination register address

(Addr1) and an LSB indicating a write. The next byte on

SDI is the data intended for the register at address Addr1.

CS is then taken high to terminate the transfer.

Multiple Byte Transfers

More efficient data transfer of multiple bytes is accom-

plished by using the LTC6948’s register address auto-

increment feature as shown in Figure 11. The serial port

master sends the destination register address in the first

byte and its data in the second byte as before, but continues

sending bytes destined for subsequent registers. Byte 1’s

address is Addr0+1, Byte 2’s address is Addr0+2, and so

on. If the register address pointer attempts to increment

past 14 (h0E), it is automatically reset to 0.

An example of an auto-increment read from the part is

shown in Figure 12. The first byte of the burst sent from

the serial bus master on SDI contains the destination reg-

ister address (Addr0) and an LSB of 1 indicating a read.

Once the LTC6948 detects a read burst, it takes SDO out

of the Hi-Z condition and sends data bytes sequentially,

beginning with data from register Addr0. The part ignores

all other data on SDI until the end of the burst.

Figure 9. Serial Port Read Sequence

Figure 10. Serial Port Single Byte Write

Figure 11. Serial Port Auto-Increment Write

Figure 12. Serial Port Auto-Increment Read

A6 A5 A4 A3 A2

7-BIT REGISTER ADDRESS

Hi-Z Hi-Z

A1 A0 1

D7X D6 D5 D4 D3 D2 D1 D0 DX

8 BITS OF DATA

1 = READ

6948 F09

MASTER–CS

MASTER–SCLK

MASTER–SDI

LTC6948–SDO

16 CLOCKS

ADDR0 + Wr

Hi-Z

MASTER–CS

MASTER–SDI

LTC6948–SDO

BYTE 0 ADDR1 + Wr BYTE 1

6948 F10

ADDR0 + Wr

Hi-Z

MASTER–CS

MASTER–SDI

LTC6948–SDO

BYTE 0 BYTE 1 BYTE 2

6948 F11

ADDR0 + Rd DON’T CARE

Hi-Z Hi-Z

MASTER–CS

MASTER–SDI

LTC6948–SDO

6948 F12

BYTE 0 BYTE 1 BYTE 2

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Table 15. Serial Port Register Contents

ADDR MSB [6] [5] [4] [3] [2] [1] LSB R/W DEFAULT

h00 * * UNLOCK ALCHI ALCLO LOCK THI TLO R

h01 * * x[5] x[4] x[3] x[2] x[1] x[0] R/W h04

h02 PDALL PDPLL PDVCO PDOUT PDFN MTCAL OMUTE POR R/W h06

h03 ALCEN ALCMON ALCCAL ALCULOK AUTOCAL AUTORST DITHEN INTN R/W h3E

h04 BD[3] BD[2] BD[1] BD[0] CPLE LDOEN LDOV[1] LDOV[0] R/W h47

h05 SEED[7] SEED[6] SEED[5] SEED[4] SEED[3] SEED[2] SEED[1] SEED[0] R/W h11

h06 RD[4] RD[3] RD[2] RD[1] RD[0] * ND[9] ND[8] R/W h08

h07 ND[7] ND[6] ND[5] ND[4] ND[3] ND[2] ND[1] ND[0] R/W hFA

h08 * * NUM[17] NUM[16] NUM[15] NUM[14] NUM[13] NUM[12] R/W h3F

h09 NUM[11] NUM[10] NUM[9] NUM[8] NUM[7] NUM[6] NUM[5] NUM[4] R/W hFF

h0A NUM[3] NUM[2] NUM[1] NUM[0] * * RSTFN CAL R/W hF0

h0B BST FILT[1] FILT[0] RFO[1] RFO[0] OD[2] OD[1] OD[0] R/W hF9

h0C LKWIN[2] LKWIN[1] LKWIN[0] LKCT[1] LKCT[0] CP[2] CP[1] CP[0] R/W h4F

h0D CPCHI CPCLO CPMID CPINV CPWIDE CPRST CPUP CPDN R/W hE4

h0E REV[3] REV[2] REV[1] REV[0] PART[3] PART[2] PART[1] PART[0] R hxx†

*unused †varies depending on version

Multidrop Configuration

Several LTC6948s may share the serial bus. In this mul-

tidrop configuration, SCLK, SDI, and SDO are common

between all parts. The serial bus master must use a separate

CS for each LTC6948 and ensure that only one device has

CS asserted at any time. It is recommended to attach a

high value resistor to SDO to ensure the line returns to a

known level during Hi-Z states.

Serial Port Registers

The memory map of the LTC6948 may be found below in

Table 15, with detailed bit descriptions found in Table 16.

The register address shown in hexadecimal format under

the ADDR column is used to specify each register. Each

(R/W). The register’s default value on device power-up or

after a reset is shown at the right.

The read-only register at address h00 is used to determine

different status flags. These flags may be instantly output

on the STAT pin by configuring register h01. See STAT

Output section that follows for more information.

The read-only register at address h0E is a ROM byte for

device identification.

STAT Output

The STAT output pin is configured with the x[5:0] bits

of register h01. These bits are used to bit-wise mask, or

enable, the corresponding status flags of status register

h00, according to Equation 2. The result of this bit-wise

Boolean operation is then output on the STAT pin.

STAT = OR (Reg00[5:0] AND Reg01[5:0]) (2)

or, expanded,

STAT = (UNLOCK AND x[5]) OR

(ALCHI AND x[4]) OR

(ALCLO AND x[3]) OR

(LOCK AND x[2]) OR

(THI AND x[1]) OR

(TLO AND x[0])

For example, if the application requires STAT to go high

whenever the ALCHI, ALCLO, or THI flags are set, then

x[4], x[3], and x[1] should be set to 1, giving a register

value of h1A.

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Block Power-Down Control

The LTC6948’s power-down control bits are located in

the device may be powered down independently. Care must

be taken with the LSB of the register, the POR (power-on-

reset) bit. When written to a 1, this bit forces a full reset

of the part’s digital circuitry to its power-up default state.

operaTion

Table 16. Serial Port Register Bit Field Summary

BITS DESCRIPTION DEFAULT

ALCCAL Auto Enable ALC During CAL Operation 1

ALCEN Always Enable ALC (Override) 1

ALCHI ALC too Hi Flag

ALCLO ALC too Low Flag

ALCMON Enable ALC Monitor for Status Flags Only 0

ALCULOK Enable ALC When PLL Unlocked 1

AUTOCAL Calibrate VCOs Whenever Registers h06 to

h0A Are Written

AUTORST Reset Modulator Whenever Registers h05

to h0A Are Written

BD[3:0] Calibration B Divider Value h4

BST REF Buffer Boost Current 1

CAL Start VCO Calibration (Auto Clears) 0

CP[2:0] CP Output Current h7

CPCHI Enable Hi-Voltage CP Output Clamp 1

CPCLO Enable Low-Voltage CP Output Clamp 1

CPDN Force CP Pump Down 0

CPINV Invert CP Phase 0

CPLE CP Linearizer Enable 0

CPMID CP Bias to Mid-Rail 1

CPRST CP Tri-State 1

CPUP Force CP Pump Up 0

CPWIDE Extend CP Pulse Width 0

DITHEN Enable Fractional Numerator Dither 1

FILT[1:0] REF Input Buffer Filter h3

INTN Integer Mode; Fractional Modulator Placed

in Standby

BITS DESCRIPTION DEFAULT

LDOEN LDO Enable 1

LDOV[1:0] LDO Voltage h3

LKCT[1:0] PLL Lock Cycle Count h1

LKWIN[2:0] PLL Lock Indicator Window h2

LOCK PLL Lock Indicator Flag

MTCAL Mute RF Output During Calibration 1

ND[9:0] N Divider Value (ND[9:0] ≥ 32) h0FA

NUM[17:0] Fractional Numerator Value h3FFF

OD[2:0] Output Divider Value (0 < OD[2:0] < 7) h1

OMUTE Mutes RF Output 1

PART[3:0] Part Code (h01 for -1, h02 for -2, h03 for

-3, h04 for -4 version)

h01, h02,

h03, h04

PDALL Full Chip Powerdown 0

PDFN Powers Down LDO and Modulator Clock 0

PDOUT Powers Down O_DIV, RF Output Buffer 0

PDPLL Powers Down REF, R_DIV, PFD, CPUMP,

N_DIV

PDVCO Powers Down VCO, N_DIV 0

POR Force Power-On-Reset 0

RD[4:0] R Divider Value (RD[4:0] > 0) h001

REV[3:0] Rev Code h1

RFO[1:0] RF Output Power h3

RSTFN Force Modulator Reset (Auto Clears) 0

SEED[7:0] Modulator Dither Seed Value h11

THI CP Clamp High Flag

TLO CP Clamp Low Flag

UNLOK PLL Unlock Flag

x[5:0] STAT Output OR Mask h04

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INTRODUCTION

A PLL is a complex feedback system that may conceptually

be considered a frequency multiplier. The system multiplies

the frequency input at REF± and outputs a higher frequency

at RF±. The PFD, charge pump, N divider, VCO and external

loop filter form a feedback loop to accurately control the

output frequency (see Figure 13).

The external loop filter is used to set the PLL’s loop

bandwidth BW. Lower bandwidths generally have better

spurious performance and lower Δ∑ modulator quantization

noise. Higher bandwidths can have better total integrated

phase noise.

The R and O divider and input frequency fREF are used to

set the output frequency resolution. When in fractional

mode, the Δ∑ modulator changes the N divider’s ratio

each PFD cycle to produce an average fractional divide

ratio. This achieves a much smaller frequency resolution

for a given fPFD as compared to integer mode.

OUTPUT FREQUENCY

When the loop is locked, the frequency fVCO (in Hz)

produced at the output of the VCO is determined by the

reference frequency fREF, the R and N divider values, and

the fractional value F, given by Equation 3:

fVCO =

REF

•N

( )

(3)

where the fractional value F is given by Equation 4:

NUM

(4)

NUM is programmable from 1 to 262143, or 218 – 1. When

using the LTC6948 in integer mode, F = 0.

The PFD frequency fPFD is given by the following equation:

fPFD =

REF

(5)

and fVCO may be alternatively expressed as:

fVCO = fPFD • (N + F) (6)

The output frequency fRF produced at the output of the O

divider is given by Equation 7:

fRF =

VCO

(7)

Using the above equations, the minimum output frequency

resolution fSTEP(MIN) produced by a unit change in the

fractional numerator NUM while in fractional mode is

given by Equation 8:

fSTEP(MIN) =

REF

R•O•2

(8)

Figure 13. PLL Loop Diagram

R_DIV

N_DIV

÷R

O_DIV

÷O

÷(N + F)

fPFD

LTC6948

REF±

(fREF)

fVCO

KPFD

KVCO

RF±

(fRF)15

LOOP FILTER

(FOURTH ORDER)

LF(s)

6948 F13

TUNE

ICP

∆∑

LF(s)

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Alternatively, to calculate the numerator step size NUMSTEP

needed to produce a given frequency step fSTEP(FRAC), use

Equation 9:

NUMSTEP =fSTEP(FRAC) •R•O•218

REF

(9)

The output frequency resolution fSTEP(INT) produced by a

unit change in N while in integer mode is given by Equa-

tion 10:

fSTEP(INT) =

REF

R•O

(10)

LOOP FILTER DESIGN

A stable PLL system requires care in designing the external

loop filter. The Linear Technology FracNWizard application,

available from www.linear.com, aids in design and simula-

tion of the complete system.

The loop design should use the following algorithm:

1) Determine the output frequency fRF and frequency step

size fSTEP based on application requirements. Using

Equations 3, 5, 7, and 8, change fREF, N, R, and O until

the application frequency constraints are met. Use the

minimum R value that still satisfies the constraints.

Then calculate B using Equation 1 and Tables 8 and

2) Select the open loop bandwidth BW constrained by

fPFD and oversampling ratio OSR. The OSR is the ratio

of fPFD to BW (see Equation 11):

OSR =

PFD

W=fPFD

OSR

(11)

where BW and fPFD are in Hz.

A stable loop, both in integer and fractional mode,

requires that the OSR is greater than or equal to 10.

Further, in fractional mode, OSR must be high enough

to allow the loop filter to reduce modulator quantiza-

tion noise to an acceptable level.

Choosing a higher order loop filter when using the Δ∑

modulator allows for a smaller OSR, and thus a larger

loop bandwidth. Linear Technology’s FracNWizard

helps choose the appropriate OSR and BW values.

3) Select loop filter component RZ and charge pump cur-

rent ICP based on BW and the VCO gain factor, KVCO.

BW (in Hz) is approximated by the following equation:

BW ≅

•R

•K

VCO

2•π•N

RZ=2•π•BW •N

•K

VCO

(12)

where KVCO is in Hz/V, ICP is in Amps, and RZ is in

Ohms. KVCO is obtained from the VCO Tuning Sensitivity

in the Electrical Characteristics. Use ICP = 5.6mA to

lower in-band noise unless component values force

a lower setting.

4) Select loop filter components CI and CP based on BW

and RZ. A reliable second-order loop filter design can

be achieved by using the following equations for the

loop capacitors (in Farads).

CI=

3.5

2•π•BW •R

(13)

CP=

7•π•BW •R

(14)

Use FracNWizard to aid in the design of higher order

loop filters.

LTC6948

6948fa

For more information www.linear.com/LTC6948

DESIGN AND PROGRAMMING EXAMPLE

This programming example uses the DC1959 with the

LTC6948-2. Assume the following parameters of interest:

fREF = 100MHz at 7dBm into 50Ω

fSTEP = 50kHz

fRF = 1921.650MHz

From the Electrical Characteristics table:

fVCO = 3.080GHz to 4.910GHz

KVCO% = 4.7%Hz/V to 7%Hz/V

Determining Divider Values

Following the loop filter design algorithm, first determine

all the divider values. The maximum fPFD while in fractional

mode is less than 100MHz, so R must be greater than 1.

R = 2

Then, using Equations 5 and 7, calculate the following

values:

O = 2

fPFD = 50MHz

Then using Equation 6:

N+F=

2•1921.650MHz

50MHz

=76.866

Therefore:

N = 76

F = 0.866

Then, from Equation 4,

NUM = 0.866 • 218 = 227017

Also, from Equation 1 and Tables 8 and 9 determine B:

B = 48 and BD[3:0] = h5

A calibration cycle takes 12 to 14 clock cycles of fCAL. This

gives a VCO calibration time of approximately:

tCAL ≅

CAL

=14•

PFD

=13.4µs

Selecting Filter Type and Loop Bandwidth

The next step in the algorithm is choosing the open loop

bandwidth. Select the minimum bandwidth resulting from

the below constraints.

1) The OSR must be at least 10 (sets absolute maximum

BW).

2) The integrated phase noise due to thermal noise should

be minimized, neglecting any modulator noise.

3) The loop bandwidth must be narrow enough to ad-

equately filter the modulator’s quantization noise.

FracNWizard reports the loop bandwidths resulting from

each of the above constraints. The quantization noise con-

strained results vary according to the shape of the external

loop filter. FracNWizard reports an optimal bandwidth for

several filter types.

FracNWizard reports the thermal noise optimized loop

bandwidth is 211kHz. Filter 3 (fourth order response) has

a quantization noise constrained BW of 150kHz, making

it a good choice. Select Filter 3 and use the smaller of the

two bandwidths (150kHz) for optimal integrated phase

noise. Use Equation 11 to calculate OSR:

OSR =

50MHz

150kHz

=333.3

applicaTions inForMaTion

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applicaTions inForMaTion

Loop Filter Component Selection

Now set loop filter resistor RZ and charge pump current

ICP. Because the KVCO varies over the VCO’s frequency

range, using the KVCO geometric mean gives good results:

KVCO =3.843 •109•0.047 •0.07

=220.4MHz / V

Using an ICP of 5.6mA, the FracNWizard uses Equation12

to determine RZ:

RZ=

2•

•150k •76

5.6m•220.4M

=58.0Ω

For the 4th order Filter 3, FracNWizard uses modified

Equations 13 and 14 to calculate CI, CP:

CI=

4.5

2•π•150k •58 =82.3nF

CP=1

10.5•π•150k •58

=3.5nF

FracNWizard calculates R1, L1, and C2 to be:

R1 = 58.0Ω

C2 = 2.3nF

L1 = 7.8µH

Status Output Programming

This example will use the STAT pin to alert the system

whenever the LTC6948 generates a fault condition. Pro-

gram x[5], x[4], x[3], x[1], x[0] = 1 to force the STAT pin

high whenever any of the UNLOCK, ALCHI, ALCLO, THI,

or TLO flags asserts:

Reg01 = h3B

Power Register Programming

For correct PLL operation all internal blocks should be

enabled. OMUTE may remain asserted (or the MUTE pin

held low) until programming is complete. For OMUTE = 1:

Reg02 = h02

VCO ALC, AUTOCAL, and AUTORST Programming

Set the ALC options (ALCMON = ALCULOK = ALCCAL=1),

the auto reset options (AUTOCAL = AUTORST = 1), and

the Δ∑ modulator modes (DITHEN = 1, INTN = 0) at the

same time:

Reg03 = h7E

The ALC will only be active during a calibration cycle or

when the loop is unlocked, but the ALCHI and ALCLO

status conditions will be monitored continuously. The VCO

will be calibrated and the Δ∑ modulator will be reset at the

end of the SPI write communication burst (assuming an

auto-increment write is used to write all registers).

LDO Programming

Use Table 13 and fPFD = 50MHz to determine V(LDO) and

LDOV[1:0]:

V(LDO) = 2.3V and LDOV[1:0] = 2

Use LDOV[1:0], LDOEN = 1 to enable the LDO, and the

previously determined BD[3:0] value to set Reg04. CPLE

should be set to 1 to reduce in-band noise and spurious

due to the Δ∑ modulator:

Reg04 = h5E

SEED Programming

The SEED[7:0] value is used to initialize the Δ∑ modulator

dither circuitry. Use the default value:

Reg05 = h11

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applicaTions inForMaTion

R and N Divider and Numerator Programming

Program registers Reg06 to Reg0A with the previously

determined R and N divider and numerator values. Because

the AUTORST and AUTOCAL bits were previously set to

1, CAL and RSTFN do not need to be set:

Reg06 = h10

Reg07 = h4C

Reg08 = h37

Reg09 = h6C

Reg0A = h90

Reference Input Settings and Output Divider

Programming

From Table 1, FILT = 0 for a 100MHz reference frequency.

Next, convert 7dBm into VP–P. For a CW tone, use the

following equation with R = 50:

P-P ≅R•10

(dBm –21)/20

(15)

This gives VP-P = 1.41V, and, according to Table 2, set

BST = 1.

Now program Reg0B, assuming maximum RF± output

power (RFO[1:0] = 3 according to Table 14) and OD[2:0] = 2:

Reg0B = h9A

Lock Detect and Charge Pump Current Programming

Next, determine the lock indicator window from fPFD. From

Table 3 we see that LKWIN[1:0] = 0 with a tLWW of 5ns for

CPLE = 1 and fVCO = 3.843GHz. The LTC6948 will consider

the loop locked as long as the phase coincidence at the

PFD is within 90°, as calculated below.

phase = 360° • tLWW • fPFD = 360 • 5n • 50M

= 90°

Choosing the correct COUNTS value depends upon the

OSR. Smaller ratios dictate larger COUNTS values, although

application requirements will vary. A COUNTS value of 32

will work for the OSR of 333. From Table 5, LKCT[1:0]=1

for 32 counts.

Using Table 6 with the previously selected ICP of 5.6mA

gives CP[3:0] = 7. This gives enough information to pro-

gram Reg0C:

Reg0C = h0D

Charge Pump Function Programming

This example uses the additional voltage clamp features

to allow the monitoring of fault conditions by setting

CPCHI=1 and CPCLO = 1. If something occurs and the

system can no longer lock to its intended frequency, the

charge pump output will move toward either GND or

VCP+, thereby setting either the TLO or THI status flags,

respectively. Disable all the other charge pump functions

(CPMID, CPINV, CPRST, CPUP, and CPDN), allowing the

loop to lock:

Reg0D = hC0

The loop should now lock. Now un-mute the output by

setting OMUTE = 0 (assumes the MUTE pin is high).

Reg02 = h04

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applicaTions inForMaTion

REFERENCE SOURCE CONSIDERATIONS

A high quality signal must be applied to the REF± inputs as

they provide the frequency reference to the entire PLL. As

mentioned previously, to achieve the part’s in-band phase

noise performance, apply a CW signal of at least 6dBm

into 50Ω, or a square wave of at least 0.5VP-P with slew

rate of at least 40V/µs.

The LTC6948 may be driven single-ended to CMOS levels

(greater than 2.7VP-P ). Apply the reference signal at REF+,

and bypass REF– to GND with a 47pF capacitor. The BST

bit must also be set to 0, according to guidelines given

in Table 2.

The LTC6948 achieves an integer mode in-band normalized

phase noise floor LNORM(INT) = –226dBc/Hz typical, and

a fractional mode phase noise floor LNORM(FRAC) = –225

dBc/Hz typical. To calculate its equivalent input phase

noise floor LIN, use the following Equation 16.

LIN = LNORM + 10 • log10 (fREF) (16)

For example, using a 10MHz reference frequency in integer

mode gives an input phase noise floor of –156dBc/Hz.

The reference frequency source’s phase noise must be at

least 3dB better than this to prevent limiting the overall

system performance.

IN-BAND OUTPUT PHASE NOISE

The in-band phase noise floor LOUT produced at fRF may

be calculated by using Equation 17.

LOUT = LNORM + 10 • log10 (fPFD) (17)

+ 20 • log10 (fRF/fPFD)

LOUT ≈ LNORM + 10 • log10 (fPFD)

+ 20 • log10 (N/O)

where LNORM is –226dBc/Hz for integer mode and

–225dBc/Hz for fractional mode. See the Typical Perfor-

mance Characteristics section for graphs showing LNORM

variation versus ICP and fVCO.

As can be seen from Equation 17 for a given PFD frequency

fPFD, the output in-band phase noise increases at a 20dB-

per-decade rate with the N divider count. So, for a given

output frequency fRF, fPFD should be as large as possible

(or N should be as small as possible) while still satisfying

the application’s frequency step size requirements.

OUTPUT PHASE NOISE DUE TO 1/f NOISE

In-band phase noise at very low offset frequencies may

be influenced by the LTC6948’s 1/f noise, depending upon

fPFD. Use the normalized in-band 1/f noise L1/f of –274dBc/

Hz with Equation 18 to approximate the output 1/f phase

noise at a given frequency offset fOFFSET.

LOUT(1/f) (fOFFSET) = L1/f + 20 • log10 (fRF) (18)

– 10 • log10 (fOFFSET)

Unlike the in-band noise floor LOUT, the 1/f noise LOUT(1/f)

does not change with fPFD, and is not constant over offset

frequency. See Figure 14 for an example of integer mode

in-band phase noise for fPFD equal to 3MHz and 100MHz.

The total phase noise will be the summation of LOUT and

LOUT(1/f).

Figure 14. Theoretical Integer Mode In-Band

Phase Noise, fRF = 2500MHz

OFFSET FREQUENCY (Hz)

PHASE NOISE (dBc/Hz)

–110

–100

100k

6948 F14

–120

–130 100 1k 10k

–90

TOTAL NOISE

fPFD = 3MHz

TOTAL NOISE

fPFD = 100MHz

1/f NOISE

CONTRIBUTION

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IB SPUR LEVEL (dBc)

–40

–50

–60

–70

–80

–90

–100

–110

6948 F15

FREQUENCY OFFSET FROM 3GHz (Hz)

1k 10k

10M

1M100k

fVCO ≈ 3GHz

fPFD = 50MHz

LOOP BW = 130kHz

CPLE = 1

N = 60

SWEPT NUM

Figure 15. Integer Boundary Spur Power vs Frequency

Offset from Boundary, LTC6948-1

INTEGER BOUNDARY SPURS

Integer boundary spurs are caused by intermodulation

between harmonics of the PFD frequency fPFD and the

VCO frequency fVCO. The coupling between the frequency

source harmonics can occur either on- or off-chip. The

spurs are located at offset frequencies defined by the beat

frequency between the reference harmonics and the VCO

frequency, and are attenuated by the loop filter. The spurs

only occur while in fractional mode.

Integer boundary spurs are most commonly seen when

the fractional value F approaches either zero or one such

that the VCO frequency offset from an integer frequency

is within the loop bandwidth:

fPFD • F ≤ BW

fPFD • (1 – F) ≤ BW

The spur will have a relatively constant power in-band,

and be attenuated by the loop out-of-band. An example

integer boundary spur measurement is shown in Figure 15.

RF OUTPUT MATCHING

The RF± outputs may be used in either single-ended or dif-

ferential configurations. Using both RF outputs differentially

will result in approximately 3dB more output power than

single-ended. Impedance matching to an external load in

both cases requires external chokes tied to VRF+. Measured

RF± S-parameters are shown below in Table17 to aid in

the design of impedance matching networks.

Table 17. Single-Ended RF Output Impedance

FREQUENCY (MHz) IMPEDANCE (Ω) S11 (dB)

100 133.0 – j16.8 –6.7

500 110.8 – j46.1 –6.8

1000 74.9 – j57.0 –6.9

1500 49.0 – j51.3 –6.7

2000 34.4 – j41.4 –6.5

2500 27.0 – j32.1 –6.5

3000 23.2 – j24.1 –6.6

3500 21.6 – j15.9 –7.1

4000 20.9 – j7.7 –7.5

4500 20.1 – j0.2 –7.4

5000 18.1 + j7.4 –6.4

5500 16.7 + j12.5 –5.6

6000 17.1 + j16.1 –5.5

6500 20.2 + j20.1 –6.2

7000 26.9 + j24.6 –7.6

7500 38.8 + j32.3 –8.8

8000 52.9 + j43.1 –8.2

applicaTions inForMaTion

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applicaTions inForMaTion

Single-ended impedance matching is accomplished using

the circuit of Figure 16, with component values found in

Table 18. Using smaller inductances than recommended

can cause phase noise degradation, especially at lower

center frequencies.

Table 18. Suggested Single-Ended Matching Component Values

fRF (MHz) LC (nH) CS (pF)

350 to 1500 180 270

1000 to 5800 68 100

Return loss measured on the DC1959 using the above

component values is shown in Figure 17. A broadband

match is achieved using an {LC, CS} of either {68nH, 100pF}

or {180nH, 270pF}. However, for maximum output power

and best phase noise performance, use the recommended

component values of Table 18. LC should be a wirewound

inductor selected for maximum Q factor and SRF, such as

the Coilcraft HP series of chip inductors.

The LTC6948’s differential RF± outputs may be combined

using an external balun to drive a single-ended load. The

advantages are approximately 3dB more output power than

each output individually and better 2nd order harmonic

performance.

For lower frequencies, transmission line (TL) baluns such as

the M/A-COM MABACT0065 and the TOKO #617DB-1673

provide good results. At higher frequencies, surface mount

(SMT) baluns such as those produced by TDK, Anaren,

and Johanson Technology, can be attractive alternatives.

See Table 19 for recommended balun part numbers versus

frequency range.

The listed SMT baluns contain internal chokes to bias RF±

and also provide input-to-output DC isolation. The pin

denoted as GND or DC FEED should be connected to the

VRF+ voltage. Figure 18 shows a surface mount balun’s

connections with a DC FEED pin.

Table 19. Suggested Baluns

fRF (MHz) PART NUMBER MANUFACTURER TYPE

350 to 900 #617DB-1673 TOKO TL

400 to 600 HHM1589B1 TDK SMT

600 to 1400 BD0810J50200 Anaren SMT

600 to 3000 MABACT0065 M/A-COM TL

1000 to 2000 HHM1518A3 TDK SMT

1400 to 2000 HHM1541E1 TDK SMT

1900 to 2300 2450BL15B100E Johanson SMT

2000 to 2700 HHM1526 TDK SMT

3700 to 5100 HHM1583B1 TDK SMT

4000 to 6000 HHM1570B1 TDK SMT

The listed TL baluns do not provide input-to-output DC

isolation and must be AC-coupled at the output. Figure 19

displays RF± connections using these baluns.

Figure 17. RF Single-Ended Return Loss

0123456

FREQUENCY (GHz)

S11 (dB)

–6

–4

–2

6947 F17

–10

–16

–8

–12

–14

68nH, 100pF

180nH, 270pF

Figure 16. Single-Ended Output Matching Schematic

RF+(–)

50Ω

TO 50Ω

LOAD

VRF

RF–(+)

6948 F16

VRF+

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Figure 20. Example Exposed Pad Land Pattern

6948 F20

applicaTions inForMaTion

Figure 18. Example SMT Balun Connection

Figure 19. Example TL Balun Connection

LTC6948

VRF

RF–

RF+

TO 50Ω

LOAD

6948 F18

BALUN

23 1

54 6

BALUN PIN CONFIGURATION

UNBALANCED PORT

GND OR DC FEED

BALANCED PORT

GND

LTC6948

VRF

RF–

RF+

TO 50Ω

LOAD

PRI

SEC

6948 F19

SUPPLY BYPASSING AND PCB LAYOUT GUIDELINES

Care must be taken when creating a PCB layout to mini-

mize power supply decoupling and ground inductances.

All power supply V+ pins should be bypassed directly to

the ground plane using a 0.1µF ceramic capacitor as close

to the pin as possible. Multiple vias to the ground plane

should be used for all ground connections, including to

the power supply decoupling capacitors.

The package’s exposed pad is a ground connection, and

must be soldered directly to the PCB land pattern. The

PCB land pattern should have multiple thermal vias to the

ground plane for both low ground inductance and also low

thermal resistance (see Figure 20 for an example). See

QFN Package Users Guide, page 8, on Linear Technology

website’s Packaging Information page for specific recom-

mendations concerning land patterns and land via solder

masks. A link is provided below.

http://www.linear.com/designtools/packaging

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REFERENCE SIGNAL ROUTING, SPURIOUS, AND

PHASE NOISE

The charge pump operates at the PFD’s comparison

frequency fPFD. The resultant output spurious energy is

small and is further reduced by the loop filter before it

modulates the VCO frequency.

However, improper PCB layout can degrade the LTC6948’s

inherent spurious performance. Care must be taken to

prevent the reference signal fREF from coupling onto the

VCO’s tune line, or into other loop filter signals. Example

suggestions are the following.

applicaTions inForMaTion

1) Do not share power supply decoupling capacitors

between same-voltage power supply pins.

2) Use separate ground vias for each power supply

decoupling capacitor, especially those connected to

VREF+, VD+, LDO, VCP+, and VVCO+.

3) Physically separate the reference frequency signal

from the loop filter and VCO.

4) Do not place a trace between the CMA, CMB, and CMC

pads underneath the package, as worse phase noise

could result.

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OUTPUT POWER (dBm)

–20

NOISE FLOOR (dBm/Hz)

–148

–150

–152

–156

–160

–154

–158

–162 –15 0

6948 TA02c

–5–10

fLO = 2100MHz

BASEBAND = 2kHz SINE

OUTPUT FREQUENCY (MHz)

1500

(dBc)

–20

–30

–40

–50

–60

–70

–80 1700 1900 23002100

6948 TA02b

2500

BASEBAND = 100kHz AT 500mVPK

LO LEAKAGE RATIO

UNADJUSTED

IMAGE REJECTION

Typical applicaTions

Driving a Modulator LO for High Image Rejection and Low Noise Floor

Measured Image Rejection and LO

Leakage Ratio vs Output Frequency

Measured Noise Floor at 70MHz

Offset vs RF Output Power

UNUSED OUTPUT

AVAILABLE FOR

OTHER USE

50Ω

0.1µF

R = 1, fPFD = 61.44MHz

N = 73.24 TO 84.64

LBW = 180kHz

O = 2

1µF

0.01µF

3.3V

0.1µF

68nH

0.01µF

3.3V

66.5Ω

100pF

1.8nF

3.3V

LTC6948-3

STAT

TUNE

MUTE

SDI

VD+

LDO

RF+

RF–

GND

SCLK

GND

VRF+

REF+

SDO

CMC

CMB

CMA

GND

BVCO

VVCO+

GND

VCP+

VREF+

REF–

GND

0.1µF

61.44MHz

51.1Ω

1µF

3.3V

1µF

0.1µF

3.3V

1µF

2.2µF

15Ω

SPI BUS

10µH

66.5Ω

68nF

2.2nF

AVAGO

VMMK-2503

4.7µF

MINI-CIRCUITS

LFCN-2600+

LPF

50Ω

1Ω

1nF

6.8pF

1nF

5V 3.3V

LTC5588-1

BASEBAND

I-CHANNEL

BASEBAND

Q-CHANNEL

GND

LOP

LOM

GND

LINOPT

GND

BBMQ

BBPQ

GND

GNDRF

GND

GNDRF

VCC1

GND

BBMI

BBPI

GND

GNDRF

VCC2

GNDRF

100nF

6948 TA02a

fLO = 1500MHz TO 2600MHz IN 117.2Hz STEPS

RF OUTPUT,

1500MHz TO 2600MHz CARRIER

PLO = 13.5dBm, ~ <–40dBc ODD HARMONIC CONTENT

~ <–30dBc EVEN HARMONIC CONTENT

0.1µF

1nF

4.7µF

1.3Ω

3.3V

1nF

0.2pF

100pF

10nH

0.01µF

1µF

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package DescripTion

Please refer to http://www.linear.com/product/LTC6948#packaging for the most recent package drawings.

UFD Package

28-Lead Plastic QFN (4mm × 5mm)

(Reference LTC DWG # 05-08-1712 Rev C)

4.00 ±0.10

(2 SIDES)

2.50 REF

5.00 ±0.10

(2 SIDES)

NOTE:

1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGHD-3).

2. DRAWING NOT TO SCALE

3. ALL DIMENSIONS ARE IN MILLIMETERS

4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE

MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE

5. EXPOSED PAD SHALL BE SOLDER PLATED

6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION

ON THE TOP AND BOTTOM OF PACKAGE

PIN 1

TOP MARK

(NOTE 6)

0.40 ±0.10

27 28

BOTTOM VIEW—EXPOSED PAD

3.50 REF

0.75 ±0.05 R = 0.115

TYP

R = 0.05

TYP

PIN 1 NOTCH

R = 0.20 OR 0.35

× 45° CHAMFER

0.25 ±0.05

0.50 BSC

0.200 REF

0.00 – 0.05

(UFD28) QFN 0816 REV C

RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS

APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED

0.70 ±0.05

0.25 ±0.05

0.50 BSC

2.50 REF

3.50 REF

4.10 ±0.05

5.50 ±0.05

2.65 ±0.05

3.10 ±0.05

4.50 ±0.05

PACKAGE OUTLINE

2.65 ±0.10

3.65 ±0.10

3.65 ±0.05

UFD Package

28-Lead Plastic QFN (4mm × 5mm)

(Reference LTC DWG # 05-08-1712 Rev C)

LTC6948

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For more information www.linear.com/LTC6948

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

revision hisTory

REV DATE DESCRIPTION PAGE NUMBER

A 04/17 Corrected defaults for ALCULOK and BD[3:0]. 24

LTC6948

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 LINEAR TECHNOLOGY CORPORATION 2014

LT 0417 REV A • PRINTED IN USA

www.linear.com/LTC6948

PART NUMBER DESCRIPTION COMMENTS

LTC6946-x Ultralow Noise and Spurious Integer-N

Synthesizer with Integrated VCO

370MHz to 6.4GHz, –226dBc/Hz Normalized In-Band Phase Noise Floor, –157dBc/Hz

Wideband Output Phase Noise Floor

LTC6945 Ultralow Noise and Spurious Integer-N

Synthesizer

350MHz to 6GHz, –226dBc/Hz Normalized In-Band Phase Noise Floor, –157dBc/Hz

Wideband Output Phase Noise Floor

LTC6947 Ultralow Noise and Spurious Fractional-N

Synthesizer

350MHz to 6GHz, –226dBc/Hz Normalized In-Band Phase Noise Floor, –157dBc/Hz

Wideband Output Phase Noise Floor

LTC6957 Low Phase Noise, Dual Output Buffer/Driver/

Logic Converter

Optimized Conversion of Sine Waves to Logic Levels, LVPECL/LVDS/CMOS Outputs,

DC-300MHz, 45fsRMS additive jitter (LVPECL)

LTC5540/LTC5541/

LTC5542/LTC5543

High Dynamic Range Downconverting Mixers 8dB Conversion Gain, 26.4dBm IIP3, 9.6dB NF, 600MHz to 4GHz

LTC5590/LTC5591/

LTC5592/LTC5593

High Linearity Dual Mixers 600MHz to 4.5GHz, 8.5dB Gain, 26.2dBm IIP3, 9.9dB NF

LTC5569 Broadband Dual Mixer 300MHz to 4GHz, 26.8dBm IIP3, 2dB Gain, 11.7dB NF, 600mW Power

LTC5588-1 Ultrahigh OIP3 I/Q Modulator 200MHz to 6GHz, 31dBm OIP3, –160.6dBm/Hz Noise Floor

5575 Direct Conversion I/Q Demodulator 800MHz to 2.7GHz, 22.6dBm IIP3, 60dBm IIP2, 12.7dB NF

relaTeD parTs

Typical applicaTions

UNUSED OUTPUT

AVAILABLE FOR

OTHER USE

50Ω

0.1µF

R = 5, fPFD = 63.1MHz

N = 66.6 TO 95.1

LBW = 218kHz

O = 1

1µF

0.01µF

3.3V

0.1µF

68nH

0.01µF

3.3V

71.5Ω

100pF

1.5nF

3.3V

fLO

= 8400MHz TO 12000MHz IN 481.4Hz STEPS

PLO = 14dBm TO 16dBm

LTC6948-4

STAT

TUNE

MUTE

SDI

VD+

LDO

RF+

RF–

GND

SCLK

GND

VRF+

REF+

SDO

CMC

CMB

CMA

GND

BVCO

VVCO+

GND

VCP+

VREF+

REF–

GND

0.1µF

315.5MHz, 15dBm

315.5MHz DATA-

CONVERTER CLOCK

CRYSTEK

CCSO-914X

51.1Ω

1µF

3.3V

1µF

0.1µF

3.3V

1µF

2.2µF

16.5Ω16.5Ω

15Ω

16.5Ω

6948 TA03a

SPI BUS

10µH

71.5Ω

47nF

4.7nF

2.2nF

MA-COM

XX1002-QH

1µF1nF

0.01µF

1µF

Generating a 12GHz LO Signal

System Phase Noise

fRF = 12.024GHz

OFFSET FREQUENCY (Hz)

–130

–140

PHASE NOISE (dBc/Hz)

–110

–80

–90

–120

–100

1k 100k 1M 10M

100M

6948 TA03b

–150

10k

RMS NOISE = 0.825°

RMS JITTER = 191fs

fPFD = 63.1MHz

LOOP BW = 218kHz

INTN = 0