10-Bit, 65/80/105 MSPS,
3 V A/D Converter
Data Sheet
AD9215
Rev. B Document Feedback
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FEATURES
Single 3 V supply operation (2.7 V to 3.3 V)
SNR = 58 dBc (to Nyquist)
SFDR = 77 dBc (to Nyquist)
Low power ADC core: 96 mW at 65 MSPS, 104 mW
@ 80 MSPS, 120 mW at 105 MSPS
Differential input with 300 MHz bandwidth
On-chip reference and sample-and-hold amplifier
DNL = ±0.25 LSB
Flexible analog input: 1 V p-p to 2 V p-p range
Offset binary or twos complement data format
Clock duty cycle stabilizer
APPLICATIONS
Ultrasound equipment
IF sampling in communications receivers
Battery-powered instruments
Hand-held scopemeters
Low cost digital oscilloscopes
FUNCTIONAL BLOCK DIAGRAM
SHA
VIN+
VIN–
REFT
REFB
DRVDD
CLK PDWN MODE
CLOCK
DUTY CYCLE
STABLIZER MODE
SELECT
DGND
OR
D9 (MSB)
02874-A-001
D0
AVDD
CORRECTION LOGIC
OUTPUT BUFFERS
10
REF
SELECT
AGND
0.5V
VREF
SENSE
AD9215
PIPELINE
ADC CORE
Figure 1.
PRODUCT DESCRIPTION
The AD9215 is a family of monolithic, single 3 V supply, 10-bit,
65/80/105 MSPS analog-to-digital converters (ADC). This family
features a high performance sample-and-hold amplifier (SHA)
and voltage reference. The AD9215 uses a multistage differential
pipelined architecture with output error correction logic to pro-
vide 10-bit accuracy at 105 MSPS data rates and to guarantee no
missing codes over the full operating temperature range.
The wide bandwidth, truly differential sample-and-hold ampli-
fier (SHA) allows for a variety of user-selectable input ranges
and offsets including single-ended applications. It is suitable for
multiplexed systems that switch full-scale voltage levels in
successive channels and for sampling single-channel inputs at
frequencies well beyond the Nyquist rate. Combined with pow-
er and cost savings over previously available ADCs, the AD9215
is suitable for applications in communications, imaging, and
medical ultrasound.
A single-ended clock input is used to control all internal conversion
cycles. A duty cycle stabilizer compensates for wide variations in the
clock duty cycle while maintaining excellent performance. The digital
output data is presented in straight binary or twos complement for-
mats. An out-of-range signal indicates an overflow condition, which
can be used with the MSB to determine low or high overflow.
Fabricated on an advanced CMOS process, the AD9215 is avail-
able in both a 28-lead surface-mount plastic package and a
32-lead chip scale package and is specified over the industrial
temperature range of −40°C to +85°C.
PRODUCT HIGHLIGHTS
1. The AD9215 operates from a single 3 V power supply and
features a separate digital output driver supply to accom-
modate 2.5 V and 3.3 V logic families.
2. Operating at 105 MSPS, the AD9215 core ADC consumes
a low 120 mW; at 80 MSPS, the power dissipation is 104
mW; and at 65 MSPS, the power dissipation is 96 mW.
3. The patented SHA input maintains excellent performance
for input frequencies up to 200 MHz and can be config-
ured for single-ended or differential operation.
4. The AD9215 is part of several pin compatible 10-, 12-, and
14-bit low power ADCs. This allows a simplified upgrade
from 10 bits to 12 bits for systems up to 80 MSPS.
5. The clock duty cycle stabilizer maintains converter per-
formance over a wide range of clock pulse widths.
6. The out of range (OR) output bit indicates when the signal
is beyond the selected input range.
AD9215 Data Sheet
Rev. B | Page 2 of 36
TABLE OF CONTENTS
Specifications ..................................................................................... 3
Absolute Maximum Ratings1 .......................................................... 6
Explanation of Test Levels ........................................................... 6
ESD Caution .................................................................................. 6
Pin Configurations and Function Descriptions ........................... 7
Equivalent Circuits ....................................................................... 8
Definitions of Specifications ....................................................... 8
Typical Performance Characteristics ........................................... 10
Applying the AD9215 Theory of Operation ............................... 14
Clock Input and Considerations .............................................. 15
Evaluation Board ........................................................................ 18
Outline Dimensions ....................................................................... 33
Ordering Guide ........................................................................... 34
REVISION HISTORY
2/13Data Sheet Changed from a REV. A to a REV. B
Changes to Figure 4 and Added EPAD Note to Pin Configura-
tions and Function Descriptions Section ..................................... 7
Changes to Voltage Reference Section ........................................ 17
Changes to Evaluation Board Section......................................... 18
Updated Outline Dimensions ...................................................... 33
Changes to Ordering Guide ......................................................... 34
2/04Data Sheet Changed from a REV. 0 to a REV. A
Renumbered Figures and Tables ..............................UNIVERSAL
Changes to Product Title ................................................................ 1
Changes to Features ........................................................................ 1
Changes to Product Description ................................................... 1
Changes to Product Highlights ..................................................... 1
Changes to Specifications ............................................................... 2
Changes to Figure 2 ......................................................................... 4
Changes to Figures 9 to 11 ........................................................... 10
Added Figure 14 ............................................................................ 10
Added Figures 16 and 18 .............................................................. 11
Changes to Figures 21 to 24 and 25 to 26 ................................... 12
Deleted Figure 25 ........................................................................... 12
Changes to Figures 28 and 29 ...................................................... 13
Changes to Figure 31 ..................................................................... 14
Changes t0 Figure 35 ..................................................................... 16
Changes to Figures 50 through 58............................................... 26
Added Table 11 .............................................................................. 31
Updated Outline Dimensions ...................................................... 32
Changes to Ordering Guide ......................................................... 33
5/03Revision 0: Initial Version
Data Sheet AD9215
Rev. B | Page 3 of 36
SPECIFICATIONS
AVDD = 3 V, DRVDD = 2.5 V, specified maximum conversion rate, 2 V p-p differential input, 1.0 V internal reference, unless otherwise
noted.
Table 1. DC Specifications
AD9215BRU-65/
AD9215BCP-65
AD9215BRU-80/
AD9215BCP-80
AD9215BRU-105/
AD9215BCP-105
Parameter
Temp
Test
Level
Min
Typ
Max
Min
Typ
Max
Min
Typ
Max
Unit
RESOLUTION Full VI 10 10 10 Bits
ACCURACY
No Missing Codes Full VI Guaranteed Guaranteed Guaranteed
Offset Error1 Full VI ±0.3 ±2.0 ±0.3 ±2.0 ±0.3 ±2.0 % FSR
Gain Error1 Full VI 0 +1.5 +4.0 +1.5 +4.0 +1.5 +4.0 % FSR
Differential Nonlinearity (DNL)2 Full VI 1.0 ±0.5 +1.0 1.0 ±0.5 +1.0 1.0 ±0.6 +1.2 LSB
Integral Nonlinearity (INL)2 Full VI ±0.5 ±1.2 ±0.5 ±1.2 ±0.65 ±1.2 LSB
TEMPERATURE DRIFT
Offset Error1 Full V +15 +15 +15 ppm/°C
Gain Error1 Full V +30 +30 +30 ppm/°C
Reference Voltage (1 V Mode) Full V ±230 ±230 ±230 ppm/°C
INTERNAL VOLTAGE REFERENCE
Output Voltage Error (1 V Mode) Full VI ±2 ±35 ±2 ±35 ±2 ±35 mV
Load Regulation @ 1.0 mA Full V 0.2 0.2 0.2 mV
Output Voltage Error (0.5 V Mode) Full V ± 1 ±1 ±1 mV
Load Regulation @ 0.5 mA Full V 0.2 0.2 0.2 mV
INPUT REFERRED NOISE
VREF = 0.5 V 25°C V 0.8 0.8 0.8 LSB rms
VREF = 1.0 V 25°C V 0.4 0.4 0.4 LSB rms
ANALOG INPUT
Input Span, VREF = 0.5 V Full IV 1 1 1 V p-p
Input Span, VREF = 1.0 V Full IV 2 2 2 V p-p
Input Capacitance3 Full V 2 2 2 pF
REFERENCE INPUT RESISTANCE Full V 7 7 7 kΩ
POWER SUPPLIES
Supply Voltage
AVDD Full IV 2.7 3.0 3.3 2.7 3.0 3.3 2.7 3.0 3.3 V
DRVDD Full IV 2.25 2.5 3.6 2.25 2.5 3.6 2.25 2.5 3.6 V
Supply Current
IAVDD2 Full VI 32 35 34.5 39 40 44 mA
IDRVDD2 25°C V 7.0 8.6 11.3 mA
PSRR Full V ± 0.1 ± 0.1 ± 0.1 % FSR
POWER CONSUMPTION
Sine Wave Input2
IAVDD2 Full VI 96 104 120 mW
IDRVDD2 25°C V 18 20 25 mW
Standby Power4 25°C V 1.0 1.0 1.0 mW
1 With a 1.0 V internal reference.
2 Measured at fIN = 2.4 MHz, full-scale sine wave, with approximately 5 pF loading on each output bit.
3 Input capacitance refers to the effective capacitance between one differential input pin and AGND. Refer to Figure 5 for the equivalent analog input structure.
4 Standby power is measured with a dc input, the CLK pin inactive (i.e., set to AVDD or AGND).
AD9215 Data Sheet
Rev. B | Page 4 of 36
AVDD = 3 V, DRVDD = 2.5 V, specified maximum conversion rate, 2 V p-p differential input, 1.0 V internal reference,
AIN = −0.5 dBFS, MODE = AVDD/3 (duty cycle stabilizer [DCS] enabled), unless otherwise noted.
Table 2. AC Specifications
AD9215BRU-65/
AD9215BCP-65
AD9215BRU-80/
AD9215BCP-80
AD9215BRU-105/
AD9215BCP-105
Parameter
Temp
Test
Level
Min
Typ
Max
Min
Typ
Max
Min
Typ
Max
Unit
SIGNAL-TO-NOISE RATIO (SNR)
fIN = 2.4 MHz Full VI 56.0 58.5 56.0 58.5 57.5 dB
25°C I 57.0 59.0 57.0 59.0 56.6 58.5 dB
fIN = Nyquist1 Full VI 56.0 58.0 56.0 58.0 57.5 dB
25°C I 56.5 58.5 56.5 58.5 56.4 58.0 dB
fIN = 70 MHz 25°C V 58.0 57.8 dB
fIN = 100 MHz 25°C V 57.5 57.7 dB
SIGNAL-TO-NOISE AND DISTORTION (SINAD)
fIN = 2.4 MHz Full VI 55.8 58.5 55.7 58.5 57.6 dB
25°C I 56.5 59.0 56.8 58.5 56.5 58.2 dB
fIN = Nyquist1 Full VI 55.8 58.0 55.5 58.0 57.3 dB
25°C I 56.3 58.5 56.3 58.5 56.1 57.8 dB
fIN = 70 MHz 25°C V 56.0 57.7 dB
fIN = 100 MHz 25°C V 55.5 57.4 dB
EFFECTIVE NUMBER OF BITS (ENOB)
fIN = 2.4 MHz Full VI 9.1 9.5 9.0 9.5 9.3 Bits
25°C I 9.2 9.6 9.3 9.5 9.2 9.5 Bits
fIN = Nyquist1 Full VI 9.1 9.4 9.0 9.4 9.4 Bits
25°C I 9.1 9.5 9.0 9.5 9.1 9.4 Bits
fIN = 70 MHz 25°C V 9.1 9.4 Bits
fIN = 100 MHz 25°C V 9.0 9.3 Bits
WORST HARMONIC (Second or Third)
fIN = 2.4 MHz Full VI −78 −64 −78 −64 −78 dBc
25°C I −80 −65 −80 −65 −84 −70 dBc
fIN = Nyquist1 Full VI −77 −64 −76 −63 −74 dBc
25°C I −78 −65 −78 −65 −75 −61 dBc
fIN = 70 MHz 25°C V −70 −75 dBc
fIN = 100 MHz 25°C V −70 −74 dBc
WORST OTHER (Excluding Second or Third)
fIN = 2.4 MHz Full VI −77 −67 −77 −66 −73 dBc
25°C I −78 −68 −77 −68 −75 −66 dBc
fIN = Nyquist1 Full VI −77 −67 −77 −66 −71 dBc
25°C I −78 −68 −77 −68 −75 −63 dBc
fIN = 70 MHz 25°C V −80 -75 dBc
fIN = 100 MHz 25°C V −80 −75 dBc
TWO-TONE SFDR (AIN = –7 dBFS)
fIN1 = 70.3 MHz, fIN2 = 71.3 MHz 25°C V 75 75 dBc
fIN1 = 100.3 MHz, fIN2 = 101.3 MHz 25°C V 74 74 dBc
ANALOG BANDWIDTH 25°C V 300 300 300 MHz
1 Tested at fIN = 35 MHz for AD9215-65; fIN = 39 MHz for AD9215-80; and fIN = 50 MHz for AD9215-105.
Data Sheet AD9215
Rev. B | Page 5 of 36
Table 3. Digital Specifications
AD9215BRU-65/
AD9215BCP-65
AD9215BRU-80/
AD9215BCP-80
AD9215BRU-105/
AD9215BCP-105
Parameter
Temp
Test
Level
Min
Typ
Max
Min
Typ
Max
Min
Typ
Max
Unit
LOGIC INPUTS (CLK, PDWN)
High Level Input Voltage Full IV 2.0 2.0 2.0 V
Low Level Input Voltage Full IV 0.8 0.8 0.8 V
High Level Input Current
Full
IV
−650
+10
−650
+10
−650
+10
µA
Low Level Input Current Full IV −70 +10 70 +10 −70 +10 µA
Input Capacitance Full V 2 2 2 pF
LOGIC OUTPUTS1
DRVDD = 2.5 V
High Level Output Voltage Full IV 2.45 2.45 2.45 V
Low Level Output Voltage Full IV 0.05 0.05 0.05 V
1 Output voltage levels measured with a 5 pF load on each output.
Table 4. Switching Specifications
AD9215BRU-65/
AD9215BCP-65
AD9215BRU-80/
AD9215BCP-80
AD9215BRU-105/
AD9215BCP-105
Parameter
Temp
Test
Level
Min
Typ
Max
Min
Typ
Max
Min
Typ
Max
Unit
CLOCK INPUT PARAMETERS
Maximum Conversion Rate Full VI 65 80 105 MSPS
Minimum Conversion Rate Full V 5 5 5 MSPS
CLOCK Period Full V 15.4 12.5 9.5 ns
DATA OUTPUT PARAMETERS
Output Delay1 (tOD) Full VI 2.5 4.8 6.5 2.5 4.8 6.5 2.5 4.8 6.5 ns
Pipeline Delay (Latency) Full V 5 5 5 Cycles
Aperture Delay 25°C V 2.4 2.4 2.4 ns
Aperture Uncertainty (Jitter) 25°C V 0.5 0.5 0.5 ps rms
Wake-Up Time2 25°C V 7 7 7 ms
OUT-OF-RANGE RECOVERY TIME 25°C V 1 1 1 Cycles
02874-A-002
t
A
t
PD
N–7 N–6 N–5 N–4 N–3 N–2 N–1 N N+1 N+2
ANALOG
INPUT
CLK
DATA
OUT
N–1
NN+1 N+2
N+3
N+4 N+5 N+6 N+7
N+8
Figure 2. Timing Diagram
1 Output delay is measured from CLK 50% transition to DATA 50% transition, with 5 pF load on each output.
2 Wake-up time is dependent on the value of decoupling capacitors; typical values shown with 0.1 µF and 10 µF capacitors on REFT and REFB.
AD9215 Data Sheet
Rev. B | Page 6 of 36
ABSOLUTE MAXIMUM RATINGS1
Table 5.
Mnemonic
With
Respect to
Min
Max
Unit
ELECTRICAL
AVDD AGND −0.3 +3.9 V
DRVDD DRGND 0.3 +3.9 V
AGND DRGND −0.3 +0.3 V
AVDD DRVDD −3.9 +3.9 V
Digital Outputs DRGND −0.3 DRVDD + 0.3 V
CLK, MODE AGND −0.3 AVDD + 0.3 V
VIN+, VIN− AGND −0.3 AVDD + 0.3 V
VREF AGND −0.3 AVDD + 0.3 V
SENSE AGND −0.3 AVDD + 0.3 V
REFB, REFT AGND −0.3 AVDD + 0.3 V
PDWN
AGND
−0.3
AVDD + 0.3
V
ENVIRONMENTAL2
Operating Temperature
Junction Temperature
Lead Temperature (10 sec)
Storage Temperature
−40 +85 °C
150 °C
300 °C
−65 +150 °C
NOTES
1Absolute maximum ratings are limiting values to be applied individually, and
beyond which the serviceability of the circuit may be impaired. Functional
operability is not necessarily implied. Exposure to absolute maximum rating
conditions for an extended period of time may affect device reliability.
2Typical thermal impedances 28-lead TSSOP: θJA = 67.7°C/W, 32-lead LFCSP:
θJA = 32.7°C/W; heat sink soldered down to ground plane.
EXPLANATION OF TEST LEVELS
Test Level
I 100% production tested.
II 100% production tested at 25°C and sample tested at spec-
ified temperatures.
III Sample tested only.
IV Parameter is guaranteed by design and characterization
testing.
V Parameter is a typical value only.
VI 100% production tested at 25°C; guaranteed by design and
characterization testing for industrial temperature range;
100% production tested at temperature extremes for mili-
tary devices.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Data Sheet AD9215
Rev. B | Page 7 of 36
PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS
02874-A-003
1
2
3
4
5
6
7
8
9
10
11
12
13
14
OR
MODE
SENSE
VREF
REFB
REFT
AVDD
AGND
VIN+
VIN–
AGND
AVDD
CLK
PDWN
D9 (MSB)
D8
D7
D6
DRVDD
DRGND
D5
D4
D3
D2
D1
D0 (LSB)
DNC
28
27
26
25
24
23
22
21
20
19
18
17
16
15
AD9215
TOP VIEW
(Not to Scale)
DNC = DO NOT CONNECT
DNC
Figure 3. TSSOP (RU-28)
0
2874-A-004
19
17
20
21
18
14
13
12
11
10
9
15
16
D5
D4
D3
D2
D1
(LSB) D0
DRGND
DRVDD
32
31
30
29
28
27
26
25
AVDD
AGN
D
VIN–
VIN+
AGN
D
AVDD
REFT
REFB
22
23
24
D8
D6
D9 (MSB )
OR
D7
MODE
SENSE
VREF
1
2
3
4
5
6
7
8
DNC
CLK
DNC
PDWN
DNC
DNC
DNC
DNC
AD9215
TOP VIEW
(Not to Scale)
NOTES
1. DNC = DO NO T CONNECT.
2. IT I S RECOMMENDED THAT T HE EXPOSED PAD BE SOLDERED
TO THE GROUND PLANE FOR THE LFCSP PACKAGE. THERE IS
AN INCRE AS E D RELIABI L IT Y O F THE S OL DE R JOI NTS, AND
THE MAXIMUM THERMAL CAPABIL I T Y O F T HE PACKAGE IS
ACHIEV ED WI T H T HE EXPOSED PAD SOLDERED TO THE
CUSTOME R BOARD.
Figure 4. LFCSP (CP-32-7)
Table 6. Pin Function Descriptions
TSSOP Pin No. LFCSP Pin No. Mnemonic Description
1 21 OR Out-of-Range Indicator.
2 22 MODE Data Format and Clock Duty Cycle Stabilizer (DCS) Mode Selection.
3 23 SENSE Reference Mode Selection.
4 24 VREF Voltage Reference Input/Output.
5 25 REFB Differential Reference (Negative).
6 26 REFT Differential Reference (Positive).
7, 12 27, 32 AVDD Analog Power Supply.
8, 11 28, 31 AGND Analog Ground.
9 29 VIN+ Analog Input Pin (+).
10 30 VIN− Analog Input Pin (−).
13 2 CLK Clock Input Pin.
14 4 PDWN Power-Down Function Selection (Active High).
15 to 16 1, 3, 5 to 8 DNC Do not connect, recommend floating this pin.
17 to 22,
25 to 28
9 to 14,
17 to 20
D0 (LSB) to
D9 (MSB)
Data Output Bits.
23 15 DRGND Digital Output Ground.
24 16 DRVDD Digital Output Driver Supply. Must be decoupled to DRGND with a
minimum 0.1 μF capacitor. Recommended decoupling is 0.1 μF in parallel with 10 μF.
N/A 33 EP
Exposed Pad. It is recommended that the exposed pad be soldered to the ground plane
for the LFCSP package. There is an increased reliability of the solder joints, and the
maximum thermal capability of the package is achieved with the exposed pad soldered
to the customer board.
AD9215 Data Sheet
Rev. B | Page 8 of 36
EQUIVALENT CIRCUITS
02874-A-005
AVDD
MODE
Figure 5. Equivalent Analog Input Circuit
02874-A-006
AVDD
MODE
20k
Figure 6. Equivalent MODE Input Circuit
02874-A-007
D9–D0,
OR
DRVDD
Figure 7. Equivalent Digital Output Circuit
02874-A-008
2.6k
2.6k
AVDD
CLK
Figure 8. Equivalent Digital Input Circuit
DEFINITIONS OF SPECIFICATIONS
Aperture Delay
Aperture delay is a measure of the sample-and-hold amplifier
(SHA) performance and is measured from the rising edge of the
clock input to when the input signal is held for conversion.
Aperture Jitter
Aperture jitter is the variation in aperture delay for successive
samples and can be manifested as frequency-dependent noise
on the input to the ADC.
Clock Pulse Width and Duty Cycle
Pulse width high is the minimum amount of time that the clock
pulse should be left in the Logic 1 state to achieve rated perfor-
mance. Pulse width low is the minimum time the clock pulse
should be left in the low state. At a given clock rate, these speci-
fications define an acceptable clock duty cycle.
Differential Nonlinearity (DNL, No Missing Codes)
An ideal ADC exhibits code transitions that are exactly 1 LSB
apart. DNL is the deviation from this ideal value. Guaranteed
no missing codes to 10-bit resolution indicate that all 1024
codes, respectively, must be present over all operating ranges.
Effective Number of Bits (ENOB)
For a sine wave, SINAD can be expressed in terms of the num-
ber of bits. Using the following formula, it is possible to obtain a
measure of performance expressed as N, the effective number of
bits
N = (SINAD1.76)/6.02
Thus, the effective number of bits for a device for sine wave
inputs at a given input frequency can be calculated directly
from its measured SINAD.
Gain Error
The first code transition should occur at an analog value 1/2
LSB above negative full scale. The last transition should occur at
an analog value 1 1/2 LSB below the positive full scale. Gain
error is the deviation of the actual difference between the first
and last code transitions and the ideal difference between the
first and last code transitions.
Integral Nonlinearity (INL)
INL refers to the deviation of each individual code from a line
drawn from “negative full scale” through “positive full scale.
The point used as negative full scale occurs 1/2 LSB before the
first code transition. Positive full scale is defined as a level 1 1/2
LSB beyond the last code transition. The deviation is measured
from the middle of each particular code to the true straight line.
Maximum Conversion Rate
The clock rate at which parametric testing is performed.
Minimum Conversion Rate
The clock rate at which the SNR of the lowest analog signal
frequency drops by no more than 3 dB below the guaranteed
limit.
Offset Error
The major carry transition should occur for an analog value 1/2
LSB below VIN+ = VIN−. Zero error is defined as the deviation
of the actual transition from that point.
Out-of-Range Recovery Time
Out-of-range recovery time is the time it takes for the ADC to
reacquire the analog input after a transient from 10% above
positive full scale to 10% above negative full scale, or from 10%
below negative full scale to 10% below positive full scale.
Output Propagation Delay
The delay between the clock logic threshold and the time when
all bits are within valid logic levels.
Power Supply Rejection
The specification shows the maximum change in full scale from
the value with the supply at the minimum limit to the value
Data Sheet AD9215
Rev. B | Page 9 of 36
with the supply at its maximum limit.
Signal-to-Noise and Distortion (SINAD) Ratio
SINAD is the ratio of the rms value of the measured input sig-
nal to the rms sum of all other spectral components below the
Nyquist frequency, including harmonics but excluding dc. The
value for SINAD is expressed in decibels.
Signal-to-Noise Ratio (SNR)
SNR is the ratio of the rms value of the measured input signal to
the rms sum of all other spectral components below the Nyquist
frequency, excluding the first six harmonics and dc. The value
for SNR is expressed in decibels.
Spurious-Free Dynamic Range (SFDR)
SFDR is the difference in dB between the rms amplitude of the
input signal and the peak spurious signal.
Temperature Drift
The temperature drift for zero error and gain error specifies the
maximum change from the initial (25°C) value to the value at
TMIN or TMAX.
Total Harmonic Distortion (THD)
THD is the ratio of the rms sum of the first six harmonic com-
ponents to the rms value of the measured input signal and is
expressed as a percentage or in decibels.
Two-Tone SFDR
The ratio of the rms value of either input tone to the rms value
of the peak spurious component. The peak spurious component
may or may not be an IMD product. It may be reported in dBc
(i.e., degrades as signal levels are lowered) or in dBFS (always
related back to converter full scale).
AD9215 Data Sheet
Rev. B | Page 10 of 36
TYPICAL PERFORMANCE CHARACTERISTICS
AVDD = 3.0 V, DRVDD = 2.5 V with DCS enabled, TA = 25°C, 2 V differential input, AIN = 0.5 dBFS, VREF = 1.0 V, unless
otherwise noted.
02874-A-062
–120
0
–20
–40
–60
–80
–100
0 52.5045.9439.3832.8126.2519.6913.136.56
AMPLITUDE (dBFS)
FREQUENCY (MHz)
A
IN
= –0.5dBFS
SNR = 58.0
ENOB = 9.4 BITS
SFDR = 75.5dB
Figure 9. Single-Tone 32k FFT with f
IN
= 10.3 MHZ, f
SAMPLE
= 105 MSPS
02874-A-063
–120
0
–20
–40
–60
–80
–100
0 52.5045.9439.3832.8126.2519.6913.136.56
AMPLITUDE (dBFS)
FREQUENCY (MHz)
A
IN
= –0.5dBFS
SNR = 57.8
ENOB = 9.4 BITS
SFDR = 75.0dB
Figure 10. Single-Tone 32k FFT with f
IN
= 70.3 MHz, f
SAMPLE
= 105 MSPS
02874-A-065
–120
0
–20
–40
–60
–80
–100
0 52.5045.9439.3832.8126.2519.6913.136.56
AMPLITUDE (dBFS)
FREQUENCY (MHz)
A
IN
= –0.5dBFS
SNR = 57.7
ENOB = 9.3 BITS
SFDR = 75dB
Figure 11. Single-Tone 32k FFT with fIN = 100.3 MHz, fSAMPLE = 105 MSPS
02874-A-012
ENCODE (MSPS)
dB
70
75
80
65
60
55
50 5 15 25 35 45 55 65 75 85
AIN = –0.5dBFS
1V p-p SFDR (dBc)
2V p-p SFDR (dBc)
1V p-p SNR (dB)
2V p-p SNR (dB)
Figure 12. AD9215-80 SNR/SFDR vs. fSAMPLE, fIN = 10.3 MHz
02874-A-013
ENCODE (MSPS)
dB
70
75
80
65
60
55
505 15 25 35 45 55 65
AIN = –0.5dBFS
2V p-p SFDR (dBc)
1V p-p SFDR (dBc)
1V p-p SNR (dB)
2V p-p SNR (dB)
Figure 13. AD9215-65 SNR/SFDR vs. fSAMPLE, fIN = 10.3 MHz
02874-A-066
55
60
65
70
75
80
85
0 10080604020
dB
f
SAMPLE
(MSPS)
2V p-p SFDR
2V p-p SNR
Figure 14. AD9215-105 SNR/SFDR vs. fSAMPLE, fIN = 10.3 MHz
Data Sheet AD9215
Rev. B | Page 11 of 36
02874-A-014
ANALOG INPUT LEVEL
dB
40
50
60
70
80
30
20
10
0
–50 –40–45 –35 –25–30 –10–15–20 –5 0
2V p-p SNR (dB)
2V p-p SFDR (dBc)
1V p-p SFDR (dBc)
1V p-p SNR (dB)
80dB REFERENCE LINE
Figure 15. AD9215-80 SNR/SFDR vs.
Analog Input Drive Level, f
SAMPLE
= 80 MSPS, f
IN
= 39.1 MHz
02874-A-067
0
80
70
60
50
40
30
20
10
–90 –80 –70 –60 –50 –40 –30 –20 –10 0
dB
ANALOG INPUT LEVEL (–dBFS)
2V p-p
SNR
1V p-p SNR
2 SFDR dBc
1V p-p SFDR (dBc)
–70dBFS
REFERENCE LINE
Figure 16. AD9215-105 SNR/SFDR vs.
Analog Input Drive Level, f
SAMPLE
= 105 MSPS, f
IN
= 50.3 MHz
02874-A-015
ANALOG INPUT LEVEL
dB
40
50
60
70
80
30
20
10
0
–50 –40–45 –35 –25–30 –10–15–20 –5 0
2V p-p SNR (dB)
2V p-p SFDR (dBc)
1V p-p SFDR (dBc)
1V p-p SNR (dB)
80dB REFERENCE LINE
Figure 17. AD9215-65 SNR/SFDR vs.
Analog Input Drive Level, fSAMPLE = 65 MSPS, fIN = 30.3 MHz
02874-A-072
50
55
60
65
70
75
80
0 50 100 150 200 250 300
SFDR
SNR
dB
FREQUENCY (MHz)
Figure 18. AD9215-105 SNR/SFDR vs.
fIN, AIN = −0.5 dBFS, fSAMPLE = 105 MSPS
02874-A-016
f
IN
(MHz)
dB
50
55
60
65
70
75
80
85
0 10050 150 250200 300
2V p-p SNR (dB)
2V p-p SFDR (dBc)
Figure 19. AD9215-80 SNR/SFDR vs.
fIN, AIN = 0.5 dBFS, fSAMPLE = 80 MSPS
02874-A-017
ANALOG INPUT (MHz)
dB
50
55
60
65
70
75
80
0 10050 150 250200 300
2V p-p SNR (dB)
2V p-p SFDR (dBc)
Figure 20. AD9215-65 SNR/SFDR vs.
fIN, AIN = 0.5 dBFS, fSAMPLE = 65 MSPS
AD9215 Data Sheet
Rev. B | Page 12 of 36
02874-A-060
FREQUENCY (MHz)
dB
–120
–100
–80
–60
–40
–20
0
0 52.50039.37526.25013.125
A
IN1
, A
IN2
= –7dBFS
SFDR = 74dBc
Figure 21. Two-Tone 32k FFT with fIN1 = 70.1 MHz,
and fIN2 = 71.1 MHz, fSAMPLE = 105 MSPS
02874-A-061
FREQUENCY (MHz)
dB
–120
–100
–80
–60
–40
–20
0
0 52.50039.37526.25013.125
A
IN1
, A
IN2
= –7dBFS
SFDR = 74dBc
Figure 22. Two-Tone 32k FFT with fIN1 = 100.3 MHz,
and fIN2 = 101.3 MHz, fSAMPLE = 105 MSPS
02874-A-068
0
80
70
60
50
40
30
20
10
–65 –55 –45 –35 –25 –15 –5
dB
AIN1, AIN2 (dBFS)
SFDR
80dBFS REFERENCE LINE
Figure 23. AD9215-105 Two-Tone SFDR vs. AIN,
fIN1 = 70.1 MHz, and fIN2 = 71.1 MHz, fSAMPLE = 105 MSPS
02874-A-073
0
80
70
60
50
40
30
20
10
–60 –5–10–15–20–25–30–35–40–45–50–55
dB
A
IN
(dBFS)
SFDR
80dBFS REFERENCE LINE
Figure 24. AD9215-80 Two-Tone SFDR vs. AIN, fIN1 = 100.3 MHz, and fIN2 =
101.3 MHz, fSAMPLE = 105 MSPS
02874-A-069
30
80
75
70
65
60
55
50
45
40
35
20 30 40 50 60 70 80
dB
CLOCK DUTY CYCLE HIGH (%)
SFDR DCS ON
SFDR DCS OFF
SNR DCS ON
SNR DCS OFF
Figure 25. SINAD, SFDR vs.
Clock Duty Cycle, fSAMPLE = 105 MSPS, fIN = 50.3 MH
02874-A-070
50
55
60
65
70
75
80
40200 20406080
dBc
TEMPERATURE (C)
2V p-p SFDR (dBc)
1V p-p SFDR (dBc)
2V p-p SINAD
1V p-p SINAD
Figure 26. SINAD, SFDR vs. Temperature,
fSAMPLE = 105 MSPS, fIN = 50 MHz
Data Sheet AD9215
Rev. B | Page 13 of 36
02874-A-025
TEMPERATURE (°C)
GAIN ERROR (ppm/°C)
10
20
30
0
40
–40
–30
–20
–10
–40 –20 0 20 6040 80
Figure 27. Gain vs. Temperature External 1 V Reference
02874-A-064
–0.5
–0.4
–0.3
–0.2
–0.1
0
0.1
0.2
0.3
0.4
0.5
10248967686405123842561280
DNL (LSB)
CODE
Figure 28. AD9215-105 Typical DNL, fSAMPLE = 105 MSPS, fIN = 2.3 MHz
02874-A-074
–0.6
–0.4
–0.2
0
0.2
0.4
0.6
10248967686405123842561280
INL (LSB)
CODE
Figure 29. AD9215-105 Typical INL, fSAMPLE = 105 MSPS, fIN = 2.3 MHz
AD9215 Data Sheet
Rev. B | Page 14 of 36
APPLYING THE AD9215 THEORY OF OPERATION
The AD9215 architecture consists of a front-end SHA followed
by a pipelined switched capacitor ADC. Each stage provides
sufficient overlap to correct for flash errors in the preceding
stages. The quantized outputs from each stage are combined
into a final 10-bit result in the digital correction logic. The pipe-
lined architecture permits the first stage to operate on a new
input sample, while the remaining stages operate on preceding
samples. Sampling occurs on the rising edge of the clock.
The input stage contains a differential SHA that can be config-
ured as ac-coupled or dc-coupled in differential or single-ended
modes. Each stage of the pipeline, excluding the last, consists of
a low resolution flash ADC connected to a switched capacitor
DAC and interstage residue amplifier (MDAC). The residue
amplifier magnifies the difference between the reconstructed
DAC output and the flash input for the next stage in the pipe-
line. Redundancy is used in each one of the stages to facilitate
digital correction of flash errors.
The output-staging block aligns the data, carries out the error
correction, and passes the data to the output buffers. The output
buffers are powered from a separate supply, allowing adjust-
ment of the output voltage swing. During power-down, the
output buffers go into a high impedance state.
Analog Input and Reference Overview
The analog input to the AD9215 is a differential switched
capacitor SHA that has been designed for optimum perfor-
mance while processing a differential input signal. The SHA
input can support a wide common-mode range and maintain
excellent performance, as shown in Figure 31. An input com-
mon-mode voltage of midsupply minimizes signal-dependent
errors and provides optimum performance.
02874-A-028
H
H
VIN+
VIN–
CPAR
CPAR
T
T
0.5pF
0.5pF
T
T
Figure 30. Switched-Capacitor SHA Input
The clock signal alternatively switches the SHA between sample
mode and hold mode (see Figure 30). When the SHA is
switched into sample mode, the signal source must be capable
of charging the sample capacitors and settling within one-half
of a clock cycle. A small resistor in series with each input can
help reduce the peak transient current required from the output
stage of the driving source. Also, a small shunt capacitor can be
placed across the inputs to provide dynamic charging currents.
This passive network creates a low-pass filter at the ADCs in-
put; therefore, the precise values are dependent upon the appli-
cation. In IF undersampling applications, any shunt capacitors
should be removed. In combination with the driving source
impedance, they would limit the input bandwidth.
The analog inputs of the AD9215 are not internally dc biased.
In ac-coupled applications, the user must provide this bias ex-
ternally. VCM = AV D D /2 is recommended for optimum perfor-
mance, but the device functions over a wider range with rea-
sonable performance (see Figure 31).
02874-A-071
40
45
50
55
60
65
70
75
80
85
0.25 0.75 1.25 1.75 2.25 2.75
dB
ANALOG INPUT COMMON-MODE VOLTAGE (V)
2V p-p SFDR
2V p-p SNR
Figure 31. AD9215-105 SNR, SFDR vs. Common-Mode Voltage
For best dynamic performance, the source impedances driving
VIN+ and VIN− should be matched such that common-mode
settling errors are symmetrical. These errors are reduced by the
common-mode rejection of the ADC.
An internal differential reference buffer creates positive and
negative reference voltages, REFT and REFB, respectively, that
define the span of the ADC core. The output common mode of
the reference buffer is set to midsupply, and the REFT and
REFB voltages and span are defined as
REFT = 1/2 (AVDD + VREF)
REFB = 1/2 (AV DD VREF)
Span = 2 × (REFT REFB) = 2 × VREF
It can be seen from the equations above that the REFT and
REFB voltages are symmetrical about the midsupply voltage
and, by definition, the input span is twice the value of the VREF
voltage.
The internal voltage reference can be pin-strapped to fixed val-
ues of 0.5 V or 1.0 V or adjusted within the same range as dis-
cussed in the Internal Reference Connection section. Maximum
SNR performance is achieved with the AD9215 set to the largest
input span of 2 V p-p. The relative SNR degradation is 3 dB
Data Sheet AD9215
Rev. B | Page 15 of 36
when changing from 2 V p-p mode to 1 V p-p mode.
The SHA may be driven from a source that keeps the signal
peaks within the allowable range for the selected reference volt-
age. The minimum and maximum common-mode input levels
are defined as
VCMMIN = VREF/2
VCMMAX = (AV D D + VREF)/2
The minimum common-mode input level allows the AD9215 to
accommodate ground-referenced inputs.
Although optimum performance is achieved with a differential
input, a single-ended source may be driven into VIN+ or VIN−.
In this configuration, one input accepts the signal, while the
opposite input should be set to midscale by connecting it to an
appropriate reference. For example, a 2 V p-p signal may be
applied to VIN+ while a 1 V reference is applied to VIN−. The
AD9215 then accepts a signal varying between 2 V and 0 V. In
the single-ended configuration, distortion performance may
degrade significantly as compared to the differential case. How-
ever, the effect is less noticeable at lower input frequencies.
Differential Input Configurations
As previously detailed, optimum performance is achieved while
driving the AD9215 in a differential input configuration. For
baseband applications, the AD8138 differential driver provides
excellent performance and a flexible interface to the ADC. The
output common-mode voltage of the AD8138 is easily set to
AV D D /2, and the driver can be configured in a Sallen Key filter
topology to provide band limiting of the input signal.
02874-A-030
AD8138 AD9215
VIN+
VIN–
AVDD
AGND
1V p-p
R
R
C
C
499
499
499
523
49.9
1k
1k
0.1µF
V
CM
Figure 32. Differential Input Configuration Using the AD8138
At input frequencies in the second Nyquist zone and above, the
performance of most amplifiers is not adequate to achieve the
true performance of the AD9215. This is especially true in IF
undersampling applications where frequencies in the 70 MHz to
200 MHz range are being sampled. For these applications, differ-
ential transformer coupling is the recommended input configura-
tion. The value of the shunt capacitor is dependant on the input
frequency and source impedance and should be reduced or re-
moved. An example of this is shown in Figure 33.
02874-A-031
AD9215
VIN+
VIN–
AVDD
AGND
2Vp-p
R
R
C
C
49.9
0.1µF
AVDD
1k
1k
Figure 33. Differential Transformer-Coupled Configuration
The signal characteristics must be considered when selecting a
transformer. Most RF transformers saturate at frequencies
below a few MHz, and excessive signal power can also cause
core saturation, which leads to distortion.
Single-Ended Input Configuration
A single-ended input may provide adequate performance in
cost-sensitive applications. In this configuration, there is a deg-
radation in SFDR and distortion performance due to the large
input common-mode swing. However, if the source impedances
on each input are kept matched, there should be little effect on
SNR performance. Figure 34 details a typical single-ended input
configuration.
02874-A-032
2V p-p
R
R
C
C
49.90.1µF
10µF
10µF 0.1µF
AD9215
VIN+
VIN–
AVDD
AGND
AVDD
1k
1k
1k
1k
Figure 34. Single-Ended Input Configuration
CLOCK INPUT AND CONSIDERATIONS
Typical high speed ADCs use both clock edges to generate a
variety of internal timing signals, and as a result may be sensi-
tive to clock duty cycle. Commonly, a 5% tolerance is required
on the clock duty cycle to maintain dynamic performance char-
acteristics. The AD9215 contains a clock duty cycle stabilizer
that retimes the nonsampling edge, providing an internal clock
signal with a nominal 50% duty cycle. This allows a wide range
of clock input duty cycles without affecting the performance of
the AD9215. As shown in Figure 25, noise and distortion per-
formance are nearly flat over a 50% range of duty cycle. For best
ac performance, enabling the duty cycle stabilizer is recom-
mended for all applications.
The duty cycle stabilizer uses a delay-locked loop (DLL) to cre-
ate the nonsampling edge. As a result, any changes to the sam-
pling frequency require approximately 100 clock cycles to allow
the DLL to acquire and lock to the new rate.
AD9215 Data Sheet
Rev. B | Page 16 of 36
Table 7. Reference Configuration Summary
Selected Mode
External SENSE
Connection
Internal Op Amp
Configuration
Resulting VREF
(V)
Resulting Differential Span
(V p-p)
Externally Supplied Reference AVDD N/A N/A 2 × External Reference
Internal 0.5 V Reference VREF Voltage Follower (G = 1) 0.5 1.0
Programmed Variable
Reference
External Divider Noninverting (1 < G < 2) 0.5 × (1 + R2/R1)
2 × VREF
Internally Programmed 1 V
Reference
AGND to 0.2 V Internal Divider 1.0 2.0
Table 8. Digital Output Coding
Code
VIN+ − VIN− Input Span =
2 V p-p (V)
VIN+ − VIN− Input Span =
1 V p-p (V)
Digital Output Offset Binary
(D9••••••D0)
Digital Output Twos
Complement (D9••••••D0)
1023 1.000 0.500 11 1111 1111 01 1111 1111
512 0 0 10 0000 0000 00 0000 0000
511 −0.00195 −0.000978 01 1111 1111 11 1111 1111
0 −1.00 −0.5000 00 0000 0000 10 0000 0000
High speed, high resolution ADCs are sensitive to the quality
of the clock input. The degradation in SNR at a given full-scale
input frequency (fINPUT) due only to aperture jitter (tA) can be
calculated with the following equation
SNR Degradation = 20 × log10 [2 × π × fINPUT × tA]
In the equation, the rms aperture jitter, tA, represents the root-
sum square of all jitter sources, which include the clock input,
analog input signal, and ADC aperture jitter specification.
Undersampling applications are particularly sensitive to jitter.
The clock input should be treated as an analog signal in cases
where aperture jitter may affect the dynamic range of the
AD9215. Power supplies for clock drivers should be separated
from the ADC output driver supplies to avoid modulating the
clock signal with digital noise. Low jitter, crystal-controlled
oscillators make the best clock sources. If the clock is generated
from another type of source (by gating, dividing, or other
methods), it should be retimed by the original clock at the last
step.
Power Dissipation and Standby Mode
As shown in Figure 35, the power dissipated by the AD9215 is
proportional to its sample rate. The digital power dissipation
does not vary substantially between the three speed grades
because it is determined primarily by the strength of the digital
drivers and the load on each output bit. The maximum DRVDD
current can be calculated as
IDRVDD = VDRVDD × CLOAD × fCLOCK × N
where N is the number of output bits, 10 in the case of the
AD9215. This maximum current is for the condition of every
output bit switching on every clock cycle, which can only occur
for a full-scale square wave at the Nyquist frequency, fCLOCK/2. In
practice, the DRVDD current is established by the average
number of output bits switching, which are determined by the
encode rate and the characteristics of the analog input signal.
Digital power consumption can be minimized by reducing the
capacitive load presented to the output drivers. The data in Fig-
ure 35 was taken with a 5 pF load on each output driver.
02874-A-075
15
35
30
25
20
40
1055 15 25 35 45 55 65 75 85 95
I
AVDD
(mA)
I
DRVDD
–1
1
3
5
7
9
11
13
15
fSAMPLE
(MSPS)
AD9215-105 I
AVDD
AD9215-65/80 I
AVDD
I
DRVDD
Figure 35. Supply Current vs. fSAMPLE for fIN = 10.3 MHz
The analog circuitry is optimally biased so that each speed
grade provides excellent performance while affording reduced
power consumption. Each speed grade dissipates a baseline
power at low sample rates that increases linearly with the clock
frequency.
By asserting the PDWN pin high, the AD9215 is placed in
standby mode. In this state, the ADC typically dissipates 1 mW
if the CLK and analog inputs are static. During standby, the
output drivers are placed in a high impedance state. Reasserting
the PDWN pin low returns the AD9215 into its normal opera-
tional mode.
In standby mode, low power dissipation is achieved by shutting
down the reference, reference buffer, and biasing networks. The
Data Sheet AD9215
Rev. B | Page 17 of 36
decoupling capacitors on REFT and REFB are discharged when
entering standby mode and then must be recharged when
returning to normal operation. As a result, the wake-up time is
related to the time spent in standby mode, and shorter standby
cycles result in proportionally shorter wake-up times. With the
recommended 0.1 μF and 10 μF decoupling capacitors on REFT
and REFB, it takes approximately one second to fully discharge
the reference buffer decoupling capacitors and 7 ms to restore
full operation.
Digital Outputs
The AD9215 output drivers can be configured to interface with
2.5 V or 3.3 V logic families by matching DRVDD to the digital
supply of the interfaced logic. The output drivers are sized to
provide sufficient output current to drive a wide variety of logic
families. However, large drive currents tend to cause current
glitches on the supplies that may affect converter performance.
Applications requiring the ADC to drive large capacitive loads
or large fanouts may require external buffers or latches.
Timing
The AD9215 provides latched data outputs with a pipeline delay
of five clock cycles. Data outputs are available one propagation
delay (tOD) after the rising edge of the clock signal. Refer to Fig-
ure 2 for a detailed timing diagram.
The length of the output data lines and loads placed on them
should be minimized to reduce transients within the AD9215;
these transients can detract from the converter’s dynamic per-
formance.
The lowest typical conversion rate of the AD9215 is 5 MSPS. At
clock rates below 5 MSPS, dynamic performance may degrade.
Voltage Reference
A stable and accurate 0.5 V voltage reference is built into the
AD9215. The input range can be adjusted by varying the refer-
ence voltage applied to the AD9215, using either the internal
reference or an externally applied reference voltage. The input
span of the ADC tracks reference voltage changes linearly. Max-
imum SNR and DNL performance is achieved with the AD9215
set to the largest input span of 2 V p-p.
Internal Reference Connection
A comparator within the AD9215 detects the potential at the
SENSE pin and configures the reference into four possible
states, which are summarized in Table 1. If SENSE is grounded,
the reference amplifier switch is connected to the internal resis-
tor divider (see Figure 36), setting VREF to 1 V. Connecting the
SENSE pin to the VREF pin switches the amplifier output to the
SENSE pin, configuring the internal op amp circuit as a voltage
follower and providing a 0.5 V reference output. If an external
resistor divider is connected as shown in Figure 37, the switch is
again set to the SENSE pin. This puts the reference amplifier in a
noninverting mode with the VREF output defined as
R1
R2
VREF 15.0
02874-A-034
10F+0.1F
VREF
SENSE
0.5V
7k
7k
AD9215
VIN–
VIN+
REFT
0.1F
0.1F 10F
0.1F
REFB
SELECT
LOGIC
ADC
CORE
Figure 36. Internal Reference Configuration
In all reference configurations, REFT and REFB drive the ADC
conversion core and establish its input span. The input range of
the ADC always equals twice the voltage at the reference pin for
either an internal or an external reference.
02874-A-035
10F+0.1F
VREF
R2
R1
SENSE
0.5V
AD9215
VIN–
VIN+
REFT
0.1F
0.1F 10F
0.1F
REFB
SELECT
LOGIC
ADC
CORE
Figure 37. Programmable Reference Configuration
If the internal reference of the AD9215 is used to drive multiple
converters to improve gain matching, the loading of the refer-
ence by the other converters must be considered. Figure 38 de-
picts how the internal reference voltage is affected by loading.
AD9215 Data Sheet
Rev. B | Page 18 of 36
02874-A-036
I
LOAD
(mA)
VREF ERROR (%)
0
0.05
–0.25
–0.20
–0.15
–0.10
–0.05
0 0.5 1.0 1.5 2.0 2.5 3.0
VREF = 0.5V
VREF = 1.0V
Figure 38. VREF Accuracy vs. Load
External Reference Operation
The use of an external reference may be necessary to enhance
the gain accuracy of the ADC or improve thermal drift charac-
teristics. When multiple ADCs track one another, a single refer-
ence (internal or external) may be necessary to reduce gain
matching errors to an acceptable level. A high precision external
reference may also be selected to provide lower gain and offset
temperature drift. Figure 39 shows the typical drift characteris-
tics of the internal reference in both 1 V and 0.5 V modes.
02874-A-037
TEMPERATURE (°C)
VREF ERROR (%)
0.4
0.5
0.6
0.3
0.2
0.1
0
–40 –20 0 20 40 60 80
VREF = 1.0V
VREF = 0.5V
Figure 39. Typical VREF Drift
When the SENSE pin is tied to AVDD, the internal reference is
disabled, allowing the use of an external reference. An internal
reference buffer loads the external reference with an equivalent
7 kΩ load. The internal buffer still generates the positive and
negative full-scale references, REFT and REFB, for the ADC
core. The input span is always twice the value of the reference
voltage; therefore, the external reference must be limited to a
maximum of 1 V.
Operational Mode Selection
As discussed earlier, the AD9215 can output data in either offset
binary or twos complement format. There is also a provision for
enabling or disabling the clock duty cycle stabilizer (DCS). The
MODE pin is a multilevel input that controls the data format
and DCS state. For best ac performance, enabling the duty cycle
stabilizer is recommended for all applications. The input
threshold values and corresponding mode selections are out-
lined in Table 9.
As detailed in Table 9, the data format can be selected for either
offset binary or twos complement.
Table 9. Mode Selection
MODE Voltage Data Format Duty Cycle Stabilizer
AVDD Twos Complement Disabled
2/3 AVDD Twos Complement Enabled
1/3 AVDD Offset Binary Enabled
AGND (Default) Offset Binary Disabled
The MODE pin is internally pulled down to AGND by a 20 kΩ
resistor.
EVALUATION BOARD
The AD9215 evaluation board is no longer in production. The
following evaluation board documentation is provided for in-
formational purposes only.
The AD921