LTC3547BIDDB#TRPBF - ANALOG DEVICES

LTC3547B

3547bfb

OUTPUT CURRENT (mA)

EFFICIENCY (%)

POWER LOSS (W)

100

1 100 1000

3547b TA01b

0.1

0.01

0.001

VIN = 2.7V

VIN = 3.6V

VIN = 4.2V

Dual Monolithic

300mA Synchronous

Step-Down Regulator

The LTC

3547B is a dual, 2.25MHz, constant-frequency,

synchronous step-down DC/DC converter in a tiny 3mm ×

2mm DFN package. 100% duty cycle provides low drop-

out operation, extending battery life in portable systems.

Low output voltages are supported with the 0.6V feed-

back reference voltage. Each regulator can supply 300mA

continuous output current.

The input voltage range is 2.5V to 5.5V, making it ideal

for Li-Ion polymer and USB powered applications. Supply

current is only 1μA in shutdown.

An internally set 2.25MHz switching frequency allows the

use of tiny surface mount inductors and capacitors. Inter-

nal soft-start reduces inrush current during start-up. All

outputs are internally compensated to work with ceramic

capacitors. The LTC3547B is available in a low proﬁ le

(0.75mm) 3mm × 2mm DFN package. The LTC3547B

is also available in a ﬁ xed output voltage conﬁ guration

selected via internal resistor dividers (see Table 2).

■ Cellular Telephones

■ Digital Still Cameras

■ Wireless and DSL Modems

■ Portable Media Players

■ PDAs/Palmtop PCs

■ High Efﬁ ciency (Up to 96%) Dual Step-Down

Outputs

■ 300mA Output Current per Channel at VIN = 3V

■ 2.25MHz Constant-Frequency Operation

■ 2.5V to 5.5V Input Voltage Range

■ Low Dropout Operation: 100% Duty Cycle

■ No Schottky Diodes Required

■ Internally Compensated for All Ceramic Capacitors

■ Independent Internal Soft-Start for Each Channel

■ Available in Fixed Output Versions

■ Current Mode Operation for Excellent Line and Load

Transient Response

■ 0.6V Reference Allows Low Output Voltages

■ Short-Circuit Protected

■ Ultralow Shutdown Current: IQ < 1μA

■ Low Proﬁ le (0.75mm) 8-Lead 3mm × 2mm

DFN Package

Dual Monolithic Buck Regulator in 8-Lead 3mm × 2mm DFN

APPLICATIO S

FEATURES DESCRIPTIO

TYPICAL APPLICATIO

, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.

Burst Mode is a registered trademark of Linear Technology Corporation. All other

trademarks are the property of their respective owners. Protected by U.S. Patents,

including 6580258, 5481178, 6304066, 6127815, 6498466, 6611131.

VIN

RUN2 RUN1

LTC3547B

VFB2

SW2 SW1

VFB1

10pF 10pF

GND

VIN

2.5V TO 5.5V

VOUT2

1.8V AT

300mA

VOUT1

2.5V AT

300mA

3547b TA01

237k 150k

475k

4.7μH 4.7μH

475k

4.7μF

Efﬁ ciency vs Output Current

for VOUT = 2.5V

LTC3547B

3547bfb

VIN

LTC3547BE .............................................. –0.3V to 6V

LTC3547BI ............................................... –0.3V to 7V

VFB1, VFB2 ......................................... –0.3V to VIN +0.3V

RUN1, RUN2 ..................................... –0.3V to VIN +0.3V

SW1, SW2 (DC) ................................ –0.3V to VIN +0.3V

P-Channel Switch Source Current (DC)................500mA

N-Channel Switch Sink Current (DC) ...................500mA

Peak SW Sink and Source Current (Note 5) .........700mA

Ambient Operating Temperature Range (Note 2)

LTC3547BE .......................................... –40°C to 85°C

LTC3547BI ......................................... –40°C to 125°C

Maximum Junction Temperature........................... 125°C

Storage Temperature Range ................... –65°C to 125°C

(Note 1)

The ● denotes the speciﬁ cations which apply over the full operating

temperature range, otherwise speciﬁ cations are at TA = 25°C. VIN = 3.6V, unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

VIN VIN Operating Voltage LTC3547BE, LTC3547BE-1,

LTC3547BI

●

2.5

5.5

6.5

VUV VIN Undervoltage Lockout VIN Low to High ● 2.2 2.5 V

IFB Feedback Pin Input Current LTC3547BE, VFB = VFBREG

LTC3547BE-1, VFB = VFBREG

●

±30

μA

VFBREG1 Regulated Feedback Voltage (VFB1) LTC3547BE, 0°C ≤ TA ≤ 85°C

LTC3547BE, –40°C ≤ TA ≤ 85°C

LTC3547BI, –40°C ≤ TA ≤ 125°C

LTC3547BE-1, 0°C ≤ TA ≤ 85°C

LTC3547BE-1, –40°C ≤ TA ≤ 85°C

●

0.590

0.588

1.770

1.764

0.600

1.800

0.610

0.612

1.830

1.836

VFBREG2 Regulated Feedback Voltage (VFB2) LTC3547BE, 0°C ≤ TA ≤ 85°C

LTC3547BE, –40°C ≤ TA ≤ 85°C

LTC3547BI, –40°C ≤ TA ≤ 125°C

LTC3547BE-1, 0°C ≤ TA ≤ 85°C

LTC3547BE-1, –40°C ≤ TA ≤ 85°C

●

0.590

0.588

1.180

1.176

0.600

1.200

0.610

0.612

1.220

1.224

ELECTRICAL CHARACTERISTICS

ABSOLUTE AXI U RATI GS

TOP VIEW

DDB PACKAGE

8-LEAD (3mm × 2mm) PLASTIC DFN

1VFB1

RUN1

VIN

SW1

VFB2

RUN2

SW2

GND

TJMAX = 125°C, θJA = 76°C/W

EXPOSED PAD (PIN 9) IS GND

MUST BE SOLDERED TO PCB

ORDER INFORMATION

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LTC3547BEDDB#PBF LTC3547BEDDB#TRPBF LCPD 8-Lead (3mm × 2mm) Plastic DFN –40°C to 85°C

LTC3547BEDDB-1#PBF LTC3547BEDDB-1#TRPBF LCPF 8-Lead (3mm × 2mm) Plastic DFN –40°C to 85°C

LTC3547BIDDB #PBF LTC3547BIDDB #TRPBF LCPD 8-Lead (3mm × 2mm) Plastic DFN –40°C to 125°C

Consult LTC Marketing for parts speciﬁ ed with wider operating temperature ranges. *Temperature grades are identiﬁ ed by a label on the shipping container.

Consult LTC Marketing for information on non-standard lead based ﬁ nish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel speciﬁ cations, go to: http://www.linear.com/tapeandreel/

PIN CONFIGURATION

LTC3547B

3547bfb

VIN (V)

2.5

100

3547b G02

3 3.5 4.5 5.5

EFFICIENCY (%)

VOUT = 1.8V

IOUT = 0.1mA

IOUT = 1mA

IOUT = 10mA

IOUT = 100mA

IOUT = 300mA

10μs/DIV

VOUT

50mV/DIV

50mA/DIV

SW,

AC-COUPLED

5V/DIV

3547b G01

VIN = 3.6V

VOUT = 1.8V

ILOAD = 1mA

TEMPERATURE (ºC)

–50

400

350

300

250

200

150

100 25 75

3547b G03

–25 0 50 100

SUPPLY CURRENT (μA)

ILOAD = 0A

VIN = 5.5V

VIN = 2.7V

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

ΔVLINEREG Reference Voltage Line Regulation VIN = 2.5V to 5.5V 0.3 0.5 %/V

ΔVLOADREG Output Voltage Load Regulation ILOAD = 50mA to 300mA 0.5 %

ISInput DC Supply Current

Active Mode (Note 4)

Shutdown

VFB1 = VFB2 = 0.95 × VFBREG

RUN1 = RUN2 = 0V, VIN = 5.5V

450

0.1

700

μA

fOSC Oscillator Frequency VFB = 0.6V ●1.8 2.25 2.7 MHz

ILIM Peak Switch Current Limit

Channel 1 (300mA)

Channel 2 (300mA)

VIN = 2.5V, VFB < VFBREG, Duty Cycle < 35%

400

550

RDS(ON) Channel 1 (Note 3)

Top Switch On-Resistance

Bottom Switch On-Resistance

Channel 2 (Note 3)

Top Switch On-Resistance

Bottom Switch On-Resistance

VIN = 3.6V, ISW = 100mA

0.8

0.75

0.8

0.75

1.05

ISW(LKG) Switch Leakage Current VIN = 5V, VRUN = 0V 0.01 1 μA

tSOFTSTART Soft-Start Time VFB From 10% to 90% Full-Scale 0.450 0.650 0.850 ms

VRUN RUN Threshold High ●0.4 1 1.2 V

IRUN RUN Leakage Current ●0.01 1 μA

The ● denotes the speciﬁ cations which apply over the full operating

temperature range, otherwise speciﬁ cations are at TA = 25°C. VCC = 3.6V, unless otherwise noted.

ELECTRICAL CHARACTERISTICS

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 2: The LTC3547BE is guaranteed to meet speciﬁ ed performance

from 0°C to 85°C. Speciﬁ cations over the –40°C and 85°C operating

temperature range are assured by design, characterization and correlation

with statistical process controls. The LTC3547BI is guaranteed over the

–40°C to 125°C operating temperature range.

Note 3: The DFN switch on-resistance is guaranteed by correlation to

wafer level measurements.

Note 4: Dynamic supply current is higher due to the internal gate charge

being delivered at the switching frequency.

Note 5: Guaranteed by long-term current density limitations.

Note 6: This IC includes overtemperature protection that is intended

to protect the device during momentary overload conditions. Junction

temperature will exceed 125°C when overtemperature protection is active.

Continuous operation above the speciﬁ ed maximum operating junction

temperature may impair device reliability.

Pulse Skip Mode Operation Efﬁ ciency vs Input Voltage

Supply Current vs Temperature

TYPICAL PERFOR A CE CHARACTERISTICS

TA = 25°C unless otherwise speciﬁ ed.

LTC3547B

3547bfb

VIN (V)

2.5

RDS(ON) (Ω)

3547b G08

0.9

0.8

0.6

3 3.5 4.5

0.5

0.4

1.0

0.7

5 5.5 6

SYNCHRONOUS

SWITCH

MAIN SWITCH

TEMPERATURE (°C)

–50

RDS(ON) (Ω)

1.2

1.1

3547b G09

0.8

0.6

–25 0 50

0.5

0.4

1.3

1.0

0.9

0.7

75 100 125

MAIN SWITCH

SYNCHRONOUS

SWITCH

VIN = 2.7V

VIN = 3.6V

VIN = 4.2V

OUTPUT CURRENT (mA)

EFFICIENCY (%)

100

0.1 10 100 1000

3547b G10

VIN = 2.7V

VIN = 3.6V

VIN = 4.2V

VOUT = 1.2V

OUTPUT CURRENT (mA)

EFFICIENCY (%)

100

0.1 10 100 1000

3547b G11

VIN = 2.7V

VIN = 3.6V

VIN = 4.2V

VOUT = 1.8V

TEMPERATURE (°C)

–50

VFB (% ERROR)

3547b G07

0.5

–0.5

–25 0 50

–1.0

–1.5

1.5

1.0

75 100

Reference Voltage

vs Temperature

RDS(ON) vs Input Voltage

RDS(ON) vs Temperature

Efﬁ ciency vs Load Current

TYPICAL PERFOR A CE CHARACTERISTICS

TEMPERATURE (°C)

–50

FREQUENCY (MHz)

2.5

3547b G04

2.2

2.0

–25 0 50

1.9

1.8

2.6

2.4

2.3

2.1

75 100 125

VIN = 4.2V

VIN = 3.6V

VIN = 2.7V

TEMPERATURE (°C)

–50

LEAKAGE CURRENT (nA)

3547b G05

–25 0 50

75 100 125

SYNCHRONOUS

SWITCH

MAIN SWITCH

VIN (V)

2.5

LEAKAGE CURRENT (pA)

3547b G06

400

200

3 3.5 4.5

100

500

300

5 5.5 6

SYNCHRONOUS

SWITCH

MAIN SWITCH

Oscillator Frequency

vs Temperature

Switch Leakage vs Temperature

Switch Leakage vs Input Voltage

LTC3547B

3547bfb

LOAD CURRENT (mA)

VOUT ERROR (%)

150

3547b G13

0.8

0.6

0.2

50 100 200

–0.2

1.2

1.O

0.4

250 300 350

PULSE SKIP MODE

OPERATION

VIN = 3.6V VOUT = 1.2V

VOUT = 1.8V

VOUT = 2.5V

VIN (V)

2.5

VOUT ERROR (%)

3547b G14

0.4

0.2

–0.2

3 3.5 4.5

–0.4

–0.6

0.6

5 5.5

VOUT = 1.8V

ILOAD = 100mA

OUTPUT CURRENT (mA)

EFFICIENCY (%)

100

0.1 10 100 1000

3547b G12

VIN = 2.7V

VIN = 3.6V

VIN = 4.2V

VOUT = 2.5V

200μs/DIV

100mA/DIV

RUN

2V/DIV

VOUT

1V/DIV

3547b G15

VIN = 3.6V

VOUT = 1.8V

ILOAD = 0A

Efﬁ ciency vs Load Current

Load Regulation

Line Regulation

Start-Up From Shutdown

Load Step

TYPICAL PERFOR A CE CHARACTERISTICS

250μs/DIV

200mA/DIV

RUN

2V/DIV

VOUT

1V/DIV

3547b G16

VIN = 3.6V

VOUT = 1.8V

ILOAD = 300mA

10μs/DIV

200mA/DIV

ILOAD

200mA/DIV

VOUT,

AC-COUPLED

100mV/DIV

3547b G19

VIN = 3.6V

VOUT = 1.8V

ILOAD = 50mA TO 300mA

10μs/DIV

200mA/DIV

ILOAD

200mA/DIV

VOUT,

AC-COUPLED

100mV/DIV

3547b G18

VIN = 3.6V

VOUT = 1.8V

ILOAD = 20mA TO 300mA

10μs/DIV

200mA/DIV

ILOAD

200mA/DIV

VOUT,

AC-COUPLED

100mV/DIV

3547b G17

VIN = 3.6V

VOUT = 1.8V

ILOAD = 0mA TO 300mA

LTC3547B

3547bfb

PI FU CTIO S

UUU

VFB1 (Pin 1): Regulator 1 Output Feedback. Receives

the feedback voltage from the external resistor divider

across the regulator 1 output. Nominal voltage for this

pin is 0.6V.

RUN1 (Pin 2): Regulator 1 Enable. Forcing this pin to

VIN enables regulator 1, while forcing it to GND causes

regulator 1 to shut down.

VIN (Pin 3): Main Power Supply. Must be closely decou-

pled to GND.

SW1 (Pin 4): Regulator 1 Switch Node Connection to the

Inductor. This pin swings from VIN to GND.

GND (Pin 5): Ground. Connect to the (–) terminal of COUT

and the (–) terminal of CIN.

SW2 (Pin 6): Regulator 2 Switch Node Connection to

the Inductor. This pin swings from VIN to GND.

RUN2 (Pin 7): Regulator 2 Enable. Forcing this pin to

VIN enables regulator 2, while forcing it to GND causes

regulator 2 to shut down.

VFB2 (Pin 8): Regulator 2 Output Feedback. Receives

the feedback voltage from the external resistor divider

across the regulator 2 output. Nominal voltage for this

pin is 0.6V.

Exposed Pad (Pin 9): Electrically Connected to GND.

Must be soldered to the PCB for optimum thermal

performance.

–

–+

–

VSLEEP

ITH

SWITCHING

LOGIC

AND

BLANKING

CIRCUIT

LATCH

PULSE SKIP

–

ICOMP

IRCMP

ANTI

SHOOT-

THRU

SLOPE

COMP

SLEEP

0.6V REF OSC

OSC

REGULATOR 2 (IDENTICAL TO REGULATOR 1)

SHUTDOWN

REGULATOR 1

SW1

SW2

5Ω

3547 FD

RUN1

RUN2

VFB2

VFB1

VIN

GND

0.6V

SOFT-START

FUNCTIONAL DIAGRAM

LTC3547B

3547bfb

OPERATIO

The LTC3547B uses a constant-frequency current mode

architecture. The operating frequency is set at 2.25MHz.

Both channels share the same clock and run in-phase.

The output voltage is set by an external resistor divider

returned to the VFB pins. An error ampliﬁ er compares the

divided output voltage with a reference voltage of 0.6V and

regulates the peak inductor current accordingly.

Main Control Loop

During normal operation, the top power switch (P-channel

MOSFET) is turned on at the beginning of a clock cycle

when the VFB voltage is below the reference voltage. The

current into the inductor and the load increases until the

peak inductor current (controlled by ITH) is reached. The

RS latch turns off the synchronous switch and energy

stored in the inductor is discharged through the bottom

switch (N-channel MOSFET) into the load until the next

clock cycle begins, or until the inductor current begins to

reverse (sensed by the IRCMP comparator).

The peak inductor current is controlled by the internally

compensated ITH voltage, which is the output of the er-

ror ampliﬁ er. This ampliﬁ er regulates the VFB pin to the

internal 0.6V reference by adjusting the peak inductor

current accordingly.

At very low load currents, the LTC3547B automatically

transitions from constant frequency operation to discon-

tinuous operation, where the regulator begins skipping

pulses. Even though the total power loss decreases with

load, the efﬁ ciency of the switching regulator will drop,

because the power delivered to the output becomes very

small. For applications where light load efﬁ ciency is a

priority, consider using the LTC3547 instead.

Dropout Operation

When the input supply voltage decreases toward the out-

put voltage the duty cycle increases to 100%, which is the

dropout condition. In dropout, the PMOS switch is turned

on continuously with the output voltage being equal to the

input voltage minus the voltage drops across the internal

P-channel MOSFET and the inductor.

An important design consideration is that the RDS(ON)

of the P-channel switch increases with decreasing input

supply voltage (see Typical Performance Characteristics).

Therefore, the user should calculate the worst-case power

dissipation when the LTC3547B is used at 100% duty cycle

with low input voltage (see Thermal Considerations in the

Applications Information Section).

Soft-Start

In order to minimize the inrush current on the input by-

pass capacitor, the LTC3547B slowly ramps up the output

voltage during start-up. Whenever the RUN1 or RUN2 pin is

pulled high, the corresponding output will ramp from zero

to full-scale over a time period of approximately 650μs. This

prevents the LTC3547B from having to quickly charge the

output capacitor and thus supplying an excessive amount

of instantaneous current.

Short-Circuit Protection

When either regulator output is shorted to ground, the

corresponding internal N-channel switch is forced on for

a longer time period for each cycle in order to allow the

inductor to discharge, thus preventing current runaway.

This technique has the effect of decreasing switching

frequency. Once the short is removed, normal operation

resumes and the regulator output will return to its nominal

voltage.

(Refer to Functional Diagram )

LTC3547B

3547bfb

APPLICATIO S I FOR ATIO

WUUU

A general LTC3547B application circuit is shown in

Figure 1. External component selection is driven by the

load requirement, and begins with the selection of the

inductor L. Once the inductor is chosen, CIN and COUT

can be selected.

Inductor Selection

Although the inductor does not inﬂ uence the operat-

ing frequency, the inductor value has a direct effect on

ripple current. The inductor ripple current ΔIL decreases

with higher inductance and increases with higher VIN

or VOUT:

IL=VOUT

fO• L •1VOUT

VIN









(1)

Accepting larger values of ΔIL allows the use of low

inductances, but results in higher output voltage ripple,

greater core losses, and lower output current capability.

A reasonable starting point for setting ripple current

is 40% of the maximum output load current. So, for a

300mA regulator, ΔIL = 120mA (40% of 300mA).

Inductor Core Selection

Different core materials and shapes will change the

size/current and price/current relationship of an induc-

tor. Toroid or shielded pot cores in ferrite or permalloy

materials are small and do not radiate much energy, but

generally cost more than powdered iron core inductors

with similar electrical characteristics. The choice of which

style inductor to use often depends more on the price vs

size requirements, and any radiated ﬁ eld/EMI requirements,

than on what the LTC3547B requires to operate. Table 1

shows some typical surface mount inductors that work

well in LTC3547B applications.

Figure 1. LTC3547B General Schematic

CF2 CF1

VIN

2.5V TO 5.5V

VOUT2 VOUT1

3547b F01

R3 R1

L2 L1

COUT2

COUT1

VIN

RUN2 RUN1

LTC3547B

VFB2

SW2 SW1

VFB1

GND

Table 1. Representative Surface Mount Inductors

MANU-

FACTURER PART NUMBER VALUE

MAX DC

CURRENT DCR HEIGHT

Taiyo Yuden CB2016T2R2M

CB2012T2R2M

CB2016T3R3M

2.2μH

3.3μH

510mA

530mA

410mA

0.13Ω

0.33Ω

0.27Ω

1.6mm

1.25mm

1.6mm

Panasonic ELT5KT4R7M 4.7μH 950mA 0.2Ω 1.2mm

Sumida CDRH2D18/LD 4.7μH 630mA 0.086Ω 2mm

Murata

LQH32CN4R7M23

4.7μH 450mA 0.2Ω 2mm

Taiyo Yuden NR30102R2M

NR30104R7M

2.2μH

4.7μH

1100mA

750mA

0.1Ω

0.19Ω

1mm

FDK FDKMIPF2520D

FDKMIPF2520D

4.7μH

3.3μH

2.2μH

1100mA

1200mA

1300mA

0.11Ω

0.1Ω

0.08Ω

1mm

TDK VLF3010AT4R7-

MR70

VLF3010AT3R3-

MR87

VLF3010AT2R2-

M1RD

4.7μH

3.3μH

2.2μH

700mA

870mA

1000mA

0.24Ω

0.17Ω

0.12Ω

1mm

LTC3547B

3547bfb

APPLICATIO S I FOR ATIO

WUUU

Input Capacitor (CIN) Selection

In continuous mode, the input current of the converter

is a square wave with a duty cycle of approximately

VOUT/VIN. To prevent large voltage transients, a low equiv-

alent series resistance (ESR) input capacitor sized for

the maximum RMS current must be used. The max-

imum RMS capacitor current is given by:

IRMS ≈IMAX

VOUT(V

IN −VOUT)

VIN

(2)

Where the maximum average output current IMAX equals

the peak current minus half the peak-to-peak ripple cur-

rent, IMAX = ILIM – ΔIL/2.

This formula has a maximum at VIN = 2VOUT

, where IRMS

= IOUT/2. This simple worst-case is commonly used to

design because even signiﬁ cant deviations do not offer

much relief. Note that capacitor manufacturer’s ripple cur-

rent ratings are often based on only 2000 hours lifetime.

This makes it advisable to further derate the capacitor,

or choose a capacitor rated at a higher temperature than

required. Several capacitors may also be paralleled to meet

the size or height requirements of the design. An addi-

tional 0.1μF to 1μF ceramic capacitor is also recommended

on VIN for high frequency decoupling when not using an

all-ceramic capacitor solution.

Output Capacitor (COUT) Selection

The selection of COUT is driven by the required effective

series resistance (ESR). Typically, once the ESR require-

ment for COUT has been met, the RMS current rating

generally far exceeds the IRIPPLE(P-P) requirement. The

output ripple ΔVOUT is determined by:

VOUT ILESR +1

8fCOUT









(3)

where f = operating frequency, COUT = output capacitance

and ΔIL = ripple current in the inductor. For a ﬁ xed output

voltage, the output ripple is highest at maximum input

voltage since ΔIL increases with input voltage.

If tantalum capacitors are used, it is critical that the capaci-

tors are surge tested for use in switching power supplies.

An excellent choice is the AVX TPS series of surface mount

tantalum. These are specially constructed and tested for low

ESR so they give the lowest ESR for a given volume. Other

capacitor types include Sanyo POSCAP, Kemet T510 and

T495 series, and Sprague 593D and 595D series. Consult

the manufacturer for other speciﬁ c recommendations.

Using Ceramic Input and Output Capacitors

Higher values, lower cost ceramic capacitors are now

becoming available in smaller case sizes. Their high ripple

current, high voltage rating and low ESR make them

ideal for switching regulator applications. Because the

LTC3547B control loop does not depend on the output

capacitor’s ESR for stable operation, ceramic capacitors

can be used freely to achieve very low output ripple and

small circuit size.

However, care must be taken when ceramic capacitors are

used at the input. When a ceramic capacitor is used at the

input and the power is supplied by a wall adapter through

long wires, a load step at the output can induce ringing at

the input, VIN. At best, this ringing can couple to the output

and be mistaken as loop instability. At worst, a sudden

inrush of current through the long wires can potentially

cause a voltage spike at VIN, large enough to damage the

part. For more information, see Application Note 88.

When choosing the input and output ceramic capacitors,

choose the X5R or X7R dielectric formulations. These

dielectrics have the best temperature and voltage charac-

teristics of all the ceramics for a given value and size.

LTC3547B

3547bfb

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Setting the Output Voltage

The LTC3547B regulates the VFB1 and VFB2 pins to 0.6V

during regulation. Thus, the output voltage is set by a

resistive divider according to the following formula:

VOUT =0.6V 1+R2









(4)

Keeping the current small (<5μA) in these resistors maxi-

mizes efﬁ ciency, but making it too small may allow stray

capacitance to cause noise problems or reduce the phase

margin of the error amp loop.

To improve the frequency response of the main control

loop, a feedback capacitor (CF) may also be used. Great

care should be taken to route the VFB line away from noise

sources, such as the inductor or the SW line.

Fixed output versions of the LTC3547B (e.g. LTC3547B-1)

include an internal resistive divider, eliminating the need

for external resistors. The resistor divider is chosen such

that the VFB input current is 3μA. For these versions, the

VFB pin should be connected directly to VOUT

. Table 2 lists

the ﬁ xed output voltages available for the LTC3547B.

Table 2. Fixed Output Voltage Versions

Part Number VOUT1 VOUT2

LTC3547B Adjustable Adjustable

LTC3547B-1 1.8V 1.2V

Checking Transient Response

The regulator loop response can be checked by looking

at the load transient response. Switching regulators take

several cycles to respond to a step in load current. When

a load step occurs, VOUT immediately shifts by an amount

equal to ΔILOAD • ESR, where ESR is the effective series

resistance of COUT

. ΔILOAD also begins to charge or dis-

charge COUT generating a feedback error signal used by the

regulator to return VOUT to its steady-state value. During

this recovery time, VOUT can be monitored for overshoot

or ringing that would indicate a stability problem.

The initial output voltage step may not be within the band-

width of the feedback loop, so the standard second-order

overshoot/DC ratio cannot be used to determine the phase

margin. In addition, feedback capacitors (CF1 and CF2)

can be added to improve the high frequency response, as

shown in Figure 1. Capacitor CF provides phase lead by

creating a high frequency zero with R2 which improves

the phase margin.

The output voltage settling behavior is related to the stability

of the closed-loop system and will demonstrate the actual

overall supply performance. For a detailed explanation of

optimizing the compensation components, including a re-

view of control loop theory, refer to Application Note 76.

In some applications, a more severe transient can be

caused by switching in loads with large (>1μF) input ca-

pacitors. The discharged input capacitors are effectively

put in parallel with COUT

, causing a rapid drop in VOUT

No regulator can deliver enough current to prevent this

problem if the switch connecting the load has low resistance

and is driven quickly. The solution is to limit the turn-on

speed of the load switch driver. A Hot Swap™ controller

is designed speciﬁ cally for this purpose and usually in-

corporates current limiting, short-circuit protection, and

soft-starting.

Hot Swap is a trademark of Linear Technology Corporation.

LTC3547B

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Efﬁ ciency Considerations

The percent efﬁ ciency of a switching regulator is equal to

the output power divided by the input power times 100%.

It is often useful to analyze individual losses to determine

what is limiting the efﬁ ciency and which change would

produce the most improvement. Percent efﬁ ciency can

be expressed as:

% Efﬁ ciency = 100% – (L1 + L2 + L3 + ...)

where L1, L2, etc., are the individual losses as a percent-

age of input power.

Although all dissipative elements in the circuit produce

losses, four sources usually account for the losses in

LTC3547B circuits: 1) VIN quiescent current, 2) switching

losses, 3) I2R losses, 4) other system losses.

1) The VIN current is the DC supply current given in the

Electrical Characteristics which excludes MOSFET

driver and control currents. VIN current results in a

small (<0.1%) loss that increases with VIN, even at

no load.

2) The switching current is the sum of the MOSFET driver

and control currents. The MOSFET driver current re-

sults from switching the gate capacitance of the power

MOSFETs. Each time a MOSFET gate is switched from

low to high to low again, a packet of charge dQ moves

from VIN to ground. The resulting dQ/dt is a current out

of VIN that is typically much larger than the DC bias cur-

rent. In continuous mode, IGATECHG = fO(QT + QB), where

QT and QB are the gate charges of the internal top and

bottom MOSFET switches. The gate charge losses are

proportional to VIN and thus their effects will be more

pronounced at higher supply voltages.

3) I2R losses are calculated from the DC resistances

of the internal switches, RSW

, and external inductor,

RL. In continuous mode, the average output current

ﬂ ows through inductor L, but is “chopped” between

the internal top and bottom switches. Thus, the series

resistance looking into the SW pin is a function of both

top and bottom MOSFET RDS(ON) and the duty cycle

(DC) as follows:

RSW = (RDS(ON)TOP) • (DC) + (RDS(ON)BOT) • (1– DC)

(5)

The RDS(ON) for both the top and bottom MOSFETs can

be obtained from the Typical Performance Character-

istics curves. Thus, to obtain I2R losses:

2R losses = IOUT2 • (RSW + RL

)

4) Other “hidden” losses, such as copper trace and in-

ternal battery resistances, can account for additional

efﬁ ciency degradations in portable systems. It is very

important to include these “system” level losses in

the design of a system. The internal battery and fuse

resistance losses can be minimized by making sure that

CIN has adequate charge storage and very low ESR at

the switching frequency. Other losses, including diode

conduction losses during dead-time, and inductor

core losses, generally account for less than 2% total

additional loss.

LTC3547B

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Thermal Considerations

In a majority of applications, the LTC3547B does not dis-

sipate much heat due to its high efﬁ ciency. In the unlikely

event that the junction temperature somehow reaches

approximately 150°C, both power switches will be turned

off and the SW node will become high impedance.

The goal of the following thermal analysis is to determine

whether the power dissipated causes enough temperature

rise to exceed the maximum junction temperature (125°C)

of the part. The temperature rise is given by:

RISE = PD • θJA (6)

Where PD is the power dissipated by the regulator and

θJA is the thermal resistance from the junction of the die

to the ambient temperature.

The junction temperature, TJ, is given by:

J = TRISE + TAMBIENT (7)

As a worst-case example, consider the case when the

LTC3547B is in dropout on both channels at an input

voltage of 2.7V with a load current of 300mA and an ambi-

ent temperature of 70°C. From the Typical Performance

Characteristics graph of Switch Resistance, the RDS(ON)

of the main switch is 0.9Ω. Therefore, power dissipated

by each channel is:

D = IOUT2 • RDS(ON) = 81mV

Given that the thermal resistance of a properly soldered

DFN package is approximately 76°C/W, the junction

temperature of an LTC3547B device operating in a 70°C

ambient temperature is approximately:

J = (2 • 0.081W • 76°C/W) + 70°C = 82.3°C

which is well below the absolute maximum junction tem-

perature of 125°C.

PC Board Layout Considerations

When laying out the printed circuit board, the following

checklist should be used to ensure proper operation of

the LTC3547B. These items are also illustrated graphically

in the layout diagrams of Figures 2 and 3. Check the fol-

lowing in your layout:

1. Does the capacitor CIN connect to the power VIN (Pin 3)

and GND (Pin 5) as closely as possible? This capacitor

provides the AC current of the internal power MOSFETs

and their drivers.

2. Are the respective COUT and L closely connected?

The (–) plate of COUT returns current to GND and the

(–) plate of CIN.

3. The resistor divider, R1 and R2, must be connected

between the (+) plate of COUT1 and a ground sense line

terminated near GND (Pin 5). The feedback signals VFB1

and VFB2 should be routed away from noisy components

and traces, such as the SW lines (Pins 4 and 6), and

their trace length should be minimized.

4. Keep sensitive components away from the SW pins if

possible. The input capacitor CIN and the resistors R1,

R2, R3 and R4 should be routed away from the SW

traces and the inductors.

5. A ground plane is preferred, but if not available, keep

the signal and power grounds segregated with small

signal components returning to the GND pin at a single

point. These ground traces should not share the high

current path of CIN or COUT

6. Flood all unused areas on all layers with copper. Flood-

ing with copper will reduce the temperature rise of

power components. These copper areas should be

connected to VIN or GND.

LTC3547B

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Figure 3. LTC3547B Suggested Layout

Figure 2. LTC3547B Layout Diagram (See Board Layout Checklist)

VIN

RUN2 RUN1

LTC3547B

VFB2

SW2 SW1

VFB1

CF2 CF1

GND

VIN

2.5V TO 5.5V

VOUT2 VOUT1

3547b F02

R3 R1

L2 L1

COUT2

COUT1

BOLD LINES INDICATE HIGH CURRENT PATHS

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CF2

VIA TO VIN

CF2

VIA TO VOUT2

VOUT2

COUT2

CIN

COUT1

VOUT1

VFB1 VFB2

GND

VIA TO GND

3547b F03

SW2

SW1

VIA TO VOUT1

VIA TO GND

VIN

LTC3547B

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OUTPUT CURRENT (mA)

EFFICIENCY (%)

100

0.1 10 100 1000

VIN = 2.7V

VIN = 3.6V

VIN = 4.2V

VOUT = 1.8V

OUTPUT CURRENT (mA)

EFFICIENCY (%)

100

0.1 10 100 1000

3547b F04b

VIN = 2.7V

VIN = 3.6V

VIN = 4.2V

VOUT = 2.5V

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Design Example

As a design example, consider using the LTC3547B in

a portable application with a Li-Ion battery. The battery

provides a VIN ranging from 2.8V to 4.2V. The load on each

channel requires a maximum of 300mA in active mode

and 2mA in standby mode. The output voltages are VOUT1

= 2.5V and VOUT2 = 1.8V.

Start with channel 1. First, calculate the inductor value

for about 40% ripple current (120mA in this example) at

maximum VIN. Using a derivation of Equation 1:

L1=2.5V

2.25MHz •(120mA)•12.5V

4.2V









=3.75μH

For the inductor, use the closest standard value of 4.7μH.

A 4.7μF capacitor should be more than sufﬁ cient for this

output capacitor. As for the input capacitor, a typical value

of CIN = 4.7μF should sufﬁ ce, as the source impedance of

a Li-Ion battery is very low.

Figure 4a. Design Example Circuit

Figure 4b. Efﬁ ciency vs Output Current

VIN

RUN2 RUN1

LTC3547B

VFB2

SW2 SW1

VFB1

CF2, 10pF CF1, 10pF

GND

VIN

2.5V TO 5.5V

VOUT2

1.8V AT 300mA

VOUT1

2.5V AT 300mA

3547b F04a

280k

562k

4.7μH

887k

COUT2

4.7μF

COUT1

4.7μF

C1, C2, C3: TAIYO YUDEN JMK316BJ475ML L1, L2: MURATA LQH32CN4R7M33

The feedback resistors program the output voltage. To

maintain high efﬁ ciency at light loads, the current in these

resistors should be kept small. Choosing 2μA with the

0.6V feedback voltage makes R1~300k. A close standard

1% resistor is 280k, using Equation 4.

R2 =VOUT

0.61









•R1=887k

An optional 10pF feedback capacitor (CF1) may be used

to improve transient response.

Using the same analysis for channel 2 (VOUT2 = 1.8V),

the results are:

L2 = 3.81μH

R3 = 280k

R4 = 560k

Figure 4 shows the complete schematic for this example,

along with the efﬁ ciency curve and transient response.

LTC3547B

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Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

Figure 4c. Transient Response

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PACKAGE DESCRIPTIO

DDB Package

8-Lead Plastic DFN (3mm × 2mm)

(Reference LTC DWG # 05-08-1702 Rev B)

2.00 ±0.10

(2 SIDES)

NOTE:

1. DRAWING CONFORMS TO VERSION (WECD-1) IN JEDEC PACKAGE OUTLINE M0-229

2. DRAWING NOT TO SCALE

3. ALL DIMENSIONS ARE IN MILLIMETERS

4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE

MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE

5. EXPOSED PAD SHALL BE SOLDER PLATED

6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE

0.40 ± 0.10

BOTTOM VIEW—EXPOSED PAD

0.56 ± 0.05

(2 SIDES)

0.75 ±0.05

R = 0.115

TYP

R = 0.05

TYP

2.15 ±0.05

(2 SIDES)

3.00 ±0.10

(2 SIDES)

PIN 1 BAR

TOP MARK

(SEE NOTE 6)

0.200 REF

0 – 0.05

(DDB8) DFN 0905 REV B

0.25 ± 0.05

2.20 ±0.05

(2 SIDES)

RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS

0.61 ±0.05

(2 SIDES)

1.15 ±0.05

0.70 ±0.05

2.55 ±0.05

PACKAGE

OUTLINE

0.25 ± 0.05

0.50 BSC

PIN 1

R = 0.20 OR

0.25 × 45°

CHAMFER

0.50 BSC

10μs/DIV

200mA/DIV

ILOAD

200mA/DIV

VOUT,

AC-COUPLED

100mV/DIV

3547b F04c

VIN = 3.6V

VOUT = 2.5V

ILOAD = 20mA TO 300mA

10μs/DIV

200mA/DIV

ILOAD

200mA/DIV

VOUT,

AC-COUPLED

100mV/DIV

VIN = 3.6V

VOUT = 1.8V

ILOAD = 20mA TO 300mA

LTC3547B

3547bfb

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com

LT 0807 REV B • PRINTED IN USA

TYPICAL APPLICATIO

RELATED PARTS

VIN

RUN2 RUN1

LTC3547B

VFB2

SW2 SW1

VFB1

CF2, 10pF CF1, 10pF

GND

VIN

2.5V TO 5.5V

VOUT2

1.8V AT 300mA

VOUT1

2.5V AT 300mA

3547b TA02

280k

562k

4.7μH

887k

COUT2

4.7μF

COUT1

4.7μF

C1, COUT1, COUT2: TAIYO YUDEN JMK316BJ475ML

L1, L2: MURATA LQH32CN4R7M33

PART NUMBER DESCRIPTION COMMENTS

LTC1877 600mA (IOUT), 550kHz, Synchronous Step-Down

DC/DC Converter

95% Efﬁ ciency, VIN: 2.7V to 10V, VOUT = 0.8V, IQ = 10μA,

ISD <1μA, MS8 Package

LTC1878 600mA (IOUT), 550kHz, Synchronous Step-Down

DC/DC Converter

95% Efﬁ ciency, VIN: 2.7V to 6V, VOUT = 0.8V, IQ = 10μA, ISD <1μA,

MS8 Package

LTC1879 1.2A (IOUT), 550kHz, Synchronous Step-Down DC/DC

Converter

95% Efﬁ ciency, VIN: 2.7V to 10V, VOUT = 0.8V, IQ = 15μA,

ISD <1μA, TSSOP-16 Package

LTC3403 600mA (IOUT), 1.5MHz, Synchronous Step-Down

DC/DC Converter with Bypass Transistor

96% Efﬁ ciency, VIN: 2.5V to 5.5V, VOUT = Dynamically Adjustable,

IQ = 20μA, ISD <1μA, DFN Package

LTC3404 600mA (IOUT), 1.4MHz, Synchronous Step-Down

DC/DC Converter

95% Efﬁ ciency, VIN: 2.7V to 6V, VOUT = 0.8V, IQ = 10μA,

ISD <1μA, MS8 Package

LTC3405/LTC3405A 300mA (IOUT), 1.5MHz, Synchronous Step-Down

DC/DC Converter

96% Efﬁ ciency, VIN: 2.5V to 5.5V, VOUT = 0.8V, IQ = 20μA,

ISD <1μA, ThinSOT™ Package

LTC3406 600mA (IOUT), 1.5MHz, Synchronous Step-Down

DC/DC Converter

96% Efﬁ ciency, VIN: 2.5V to 5.5V, VOUT = 0.6V, IQ = 20μA,

ISD <1μA, ThinSOT Package

LTC3409 600mA (IOUT), 1.5MHz/2.25MHz, Synchronous

Step-Down DC/DC Converter

95% Efﬁ ciency, VIN: 1.6V to 5.5V, VOUT = 0.613V, IQ = 65μA,

DD8 Package

LTC3410 300mA (IOUT), 2.25MHz, Synchronous Step-Down

DC/DC Converter

96% Efﬁ ciency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 26μA,

ISD <1μA, SC70 Package

LTC3411 1.25A (IOUT), 4MHz, Synchronous Step-Down

DC/DC Converter

95% Efﬁ ciency, VIN: 2.5V to 5.5V, VOUT = 0.8V, IQ = 60μA,

ISD <1μA, MS Package

LTC3412 2.5A (IOUT), 4MHz, Synchronous Step-Down

DC/DC Converter

95% Efﬁ ciency, VIN: 2.5V to 5.5V, VOUT = 0.8V, IQ = 60μA,

ISD <1μA, TSSOP-16E Package

LTC3440 600mA (IOUT), 2MHz, Synchronous Buck-Boost

DC/DC Converter

95% Efﬁ ciency, VIN: 2.5V to 5.5V, VOUT = 2.5V, IQ = 25μA,

ISD <1μA, MS Package

LTC3547 Dual 300mA (IOUT), 2.25MHz, Synchronous

Step-Down DC/DC Converter

96% Efﬁ ciency, VIN: 2.5V to 5.5V, VOUT = 0.6V, IQ = 40μA,

ISD <1μA, DFN Package

ThinSOT is a trademark of Linear Technology Corporation

Dual 300mA Buck Converter with Pulse Skip Mode

VIN

RUN2 RUN1

LTC3547B-1

VFB2

SW2 SW1

VFB1

GND

VIN

2.5V TO 5.5V

VOUT2

1.2V AT 300mA

VOUT1

1.8V AT 300mA

3547b TA03

4.7μH

COUT2

4.7μF

COUT1

4.7μF

C1, COUT1, COUT2: TAIYO YUDEN JMK316BJ475ML

L1, L2: MURATA LQH32CN4R7M33

1.8V/1.2V Dual 300mA Buck Converter with Pulse Skip Mode