HCPL-7520-060 - Agilent / Hewlett-Packard

Agilent HCPL-7520

Isolated Linear Sensing IC

Data Sheet

Description

The HCPL-7520 isolated linear

current sensing IC family is

designed for current sensing in

low-power electronic motor

drives. In a typical implementa-

tion, motor current flows through

an external resistor and the

resulting analog voltage drop is

sensed by the HCPL-7520. An

output voltage is created on the

other side of the HCPL-7520

optical isolation barrier. This

single-ended output voltage is

proportional to the motor

current. Since common-mode

voltage swings of several hundred

volts in tens of nanoseconds are

common in modern switching

inverter motor drives, the HCPL-

7520 was designed to ignore very

high common-mode transient

slew rates (of at least 10 kV/µs).

The high CMR capability of the

HCPL-7520 isolation amplifier

provides the precision and

stability needed to accurately

monitor motor current in high

noise motor control environ-

ments, providing for smoother

control (less “torque ripple”) in

various types of motor control

applications.

Features

• 15 kV/µs common-mode rejection

at Vcm = 1000 V

• Compact, auto-insertable 8-pin

DIP package

• 60 ppm/°C gain drift vs.

temperature

• –0.6 mV input offset voltage

•8 µV/°C input offset voltage vs.

temperature

• 100 kHz bandwidth

• 0.06% nonlinearity, single-ended

amplifer oon

• Worldwide safety approval:

UL 1577 (3750 Vrms/1 min.) and

CSA (pending), DIN EN 60747-5-2

(Option #060 only pending)

• Advanced sigma-delta (Σ-∆)

A/D converter technology

Applications

• Low-power inverter current

sensing

• Motor phase and rail current

sensing

• Switched mode power supply

signal isolation

• General purpose low-power

current sensing and monitoring

• General purpose analog signal

isolation

The product can also be used for

general analog signal isolation

applications. For general

applications, we recommend the

HCPL-7520 (gain tolerance of

±5%). The HCPL-7520 utilizes

sigma delta (Σ-∆) analog-to-

digital converter technology to

delivery offset and gain accuracy

and stability over time and

temperature. This performance is

delivered in a compact, auto-

insert, 8-pin DIP package that

meets worldwide regulatory

safety standards. (A gull-wing

surface mount option #300 is

also available).

CAUTION: It is advised that normal static precautions be taken in handling and assembly of this

component to prevent damage and /or degradation which may be induced by ESD.

IDD1

VDD1

VIN+

VIN–

GND1

IDD2

VDD2

VOUT

VREF

GND2

–

SHIELD

Functional Diagram

Ordering Information

Specify part number followed by option number (if desired).

Example: HCPL-7520-XXX

No option = Standard DIP package, 50 per tube.

300 = Gull Wing Surface Mount Option, 50 per tube.

500 = Tape and Reel Packaging Option.

060 = DIN EN 60747-5-2 Option.

Package Outline Drawings

HCPL-7520 Standard DIP Package

9.80 ± 0.25

(0.386 ± 0.010)

1.78 (0.070) MAX.

1.19 (0.047) MAX.

A 7520

YYWW

DATE CODE

1.080 ± 0.320

(0.043 ± 0.013) 2.54 ± 0.25

(0.100 ± 0.010)

0.51 (0.020) MIN.

0.65 (0.025) MAX.

4.70 (0.185) MAX.

2.92 (0.115) MIN.

DIMENSIONS IN MILLIMETERS AND (INCHES).

5678

4321

5° TYP. 0.20 (0.008)

0.33 (0.013)

7.62 ± 0.25

(0.300 ± 0.010)

6.35 ± 0.25

(0.250 ± 0.010)

HCPL-7520 Gull Wing Surface Mount Option 300 Outline Drawing

Solder Reflow Temperature Profile

TIME (SECONDS)

TEMPERATURE (°C)

200

100

50 150100 200 250

300

SEC.

50 SEC.

SEC.

160°C

140°C

150°C

PEAK

TEMP.

245°C

PEAK

TEMP.

240°C

PEAK

TEMP.

230°C

SOLDERING

TIME

200°C

PREHEATING TIME

150°C, 90 + 30 SEC.

2.5°C ± 0.5°C/SEC.

3°C + 1°C/–0.5°C

TIGHT

TYPICAL

LOOSE

ROOM

TEMPERATURE

PREHEATING RATE 3°C + 1°C/–0.5°C/SEC.

REFLOW HEATING RATE 2.5°C ± 0.5°C/SEC.

0.635 ± 0.25

(0.025 ± 0.010)

12° NOM.

0.20 (0.008)

0.33 (0.013)

9.65 ± 0.25

(0.380 ± 0.010)

0.635 ± 0.130

(0.025 ± 0.005)

7.62 ± 0.25

(0.300 ± 0.010)

9.80 ± 0.25

(0.386 ± 0.010)

6.350 ± 0.25

(0.250 ± 0.010)

1.016 (0.040)

1.194 (0.047)

1.778 (0.070)

9.398 (0.370)

9.960 (0.390)

4.826

(0.190)TYP.

0.381 (0.015)

0.635 (0.025)

PAD LOCATION (FOR REFERENCE ONLY)

1.080 ± 0.320

(0.043 ± 0.013)

4.19

(0.165)MAX.

1.780

(0.070)

MAX.

1.19

(0.047)

MAX.

2.54

(0.100)

BSC

DIMENSIONS IN MILLIMETERS (INCHES).

TOLERANCES (UNLESS OTHERWISE SPECIFIED): xx.xx = 0.01

xx.xxx = 0.005

A 7520

YYWW

LEAD COPLANARITY

MAXIMUM: 0.102 (0.004)

Regulatory Information

The HCPL-7520 is pending

approval by the following

organizations:

DIN EN

Pending approval under DIN EN

60747-5-2 with VIORM = 891 VPEAK.

DIN EN 60747-5-2 Insulation Characteristics[1]

Description Symbol Characteristic Unit

Installation classification per DIN EN 0110-1/1997-04, Table 1

for rated mains voltage ≤ 150 Vrms I – IV

for rated mains voltage ≤ 300 Vrms I – III

for rated mains voltage ≤ 600 Vrms I – II

Climatic Classification 55/100/21

Pollution Degree (DIN EN 0110-1/1997-04) 2

Maximum Working Insulation Voltage VIORM 891 Vpeak

Input to Output Test Voltage, Method b[2]

VIORM x 1.875 = VPR, 100% production test with tm = 1 sec, partial discharge <5 pC VPR 1670 Vpeak

Input to Output Test Voltage, Method a[2]

VIORM x 1.5 = VPR, type and sample test, tm = 60 sec, partial discharge <5 pC VPR 1336 Vpeak

Highest Allowable Overvoltage (transient overvoltage tini = 10 sec) VIOTM 6000 Vpeak

Safety-limiting values – maximum values allowed in the event of a failure.

Case Temperature TS175 °C

Input Current[3] IS, INPUT 400 mA

Output Power[3] PS, OUTPUT 600 mW

Insulation Resistance at TS, VIO = 500 V RS>109Ω

Notes:

1. Insulation characteristics are guaranteed only within the safety maximum ratings which must be ensured by protective circuits within the

application. Surface Mount Classifications is Class A in accordance with CECC00802.

2. Refer to the optocoupler section of the Isolation and Control Components Designer’s Catalog, under Product Safety Regulations section, (DIN EN

60747-5-2) for a detailed description of Method a and Method b partial discharge test profiles.

3. Refer to the following figure for dependence of PS and IS on ambient temperature.

Pending approval under UL 1577,

component recognition program

up to VISO = 3750 VRMS expected

prior to product release. File

E55361.

CSA

Pending approval under CSA

Component Acceptance Notice

#5, File CA 88324 expected prior

to product release.

OUTPUT POWER – P

, INPUT CURRENT – I

– CASE TEMPERATURE – °C

200

600

400

800

50 75 100

200

150 175

(mW)

125

100

300

500

700 I

(mA)

Insulation and Safety Related Specifications

Parameter Symbol Value Unit Conditions

Minimum External Air Gap L(101) 7.4 mm Measured from input terminals to output terminals,

(clearance) shortest distance through air.

Minimum External Tracking L(102) 8.0 mm Measured from input terminals to output terminals,

(creepage) shortest distance path along body.

Minimum Internal Plastic Gap 0.5 mm Through insulation distance conductor to conductor,

(internal clearance) usually the straight line distance thickness between the

emitter and detector.

Tracking Resistance CTI >175 V DIN IEC 112 Part 1

(comparative tracking index)

Isolation Group IIIa Material Group (DIN EN 0110-1/1997-04)

Absolute Maximum Ratings

Parameter Symbol Min. Max. Units Note

Storage Temperature TS–55 125 °C

Operating Temperature TA–40 100 °C

Supply Voltage VDD1_max, VDD1_max 06 V

Steady-State Input Voltage VIN+, VIN- –2.0 VDD1 + 0.5-V

Two Second Transient Input Voltage VIN+, VIN- –6.0 VDD1 + 0.5-V

Output Voltage VOUT –0.5 VDD2 + 0.5-V

Reference Input Voltage VREF 0.0 VDD2 + 0.5-V

Reference Input Current IREF 20-mA

Lead Solder Temperature 260°C for 10 sec., 1.6 mm below seating plane

Solder Reflow Temperature Profile See Package Outline Drawings section

Recommended Operating Conditions

Parameter Symbol Min. Max. Units Note

Operating Temperature TA–40 85 °C

Supply Voltage VDD1, VDD2 4.5 5.5 V

Input Voltage (accurate and linear) VIN+, VIN- –200 200 mV

Input Voltage (functional) VIN+, VIN- –2.0 2.0 V

Reference Input Voltage VREF 4.0 VDD2 V

Electrical Specifications (DC)

Unless otherwise noted, all typicals and figures are at the nominal operation conditions of VIN+ = 0 V,

VIN- = 0 V, VREF = 4.0 V, VDD1 = VDD2 = 5.0 V and TA = 25°C; all Minimum/Maximum specifications are within the

Recommended Operating Conditions.

Test

Parameter Symbol Min. Typ. Max. Units Conditions Fig. Note

Input Offset Voltage VOS –6 –1 6 mV VIN+ = 0 V 6 1

Magnitude of Input Offset ∆Vos/∆T820µV/°C7

Change vs. Temperature

Gain G VREF/0.512 VREF/0.512 V/V -0.2 V < VIN+ 82

– 5% + 5% < 0.2 V

TA = 25°C

Magnitude of Gain Change ∆G/∆T 60 300 ppm/°C -0.2 V < VIN+ 9

vs. Temperature < 0.2 V

VOUT 200 mV Nonlinearity NL200 0.06 0.55 % -0.2 V < VIN+ 10 3,4

< 0.2 V

Magnitude of VOUT 200 mV |dNL200/dT| 0.0004 %/°C -0.2 V < VIN+ 11

Nonlinearity Change < 0.2 V

vs. Temperature

VOUT 100 mV Nonlinearity NL100 0.04 0.4 % -0.1 V < VIN+ 3,5

< 0.1 V

Input Supply Current IDD1 11.7 16 mA 1,2,3

Output Supply Current IDD2 9.9 16 mA 1,2,3

Reference Voltage Input IREF 0.26 1 mA

Current

Input Current IIN+ –0.6 5 µAV

IN+ = 0 V 4

Magnitude of Input Bias |dIIN/dT| 0.45 nA/°C

Current vs. Termperature

Coefficient

Maximum Input Voltage |VIN+|MAX 256 mV 5

before VOUT Clipping

Equivalent Input Impedance RIN 700 kΩ

VOUT Output Impedance ROUT 15 Ω

Input DC Common-Mode CMRRIN 63 dB 7

Rejection Ratio

Package Characteristics

Parameter Symbol Min. Typ. Max. Units Test Conditions Fig. Note

Input-Output Momentary VISO 3750 Vrms TA = 25°C, RH < 50% 6

Withstand Voltage

Input-Output Resistance RI-O >109ΩVI-O = 500 V

Input-Output Capacitance CI-O 1.4 pF Freq = 1 MHz

Notes:

General Note: Typical values were taken from a sample of nominal units operating at nominal conditions (VDD1 = VDD2 = 5 V, VREF = 4.0 V,

Temperature = 25°C) unless otherwise stated. Nominal plots shown from Figure 1 to 11 represented the drift of these nominal units from their

nominal operating conditions.

1. Input Offset Voltage is defined as the DC Input Voltage required to obtain an output voltage of VREF/2.

2. Gain is defined as the slope of the best-fit line of the output voltage vs. the differential input voltage (VIN+ - VIN-) over the specified input range.

Gain is derived from VREF/512 mV; e.g. VREF = 5.0, gain will be 9.77 V/V.

3. Nonlinearity is defined as half of the peak-to-peak output deviation from the best-fit gain line, expressed as a percentage of the full-scale output

voltage range.

4. NL200 is the nonlinearity specified over an input voltage range of ±200 mV.

5. NL100 is the nonlinearity specified over an input voltage range of ±100 mV.

6. In accordance with UL1577, each optocoupler is proof tested by applying an insulation test voltage ≥4500 Vrms for 1 second (leakage detection

current limit, II-O < 5 µA). This test is performed before the 100% production test for the partial discharge (method b) shown in

DIN EN 60747-5-2 Insulation Characteristic Table, if applicable.

7. CMRR is defined as the ratio of the differential signal gain (signal applied differentially between pins 2 and 3) to the common-mode gain (input

pins tied together and the signal applied to both inputs at the same time), expressed in dB.

Switching Specifications (AC)

Over recommended operating conditions unless otherwise specified.

Parameter Symbol Min. Typ. Max. Units Test Conditions Fig. Note

VIN to VOUT Signal Delay (50 – 10%) tPD10 2.2 4 µsV

IN+ = 0 mV to 200 mV step 13

VIN to VOUT Signal Delay (50 – 50%) tPD50 3.4 5 µs

VIN to VOUT Signal Delay (50 – 90%) tPD90 5.2 9.9 µs

VOUT Rise Time (10 – 90%) tR3.0 7 µs

VOUT Fall Time (10 – 90%) tF3.2 7 µs

VOUT Bandwidth (-3 dB) BW 50 100 kHz VIN+ = 200 mVpk-pk 14

VOUT Noise NOUT 31.5 mVrms VIN+ = 0 V

Common Mode Transient CMTI 10 15 kV/µsT

A = 25°C, VCM = 1000 V 15

Immunity

Figure 1. Supply current vs. supply voltage.

Figure 6. Input offset change vs. supply

voltage.

Figure 5. Output voltage vs. input voltage.Figure 4. Input current vs. input voltage.

Figure 3. Supply current vs. input voltage.Figure 2. Supply current vs. temperature.

Figure 9. Gain change vs. temperature.Figure 8. Gain change vs. supply voltage.Figure 7. Input offset change vs. temperature.

– SUPPLY CURRENT – mA

– SUPPLY VOLTAGE – V

4.7 4.9

4.5 5.55.3

5.1

DD1

DD2

– TEMPERATURE – °C

9.5

9.0

7.5

-20

11.0

20 80

7.0

10.5

-40 100

8.5

040

– SUPPLY CURRENT – mA

DD1

DD2

8.0

10.0

9.0

8.0

5.0

-0.2

12.0

0 0.2

4.0

11.0

-0.3 0.3

7.0

-0.1

– SUPPLY CURRENT – mA

0.1

6.0

10.0

– INPUT VOLTAGE – V

DD1

DD2

-0.4

-0.6

-1.2

-0.2

0.2

0 0.2

-1.4

-0.3 0.3

-0.8

-0.1

– INPUT CURRENT – µA

0.1

-1.0

-0.2

– INPUT VOLTAGE – V

2.5

2.0

0.5

-0.2

4.0

0 0.2

3.5

-0.3 0.3

1.5

-0.1

– OUTPUT VOLTAGE – V

0.1

1.0

3.0

– INPUT VOLTAGE – V

∆VOS – INPUT OFFSET CHANGE – µV

– SUPPLY VOLTAGE – V

1.0

2.5

4.7 4.9

-2.0

4.5 5.55.3

1.5

-1.5

5.1

DD1

DD2

2.0

0.5

-1.0

-0.5

∆GAIN – GAIN CHANGE – %

– SUPPLY VOLTAGE – V

0.010

0.020

4.7 4.9

-0.010

4.5 5.55.3

0.015

0.005

-0.005

5.1

DD1

DD2

– TEMPERATURE – °C

0.3

0.2

-0.2

-20

0.7

20 80

-0.3

0.6

-40 100

0.1

040

∆GAIN – GAIN CHANGE – %

-0.1

0.4

0.5

∆V

– INPUT OFFSET CHANGE – mV

– TEMPERATURE – °C

-0.5

-20

2.0

-2.0

1.5

-40 10020 40

-1.5

0.5

060

-1.0

1.0

TYPICAL

MAXIMUM

Figure 15. CMTI test circuit.Figure 14. Bandwidth.

Figure 13. Propagation delay vs. temperature.Figure 12. Propagation delay test circuit.

Figure 11. Nonlinearity vs. temperature.Figure 10. Nonlinearity vs. supply voltage.

NL – NONLINEARITY – %

– SUPPLY VOLTAGE – V

0.046

0.050

4.7 4.9

0.040

4.5 5.55.3

0.048

0.044

0.042

5.1

DD1

DD2

– TEMPERATURE – °C

0.07

-20

0.09

20 80

0.05

-40 100040

NL – NONLINEARITY – %

0.06

0.08

0.1 µF

DD2

OUT

HCPL-7520

0.1 µF

REF

DD1

0.1 µF

– TEMPERATURE – °C

-20

20 80

-40 100040

– PROPAGATION DELAY – µs

Tp5010

Tp5050

Tp5090

Trise

0.1 µF

DD2

OUT

HCPL-7520

78L05

IN OUT

0.1

µF

0.1

µF

9 V

PULSE GEN.

+–

REF

FREQUENCY – kHz

-1

-2

-5

100.0

-6

0.1 1000.0

-3

1.0 10.0

GAIN – dB

-4

Application Information

Power Supplies and Bypassing

The recommended supply

connections are shown in Figure 16.

A floating power supply (which in

many applications could be the

same supply that is used to drive

the high-side power transistor) is

regulated to 5 V using a simple

zener diode (D1); the value of

resistor R4 should be chosen to

supply sufficient current from

the existing floating supply. The

voltage from the current sensing

resistor (Rsense) is applied to the

input of the HCPL-7520 through

an RC anti-aliasing filter (R2 and

C2). Although the application circuit

is relatively simple, a few recom-

mendations should be followed

to ensure optimal performance.

The power supply for the

HCPL -7520 is most often

obtained from the same supply

used to power the power

transistor gate drive circuit. If a

dedicated supply is required, in

many cases it is possible to add

an additional winding on an existing

transformer. Otherwise, some

sort of simple isolated supply can

be used, such as a line powered

transformer or a high-frequency

DC-DC converter.

An inexpensive 78L05 three-terminal

regulator can also be used to

reduce the floating supply voltage

to 5 V. To help attenuate high-

frequency power supply noise or

ripple, a resistor or inductor can be

used in series with the input of the

regulator to form a low-pass

filter with the regulator’s input

bypass capacitor.

Figure 16. Recommended supply and sense resistor connections.

MOTOR

HV-

HV+

RSENSE

GATE DRIVE

CIRCUIT

VDD1

VIN+

VIN-

GND1

HCPL-7520

0.1 µF

0.01 µF

39 Ω

5.1 V

FLOATING

POSITIVE

SUPPLY

As shown in Figure 17, 0.1 µF

bypass capacitors (C1, C2)

should be located as close as

possible to the pins of the

HCPL-7520. The bypass

capacitors are required because

of the high-speed digital nature

of the signals inside the HCPL-7520.

A 0.01 µF bypass capacitor (C2) is

also recommended at the input due

to the switched-capacitor nature

of the input circuit. The input

bypass capacitor also forms part

of the anti-aliasing filter, which is

recommended to prevent high

frequency noise from aliasing down to

lower frequencies and interfering

with the input signal. The input filter

also performs an important

reliability function—it reduces

transient spikes from ESD events

flowing through the current

sensing resistor.

PC Board Layout

The design of the printed circuit

board (PCB) should follow good

layout practices, such as keeping

bypass capacitors close to the

supply pins, keeping output

signals away from input signals,

the use of ground and power

planes, etc. In addition, the

layout of the PCB can also affect

the isolation transient immunity

(CMTI) of the HCPL-7520,

due primarily to stray capacitive

coupling between the input and

the output circuits. To obtain

optimal CMTI performance, the

layout of the PC board should

minimize any stray coupling by

maintaining the maximum

possible distance between the

input and output sides of the

circuit and ensuring that any

ground or power plane on the

PC board does not pass directly

below or extend much wider than

the body of the HCPL-7520.

Figure 17. Recommended HCPL-7520 application circuit.

MOTOR

HV-

HV+

SENSE

FLOATING

POSITIVE

SUPPLY

GATE DRIVE

CIRCUIT

HCPL-7520

0.1 µF

0.01 µF

68 Ω

IN OUT

0.1 µF

78L05

C4 C5 C6

DD1

IN+

IN-

GND1

DD2

OUT

REF

GND2

A/D

REF

GND

µC

C6 = 150 pF

C4 = C5 = 0.1 µF

+5 V

Current Sensing Resistors

The current sensing resistor

should have low resistance (to

minimize power dissipation), low

inductance (to minimize di/dt

induced voltage spikes which

could adversely affect operation),

and reasonable tolerance (to

maintain overall circuit accuracy).

Choosing a particular value for

the resistor is usually a compromise

between minimizing power

dissipation and maximizing accuracy.

Smaller sense resistance

decreases power dissipation,

while larger sense resistance

can improve circuit accuracy by

utilizing the full input range of

the HCPL -7520.

The first step in selecting a sense

resistor is determining how much

current the resistor will be sensing.

The graph in Figure 18 shows the

RMS current in each phase of a

three-phase induction motor as a

function of average motor output

power (in horsepower, hp) and

motor drive supply voltage. The

maximum value of the sense resistor

is determined by the current

being measured and the maximum

recommended input voltage

of the isolation amplifier. The

maximum sense resistance can be

calculated by taking the maximum

recommended input voltage

and dividing by the peak current

that the sense resistor should see

during normal operation. For

example, if a motor will have a

maximum RMS current of 10 A

and can experience up to 50%

overloads during normal operation,

then the peak current is

21.1 A (=10 x 1.414 x 1.5).

Assuming a maximum input

voltage of 200 mV, the maximum

value of sense resistance in this

case would be about 10 mΩ.

The maximum average power

dissipation in the sense resistor

can also be easily calculated by

multiplying the sense resistance

times the square of the maximum

RMS current, which is about 1 W

in the previous example. If the

power dissipation in the sense

resistor is too high, the resistance

can be decreased below the

maximum value to decrease

power dissipation. The minimum

value of the sense resistor is

limited by precision and accuracy

requirements of the design. As

the resistance value is reduced,

the output voltage across the

resistor is also reduced, which

means that the offset and noise,

which are fixed, become a larger

percentage of the signal amplitude.

The selected value of the sense

resistor will fall somewhere

between the minimum and

maximum values, depending on

the particular requirements of

a specific design.

When sensing currents large

enough to cause significant

heating of the sense resistor, the

temperature coefficient (tempco)

of the resistor can introduce

nonlinearity due to the signal

dependent temperature rise of the

resistor. The effect increases as

the resistor-to-ambient thermal

resistance increases. This effect

can be minimized by reducing the

thermal resistance of the current

sensing resistor or by using a

resistor with a lower tempco.

Lowering the thermal resistance

can be accomplished by

repositioning the current sensing

resistor on the PC board, by

using larger PC board traces to

carry away more heat, or by

using a heat sink. For a two-terminal

current sensing resistor, as the

value of resistance decreases, the

resistance of the leads become a

significant percentage of the total

resistance. This has two primary

effects on resistor accuracy.

First, the effective resistance of

the sense resistor can become

dependent on factors such as

how long the leads are, how

they are bent, how far they are

inserted into the board, and how

far solder wicks up the leads

during assembly (these issues

will be discussed in more detail

shortly). Second, the leads are

typically made from a material,

such as copper, which has a

much higher tempco than the

material from which the resistive

element itself is made, resulting

in a higher tempco overall. Both

of these effects are eliminated

when a four-terminal current

sensing resistor is used. A

four-terminal resistor has two

additional terminals that are

Kelvin-connected directly across

the resistive element itself; these

two terminals are used to monitor

the voltage across the resistive

element while the other two

terminals are used to carry the

load current. Because of the

Kelvin connection, any voltage

drops across the leads carrying

the load current should have no

impact on the measured voltage.

Figure 18. Motor output horsepower vs. motor

phase current and supply voltage.

15 20 25 30

MOTOR PHASE CURRENT – A (rms)

MOTOR OUTPUT POWER – HORSEPOWER

5350

440

380

220

120

When laying out a PC board for

the current sensing resistors, a

couple of points should be kept

in mind. The Kelvin connections

to the resistor should be brought

together under the body of the

resistor and then run very close

to each other to the input of the

HCPL-7520; this minimizes the

loop area of the connection and

reduces the possibility of stray

magnetic fields from interfering

with the measured signal. If the

sense resistor is not located on the

same PC board as the HCPL-7520

circuit, a tightly twisted pair of

wires can accomplish the same

thing. Also, multiple layers of the

PC board can be used to increase

current carrying capacity.

Numerous plated-through vias

should surround each non-Kelvin

terminal of the sense resistor

to help distribute the current

between the layers of the PC

board. The PC board should use

2 or 4 oz. copper for the layers,

resulting in a current carrying

capacity in excess of 20 A.

Making the current carrying

traces on the PC board fairly

large can also improve the sense

resistor’s power dissipation

capability by acting as a heat sink.

Liberal use of vias where the load

current enters and exits the PC

board is also recommended.

Sense Resistor Connections

The recommended method for

connecting the HCPL-7520 to the

current sensing resistor is shown

in Figure 17. VIN+ (pin 2 of the

HPCL-7520) is connected to the

positive terminal of the sense

resistor, while VIN- (pin 3) is

shorted to GND1 (pin 4), with the

powersupply return path

functioning as the sense line to

the negative terminal of the

current sense resistor. This

allows a single pair of wires or

PC board traces to connect the

HCPL-7520 circuit to the sense

resistor. By referencing the input

circuit to the negative side of the

sense resistor, any load current

induced noise transients on the

resistor are seen as a common-

mode signal and will not

interfere with the current-sense

signal. This is important because

the large load currents flowing

through the motor drive, along

with the parasitic inductances

inherent in the wiring of the

circuit, can generate both noise

spikes and offsets that are relatively

large compared to the small

voltages that are being measured

across the current sensing resistor.

If the same power supply is used

both for the gate drive circuit

and for the current sensing

circuit, it is very important that

the connection from GND1 of the

HCPL-7520 to the sense resistor

be the only return path for supply

current to the gate drive power supply

in order to eliminate potential ground

loop problems. The only direct

connection between the

HCPL-7520 circuit and the gate

drive circuit should be the positive

power supply line.

FREQUENTLY ASKED QUESTIONS ABOUT THE HCPL-7520

1. THE BASICS

1.1: Why should I use the HCPL-7520 for sensing current when Hall-effect sensors are available which

don’t need an isolated supply voltage?

Available in an auto-insertable, 8-pin DIP package, the HCPL-7520 is smaller than and has better linearity,

offset vs. temperature and Common Mode Rejection (CMR) performance than most Hall-effect sensors.

Additionally, often the required input-side power supply can be derived from the same supply that powers the

gate-drive optocoupler.

2. SENSE RESISTOR AND INPUT FILTER

2.1: Where do I get 10 mΩ resistors? I have never seen one that low.

Although less common than values above 10 Ω, there are quite a few manufacturers of resistors suitable for

measuring currents up to 50 A when combined with the HCPL-7520. Example product information may be

found at Dale’s web site (http://www.vishay.com/vishay/dale) and Isotek’s web site (http://www.isotekcorp.com)

and Iwaki Musen Kenkyusho’s website (http://www.iwakimusen.co.jp) and Micron Electric’s website

(http://www.micron-e.co.jp).

2.2: Should I connect both inputs across the sense resistor instead of grounding VIN- directly to pin 4?

This is not necessary, but it will work. If you do, be sure to use an RC filter on both pin 2 (VIN+) and pin 3

(VIN-) to limit the input voltage at both pads.

2.3: Do I really need an RC filter on the input? What is it for? Are other values of R and C okay?

The input anti-aliasing filter (R=39 Ω, C=0.01 µF) shown in the typical application circuit is recommended

for filtering fast switching voltage transients from the input signal. (This helps to attenuate higher signal

frequencies which could otherwise alias with the input sampling rate and cause higher input offset voltage.)

Some issues to keep in mind using different filter resistors or capacitors are:

1. (Filter resistor:) The equivalent input resistance for HCPL-7520 is around 700 kΩ. It is therefore best to

ensure that the filter resistance is not a significant percentage of this value; otherwise the offset voltage will

be increased through the resistor divider effect. [As an example, if Rfilt = 5.5 kΩ, then VOS = (Vin * 1%)

= 2 mV for a maximum 200 mV input and VOS will vary with respect to Vin.]

2. The input bandwidth is changed as a result of this different R-C filter configuration. In fact this is one of

the main reasons for changing the input-filter R-C time constant.

3. (Filter capacitance:) The input capacitance of the HCPL-7520 is approximately 1.5 pF. For proper

operation the switching input-side sampling capacitors must be charged from a relatively fixed (low

impedance) voltage source. Therefore, if a filter capacitor is used it is best for this capacitor to be a few

orders of magnitude greater than the CINPUT (A value of at least 100 pF works well.)

2.4: How do I ensure that the HCPL-7520 is not destroyed as a result of short circuit conditions which

cause voltage drops across the sense resistor that exceed the ratings of the HCPL-7520’s inputs?

Select the sense resistor so that it will have less than 5 V drop when short circuits occur. The only other

requirement is to shut down the drive before the sense resistor is damaged or its solder joints melt. This

ensures that the input of the HCPL-7520 can not be damaged by sense resistors going open-circuit.

3. ISOLATION AND INSULATION

3.1: How many volts will the HCPL-7520 withstand?

The momentary (1 minute) withstand voltage is 3750 V rms per UL 1577 and CSA Component Acceptance

Notice #5.

4. ACCURACY

4.1: Does the gain change if the internal LED light output degrades with time?

No. The LED is used only to transmit a digital pattern. Agilent has accounted for LED degradation in the

design of the product to ensure long life.

5. MISCELLANEOUS

5.1: How does the HCPL-7520 measure negative signals with only a +5 V supply?

The inputs have a series resistor for protection against large negative inputs. Normal signals are no more

than 200 mV in amplitude. Such signals do not forward bias any junctions sufficiently to interfere with

accurate operation of the switched capacitor input circuit.

www.agilent.com/semiconductors

For product information and a complete list of

distributors, please go to our web site.

For technical assistance call:

Americas/Canada: +1 (800) 235-0312 or

(916) 788-6763

Europe: +49 (0) 6441 92460

China: 10800 650 0017

Hong Kong: (+65) 6756 2394

India, Australia, New Zealand: (+65) 6755 1939

Japan: (+81 3) 3335-8152(Domestic/Interna-

tional), or 0120-61-1280(Domestic Only)

Korea: (+65) 6755 1989

Singapore, Malaysia, Vietnam, Thailand,

Philippines, Indonesia: (+65) 6755 2044

Taiwan: (+65) 6755 1843

Data subject to change.

July 8, 2003

5988-9695EN