MIC26950
12A Hyper Speed ControlTM
Synchronous DC-DC Buck Regulator
SuperSwitcher IITM
Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
General Description
The Micrel MIC26950 is a constant-frequency, synchronous
buck regulator featuring a unique digitally modified adaptive
ON-time control architecture. The MIC26950 operates over
an input supply range of 4.5V to 26V and provides a
regulated output at up to 12A of output current. The output
voltage is adjustable down to 0.8V with a typical accuracy of
±1%, and the device operates at a switching frequency of
300kHz.
Micrel’s Hyper Speed ControlTM architecture allows for ultra-
fast transient response while reducing the output capacitance
and also makes (High VIN)/(Low VOUT) operation possible.
This digitally modified adaptive tON ripple control architecture
combines the advantages of fixed-frequency operation and
fast transient response in a single device.
The MIC26950 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce
inrush current, fold-back current limit, “hiccup” mode short-
circuit protection and thermal shutdown.
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
Features
• Hyper Speed ControlTM architecture enables
- High delta V operation (VIN = 26V and VOUT = 0.8V)
- Small output capacitance
• 4.5V to 26V input voltage
• Adjustable output from 0.8V to 5.5V (±1% accuracy)
• Any CapacitorTM Stable
- Zero-ESR to high-ESR output capacitance
• 12A output current capability
• 300kHz switching frequency
• Internal compensation
• Up to 95% efficiency
• 6ms internal soft-start
• Foldback current-limit and “hiccup” mode short-circuit
protection
• Thermal shutdown
• Supports safe start-up into a pre-biased load
• –40°C to +125°C junction temperature range
• 28-pin 5mm X 6mm MLF® package
Applications
• Distributed power systems
• Communications/networking infrastructure
• Set-top box, gateways and routers
• Printers, scanners, graphic cards and video cards
____________________________________________________________________________________________________________
Typical Application
Ef ficiency (V IN = 12V)
vs. O utput Current
60
65
70
75
80
85
90
95
100
03691215
O UTPUT CURRENT ( A)
EFFICIE NCY ( %)
5.0V
3.3V
2.5V
1.5V
1.0V
July 2011 M9999-070111-C
Micrel, Inc. MIC26950
July 2011 2 M9999-070111-C
Ordering Information
Part Number Voltage Switching Frequency Junction Temperature
Range Package Lead Finish
MIC26950YJL Adjustable 300kHz –40°C to +125°C 28-pin 5mm × 6mm MLF® Pb-Free
Pin Configur ation
28-Pin 5mm X 6mm MLF® (YJL)
Pin Description
Pin
Number Pin Name Pin Function
13,14,15,
16,17,18,
19
PVIN High-Side N-internal MOSFET Drain Connection (Input): The PVIN operating voltage range is from
4.5V to 26V. Input capacitors between PVIN and the power ground (PGND) are required.
24 EN
Enable (Input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or
floating = enable, logic low = shutdown. In the off state, supply current of the device is greatly reduced
(typically 0.8mA).
25 FB
Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated
to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
26 SGND
Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin to
the PGND Pad on the top layer.
27 VDD
VDD Bias (Input): Power to the internal reference and control sections of the MIC26950. The VDD
operating voltage range is from 4.5V to 5.5V. A 2.2µF ceramic capacitor from the VDD pin-to-PGND is
recommended for clean operation.
2, 5, 6,
7, 8, 21 PGND
Power Ground. PGND is the ground path for the MIC26950 buck converter power stage. The PGND
pin connects to the sources of low-side N-Channel internal MOSFETs, the negative terminals of input
capacitors, and the negative terminals of output capacitors. The loop for the power ground should be
as small as possible and separate from the Signal ground (SGND) loop.
22 CS
Current Sense (Input): High current output driver return. The CS pin connects directly to the switch
node. Due to the high speed switching on this pin, the CS pin should be routed away from sensitive
nodes. CS pin also senses the current by monitoring the voltage across the low-side internal
MOSFET during OFF-time.
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July 2011 3 M9999-070111-C
Pin Description (Continued)
Pin
Number Pin Name Pin Function
20 BST
Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky
diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1F is connected
between the BST pin and the SW pin.
4, 9, 10,
11, 12 SW Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET
drain.
23 VIN Power Supply Voltage (Input): Requires bypass capacitor to SGND.
1, 3, 28 NC No Connect.
Micrel, Inc. MIC26950
July 2011 4 M9999-070111-C
Absolute Maximum Ratings(1,2)
PVIN to PGND................................................ −0.3V to +28V
VIN to PGND ....................................................−0.3V to PVIN
VDD to PGND ................................................... −0.3V to +6V
VSW, VCS to PGND ..............................−0.3V to (PVIN +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND .................................................. −0.3V to 34V
VEN to PGND ......................................−0.3V to (VDD + 0.3V)
VFB to PGND....................................... −0.3V to (VDD + 0.3V)
PGND to SGND ........................................... −0.3V to +0.3V
Junction Temperature ..............................................+150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C
Operating Ratings(3)
Supply Voltage (PVIN, VIN)............................... 4.5V to 26V
Output Voltage Range (VOUT)........................... 0.8V to 5.5V
Bias Voltage (VDD)............................................ 4.5V to 5.5V
Enable Input (VEN)................................................. 0V to VDD
Junction Temperature (TJ) ........................ −40°C to +125°C
Maximum Power Dissipation......................................Note 4
Package Thermal Resistance(4)
5mm x 6mm MLF®(θJA) .....................................36°C/W
Electrical Characteristics(5)
PVIN = VIN = 12V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C TJ +125°C.
Parameter Condition Min. Typ. Max. Units
Power Supply Input
Input Voltage Range (VIN, PVIN) 4.5 26 V
VDD Bias Voltage
Operating Bias Voltage (VDD) 4.5 5 5.5 V
Undervoltage Lockout Trip Level VDD Rising 2.4 2.7 3.2 V
UVLO Hysteresis 50 mV
Quiescent Supply Current VFB = 1.5V 1.4 3 mA
Shutdown Supply Current VDD = VBST = 5.5V, VIN = 26V
SW = unconnected, VEN = 0V
0.7 2 mA
Reference
0°C TJ 85°C (±1.0%) 0.792 0.8 0.808
Feedback Reference Voltage −40°C TJ 125°C (±1.5%) 0.788 0.8 0.812 V
Load Regulation IOUT = 0A to 12A 0.2 %
Line Regulation VIN = (VOUT + 3.0V) to 26V 0.1 %
FB Bias Current VFB = 0.8V 5 nA
Enable Control
EN Logic Level High 4.5V < VDD < 5.5V 1.2 0.85 V
EN Logic Level Low 4.5V < VDD < 5.5V 0.78 0.4 V
EN Bias Current VEN = 0V 50 µA
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. See “Applications Information.”
5. Specification for packaged product only.
Micrel, Inc. MIC26950
July 2011 5 M9999-070111-C
Electrical Characteristics(5)
PVIN = VIN = 12V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C TJ +125°C.
Parameter Condition Min. Typ. Max. Units
Oscillator
Switching Frequency (6) 225 300 375 kHz
Maximum Duty Cycle (7) V
FB = 0V 87 %
Minimum Duty Cycle VFB > 0.8V 0 %
Minimum Off-Time 360 ns
Soft-Start
Soft-Start Time 6 ms
Short-Circuit Protection
Current-Limit Threshold VFB = 0.8V 13.2 27 A
Short-Circuit Current VFB = 0V 8 A
Internal FETs
Top-MOSFET RDS (ON) I
SW = 1A 17 m
Bottom-MOSFET RDS (ON) I
SW = 1A 6 m
SW Leakage Current PVIN = 26V, VSW = 26V, VEN = 0V, VBST = 31.5 V 60 µA
VIN Leakage Current PVIN = 26V, VSW = 0V, VEN = 0V, VBST = 31.5V 25 µA
Thermal Protection
Over-temperature Shutdown TJ Rising 155 °C
Over-temperature Shutdown
Hysteresis 10 °C
Notes:
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns.
Micrel, Inc. MIC26950
July 2011 6 M9999-070111-C
Typical Characteristics
V
IN
Operat ing Supply Current
vs. Input Vo ltage
0.0
2.0
4.0
6.0
8.0
10.0
4 10162228
I NPUT VO L TAGE ( V)
SUPPLY CURRENT (m A)
V
OUT
= 1.2V
V
DD
= 5V
SWITCHING
I
OUT
= 0A
V
IN
Shutdown Current
vs. Input Voltage
0
4
8
12
16
20
410162228
INPUT VOLTA G E ( V)
SHUTDOWN CURRENT ( µA)
V
DD
= 5V
V
EN
= 0V
V
DD
Operating Supply Current
vs. Input Voltage
0
4
8
12
16
20
410162228
INPUT VOLTA G E ( V)
SUPPLY CURRENT (m A)
V
OUT
= 1.2V
V
DD
= 5V
SWITCHING
Feedback Voltage
vs. Input Voltage
0.792
0.794
0.796
0.798
0.800
0.802
0.804
0.806
0.808
4 10162228
INPUT VOLTAGE (V)
FEEDBACK VOLT AG E ( V)
V
OUT
= 1.2V
V
DD
= 5V
I
OUT
= 0A
Total Regulation
vs. Input Voltage
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
4 10162228
INPUT VOLTA G E ( V)
TOTAL REGULATION (%)
V
OUT
= 1.2V
V
DD
= 5V
I
OUT
= 0A to 12A
Current Limit
vs. Input Voltage
0
5
10
15
20
25
30
4 8 12 16 20 24 28
INPUT VOLTAGE (V)
CURRENT LIMIT (A)
V
OUT
= 1.2V
V
DD
= 5V
Switching Frequency
vs. Input Voltage
210
255
300
345
390
410162228
INPUT VOLTAGE (V)
SWITCHING F REQUENCY ( k Hz )
V
OUT
= 1.2V
V
DD
= 5V
I
OUT
= 0A
Micrel, Inc. MIC26950
July 2011 7 M9999-070111-C
Typical Characteristics (Continued)
V
DD
Operating Supply C urrent
vs. Temperature
0.0
2.0
4.0
6.0
8.0
10.0
-50 -20 10 40 70 100 130
TEM P ERATURE ( °C)
SUPPLY CURRENT ( m A)
V
IN
= 12V
V
OUT
= 1.2V
V
DD
= 5V
I
OUT
= 0A
SW ITCHING
V
DD
Shut d own C urrent
vs. Temperature
0
0.2
0.4
0.6
0.8
1
-50 -20 10 40 70 100 130
TEM PE RATURE ( °C)
SUPPLY CURRENT (mA)
V
IN
= 12V
I
OUT
= 0A
V
DD
= 5V
V
EN
= 0V
VDD U VLO Threshold
vs. Temperat ure
2.3
2.4
2.5
2.6
2.7
2.8
-50 -20 10 40 70 100 130
TEM P ERATURE ( °C)
V
DD
THRESHOL D (V)
Rising
Falling
V
IN
= 12V
V
IN
Operating Supply C urrent
vs. Te m p era ture
0.0
2.0
4.0
6.0
8.0
10.0
-50 -20 10 40 70 100 130
TEMPERATURE (° C)
SUPPLY CURRENT (m A)
V
IN
= 12V
V
OUT
= 1.2V
V
DD
= 5V
I
OUT
= 0A
SW ITCHING
V
IN
Shut dow n C urrent
vs. Temperature
0.0
2.0
4.0
6.0
8.0
10.0
-50 -20 10 40 70 100 130
TEM P ERATURE ( °C)
SUPPLY CURRE NT ( µA)
V
IN
= 12V
V
DD
= 5V
I
OUT
= 0A
Current Limi t
vs. Te m p era ture
0
5
10
15
20
25
30
35
40
-50 -20 10 40 70 100 130
TEMPERA T URE (°C)
CURRENT L IMIT ( A)
V
IN
= 12V
V
OUT
= 1.2V
V
DD
= 5V
Feedback Vol tage
vs. Temperature
0.792
0.794
0.796
0.798
0.800
0.802
0.804
0.806
0.808
-50 -20 10 40 70 100 130
TEMPERA TURE (°C)
FEEDBAC K VOLTAG E ( V )
VIN = 12V
VOUT = 1.2V
VDD = 5V
IOUT = 0A
Load Regul ati on
vs. Temperature
0.0%
0.2%
0.4%
0.6%
0.8%
1.0%
-50 -20 10 40 70 100 130
TEM PERATURE (° C)
LOAD REGULATION ( % )
V
IN
= 12V
V
OUT
= 1.2V
V
DD
= 5V
I
OUT
= 0A to 12A
Line Regulation
vs. Temperat ure
0.0%
0.1%
0.2%
0.3%
0.4%
0.5%
-50 -20 10 40 70 100 130
TEM P ERATURE (°C)
LINE REGULATION (% )
V
IN
=6V to 26V
V
OUT
= 1.2V
V
DD
= 5V
Swi tching Frequency
vs. Temperature
255
270
285
300
315
330
345
-50 -20 10 40 70 100 130
TEMPERAT URE (°C)
SWITCHI NG FREQ UE NCY ( k Hz)
V
IN
= 12V
V
OUT
= 1.2V
V
DD
= 5V
I
OUT
= 0A
EN Bias Current
vs. Temperature
0
20
40
60
80
100
-50 -20 10 40 70 100 130
TEM P ERATURE ( °C)
EN BIAS CURRENT ( µA)
V
IN
= 12V
V
OUT
= 1.2V
V
DD
= 5V
Micrel, Inc. MIC26950
July 2011 8 M9999-070111-C
Typical Characteristics (Continued)
Efficiency
vs. Out put Current
60
65
70
75
80
85
90
95
036912
O UTPUT CURRENT (A)
EFFI CIENCY (% )
V
OUT
= 1.2V
V
DD
= 5V
24V
IN
12V
IN
6V
IN
18V
IN
Feedback Vol tage
vs. Out p ut Current
0.792
0.794
0.796
0.798
0.800
0.802
0.804
0.806
0.808
036912
O UT PUT CU RRE NT (A)
FEEDBACK VO LT AGE (V)
V
IN
= 12V
V
OUT
= 1.2V
V
DD
= 5V
Feedback Vol tage (% )
vs. Out p ut Current
-0.5%
-0.4%
-0.3%
-0.2%
-0.1%
0.0%
0.1%
036912
O UTPUT CURRENT (A)
FEEDBACK VO LTAGE (% )
V
IN
= 12V
V
OUT
= 1.2V
V
DD
= 5V
Li ne Regulati o n
vs. Output Current
0.0%
0.1%
0.2%
0.3%
0.4%
0.5%
036912
O UTP UT CURRENT (A)
LI NE REGULATION (% )
V
IN
= 6V to 26V
V
OUT
= 1.2V
V
DD
= 5V
Swi tching Frequency
vs. O utput Current
210
240
270
300
330
360
390
036912
O UT PUT CURRE NT (A)
SWITCHI NG FREQ UENCY (kHz)
V
IN
= 12V
V
OUT
= 1.2V
V
DD
= 5V
Output Voltage (V
IN
= 5V)
vs. O utput C urrent
3.4
3.6
3.8
4
4.2
4.4
4.6
4.8
5
0 3 6 9 12 15
O UTPUT CURRENT (A)
OUTPUT VOLTAGE (V)
V
IN
= 5V
V
FB
< 0.8V
V
DD
= 5V
T
A
25ºC
85ºC
125ºC
Die Temperature* (V
IN
= 5V)
vs. O u tput C urrent
0
20
40
60
80
100
036912
O UTPUT CURRENT ( A)
DI E TEMPERATURE (° C)
V
IN
= 5V
V
OUT
= 1.2V
V
DD
= 5V
Di e Temperature* (V
IN
= 12V)
vs. O utput C urrent
0
20
40
60
80
100
036912
O UTP UT CURRENT (A)
DI E TEMPERATURE ( ° C)
V
IN
= 12V
V
OUT
= 1.2V
V
DD
= 5V
Di e Temperature* (V
IN
= 24V)
vs. O utput C urren t
0
20
40
60
80
100
036912
O UTP UT CURR ENT (A)
DI E T EMPERATURE ( °C)
V
IN
= 24V
V
OUT
= 1.2V
V
DD
= 5V
Ef ficiency (V
IN
= V
DD
= 5V)
vs. Output Current
70
75
80
85
90
95
100
03691215
O UTPUT CURRENT (A)
EFFI CIENCY (%)
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
Efficiency (V
IN
= 12V)
vs. O utput Current
70
75
80
85
90
95
100
0 3 6 9 12 15
O UTP UT CURREN T ( A)
EFFICIENCY (% )
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
Efficiency (V
IN
= 24)
vs. O utput Current
70
75
80
85
90
95
03691215
O UT PUT CURRENT (A)
EFFICIENCY (%)
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
Micrel, Inc. MIC26950
July 2011 9 M9999-070111-C
Th erma l Derating*
vs. Ambient Temperature
0
4
8
12
16
20
-50-250 255075100125
A MBIE NT TE MPERAT URE (°C)
OUT PUT CURRENT ( A)
V
IN
= 5V
V
OUT
= 0.8, 1.2, 1.5V
1.5V
0.8V
Thermal Derating*
vs. Ambi en t Temperat ure
0
4
8
12
16
20
-50 -25 0 25 50 75 100 125
A MBIENT TEMPERATURE (°C)
OUT PUT CURRENT (A)
V
IN
= 5V
V
OUT
= 1.8, 2.5, 3.3V
3.3V
1.8V
Thermal Derati ng*
vs. Ambient Temperatu re
0
5
10
15
20
25
-50 -25 0 25 50 75 100 125
A M BI ENT TEMPERATURE ( ° C)
OUT PUT CURRENT (A)
V
IN
= 12V
V
OUT
= 0.8, 1.2, 1.8V
1.8V
0.8V
Thermal Derati ng*
vs. Ambient Temperatu re
0
4
8
12
16
20
-50 -25 0 25 50 75 100 125
A M BIENT TEMPERATURE (°C)
OUT PUT CURRENT ( A)
V
IN
= 12V
V
OUT
= 2.5, 3.3, 5V
5V
2.5V
Th erma l Der ating*
vs. Ambi ent Temperat ure
0
4
8
12
16
20
-50-250 255075100125
A M BIENT TEMPERATURE ( °C)
O UT PUT CUR RENT (A)
V
IN
= 24V
V
OUT
= 0.8, 1.2, 2.5V
2.5V
0.8V
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26950 case mounted on a 5 square inch 4 layer, 0.62”,
FR-4 PCB with 2oz finish copper weight per layer, see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient
temperature and proximity to other heat emitting components.
Micrel, Inc. MIC26950
July 2011 10 M9999-070111-C
Functional Characteristics
Micrel, Inc. MIC26950
July 2011 11 M9999-070111-C
Functional Characteristics (Continued)
Micrel, Inc. MIC26950
July 2011 12 M9999-070111-C
Functional Characteristics (Continued)
Micrel, Inc. MIC26950
July 2011 13 M9999-070111-C
Functional Diagram
Figure 1. MIC26950 Block Diagram
Micrel, Inc. MIC26950
July 2011 14 M9999-070111-C
Functional Description
The MIC26950 is an adaptive ON-time synchronous
step-down DC-DC regulator. It is designed to operate
over a wide input voltage range from 4.5V to 26V and
provides a regulated output voltage at up to 12A of
output current. A digitally modified adaptive ON-time
control scheme is employed in to obtain a constant
switching frequency and to simplify the control
compensation. Over-current protection is implemented
without the use of an external sense resistor. The device
includes an internal soft-start function which reduces the
power supply input surge current at start-up by
controlling the output voltage rise time.
Theory of Operation
Figure 1 illustrates the block diagram for the control loop
of the MIC26950. The output voltage is sensed by the
MIC26950 feedback pin FB via the voltage divider R1
and R2, and compared to a 0.8V reference voltage VREF
at the error comparator through a low gain
transconductance (gm) amplifier. If the feedback voltage
decreases and the output of the gm amplifier is below
0.8V, then the error comparator will trigger the control
logic and generate an ON-time period. The ON-time
period length is predetermined by the “FIXED tON
ESTIMATION” circuitry:
300kHzV
V
t
IN
OUT
ed)ON(estimat ×
= (1)
where VOUT is the output voltage and VIN is the power
stage input voltage.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
360ns, the MIC26950 control logic will apply the tOFF(min)
instead. tOFF(min) is required to maintain enough energy in
the boost capacitor (CBST) to drive the high-side
MOSFET. The maximum duty cycle is obtained from the
360ns tOFF(min):
SS
OFF(min)S
max t
360ns
1
t
tt
D−=
−
= (2)
where tS = 1/300kHz = 3.33μs.
It is not recommended to use MIC26950 with a OFF-time
close to tOFF(min) during steady-state operation. Also, as
VOUT increases, the internal ripple injection will increase
and reduce the line regulation performance. Therefore,
the maximum output voltage of the MIC26950 should be
limited to 5.5V. Please refer to “Setting Output Voltage”
subsection in “Application Information” for more details.
The actual ON-time and resulting switching frequency
will vary with the part-to-part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications, such as 26V to 1.0V. The minimum tON
measured on the MIC26950 evaluation board is about
184ns. During load transients, the switching frequency is
changed due to the varying OFF-time.
To illustrate the control loop operation, will be analyzed
both the steady-state and load transient scenarios. For
easy analysis, the gain of the gm amplifier is assumed to
be 1. With this assumption, the inverting input of the
error comparator is the same as the feedback voltage.
Figure 2 shows the MIC26950 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON-time period. The ON-
time is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
Figure 2. MIC26950 Control Loop Timing
Micrel, Inc. MIC26950
July 2011 15 M9999-070111-C
Figure 3 shows the operation of the MIC26950 during a
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC26950 converter.
Figure 3. MIC26950 Load Transien t Response
Unlike true current-mode control, the MIC26950 uses the
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC26950 control loop has the advantage
of eliminating the need for slope compensation.
In order to meet the stability requirements, the
MIC26950 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV. If a low ESR output capacitor is selected,
the feedback voltage ripple may be too small to be
sensed by the gm amplifier and the error comparator.
Also, the output voltage ripple and the feedback voltage
ripple are not necessarily in phase with the inductor
current ripple if the ESR of the output capacitor is very
low. In these cases, ripple injection is required to ensure
proper operation. Please refer to “Ripple Injection”
subsection in “Application Information” for more details
about the ripple injection technique.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC26950 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 6ms with a 9.7mV step. Therefore, the
output voltage is controlled to increase slowly by a stair-
case VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function behavior
correctly.
Current Limit
The MIC26950 uses the RDS(ON) of the internal low-side
power MOSFET to sense over-current conditions. This
method will avoid adding cost, board space and power
losses taken by a discrete current sense resistor. The
low-side MOSFET is used because it displays much
lower parasitic oscillations during switching than the
high-side MOSFET.
In each switching cycle of the MIC26950 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. If the inductor current is
greater than 27A, then the MIC26950 turns off the high-
side MOSFET and a soft-start sequence is triggered.
This mode of operation is called “hiccup mode” and its
purpose is to protect the downstream load in case of a
hard short. The current limit threshold has a fold back
characteristic related to the feedback voltage. As shown
in Figure 4.
Current Limit Threshold
vs. Feedback Voltage
0.0
5.0
10.0
15.0
20.0
25.0
30.0
0.0 0.2 0.4 0.6 0.8 1.0
FEEDBACK VOLTAGE (V)
CURRENT LIMIT THRESHOLD (A)
Figure 4. MIC26950 Current Li mitin g Circuit
Micrel, Inc. MIC26950
July 2011 16 M9999-070111-C
MOSFET Gate Drive
The Block Diagram of Figure 1 shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended) and
CBST. This circuit supplies energy to the high-side drive
circuit. Capacitor CBST is charged, while the low-side
MOSFET is on, and the voltage on the SW pin is
approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the high-side MOSFET turns on, the voltage on
the SW pin increases to approximately VIN. Diode D1 is
reversed biased and CBST floats high while continuing to
keep the high-side MOSFET on. The bias current of the
high-side driver is less than 10mA so a 0.1F to 1F is
sufficient to hold the gate voltage with minimal droop for
the power stroke (high-side switching) cycle, i.e. BST =
10mA x 3.33s/0.1F = 333mV.
When the low-side MOSFET is turned back on, CBST is
recharged through D1. A small resistor RG, which is in
series with CBST, can be used to slow down the turn-on
time of the high-side N-channel MOSFET.
The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD – VDIODE, where VDIODE is the voltage drop across
D1. An approximate 30ns delay between the high-side
and low-side driver transitions is used to prevent current
from simultaneously flowing unimpeded through both
MOSFETs.
Micrel, Inc. MIC26950
July 2011 17 M9999-070111-C
Application Information
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by the Equation 4:
OUT(max)swIN(max)
OUTIN(max)OUT
I20% f V
)V(VV
L×××
−×
= (4)
where:
fSW = switching frequency, 300kHz
20% = ratio of AC ripple current to DC output current
VIN(max) = maximum power stage input voltage
The peak-to-peak inductor current ripple is:
L f V
)V(VV
I
swIN(max)
OUTIN(max)OUT
L(pp) ××
−×
=Δ (5)
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
IL(pk) =IOUT(max) + 0.5 × IL(pp) (6)
The RMS inductor current is used to calculate the I2R
losses in the inductor.
12
I
II
2
L(PP)
2
OUT(max)L(RMS) += (7)
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC26950 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor. Copper loss in the
inductor is calculated by Equation 8:
PINDUCTOR(Cu) = IL(RMS)
2 × RWINDING (8)
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
PWINDING(Ht) = RWINDING(20°C) ×
1 + 0.0042 × (TH – T20°C)) (9)
where:
TH = temperature of wire under full load
T20°C = ambient temperature
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
The type of the output capacitor is usually determined by
its ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are tantalum, low-ESR aluminum electrolytic, OS-
CON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view.
Micrel, Inc. MIC26950
July 2011 18 M9999-070111-C
The maximum value of ESR is calculated:
L(PP)
OUT(pp)
CI
V
ESR OUT ≤ (10)
where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
IL(PP) = peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 11:
()
2
CL(PP)
2
SWOUT
L(PP)
OUT(pp) OUT
ESRI
8fC
I
V×+
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
××
=
(11)
where:
D = duty cycle
COUT = output capacitance value
fSW = switching frequency
As described in the “Theory of Operation” subsection in
“Functional Description”, the MIC26950 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 12:
12
I
IL(PP)
(RMS)COUT = (12)
The power dissipated in the output capacitor is:
OUTOUTOUT C
2
(RMS)C)DISS(C ESRIP ×= (13)
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend upon the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
VIN = IL(pk) × ESRCIN (14)
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
D)(1DII OUT(max)CIN(RMS) −××≈ (15)
The power dissipated in the input capacitor is:
PDISS(CIN) = ICIN(RMS)
2 × ESRCIN (16)
Ripple Injection
The VFB ripple required for proper operation of the
MIC26950 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
the MIC26950 will lose control and the output voltage is
not regulated. In order to have some amount of VFB
ripple, a ripple injection method is applied for low output
voltage ripple applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1) Enough ripple at the feedback voltage due to the large
ESR of the output capacitors.
Micrel, Inc. MIC26950
July 2011 19 M9999-070111-C
As shown in Figure 5a, the converter is stable without
any ripple injection. The feedback voltage ripple is:
(pp)
LCFB(pp) IESR
R2R1
R2
VOUT ××
+
= (17)
where IL(pp) is the peak-to-peak value of the inductor
current ripple.
2) Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 5b. The typical Cff value is between 1nF and
100nF. With the feedforward capacitor, the feedback
voltage ripple is very close to the output voltage ripple:
(pp)
LFB(pp) IESRV×≈ (18)
3) Virtually no ripple at the FB pin voltage due to the very
low ESR of the output capacitors.
Figure 5a. Enough Ripple at FB
Figure 5b. Inadequate Ripple at FB
Figure 5c. Invisible Ripple at FB
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 5c. The injected ripple
is:
τ
×
××××=
SW
divINFB(pp) f
1
D)-(1DKVV (19)
R1//R2R
R1//R2
K
inj
div +
= (20)
where
VIN = Power stage input voltage
D = duty cycle
fSW = switching frequency
τ = (R1//R2//Rinj) × Cff
In equations (19) and (20), it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
f
1
SW
<<=
×
ττ
(21)
If the voltage divider resistors R1 and R2 are in the k
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in k range.
Micrel, Inc. MIC26950
July 2011 20 M9999-070111-C
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 22:
In addition to the external ripple injection added at the
FB pin, internal ripple injection is added at the inverting
input of the comparator inside the MIC26950, as shown
in Figure 7. The inverting input voltage VINJ is clamped to
1.2V. As VOUT is increased, the swing of VINJ will be
clamped. The clamped VINJ reduces the line regulation
because it is reflected back as a DC error on the FB
terminal. Therefore, the maximum output voltage of the
MIC26950 should be limited to 5.5V to avoid this line
regulation problem.
D)(1D
f
V
V
KSW
IN
FB(pp)
div −×
×
×=
τ
(22)
Then the value of Rinj is obtained as:
1)
K
1
((R1//R2)R
div
inj −×= (23)
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
Setting Output Voltage
The MIC26950 requires two resistors to set the output
voltage as shown in Figure 6.
Figure 7. Internal Ripple Injection
Thermal Measurements
Measuring the IC’s case temperature is recommended to
ensure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
Figure 6. Voltage-Divider Configuration
The output voltage is determined by the equation: Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer.
If a thermal couple wire is used, it must be constructed
of 36 gauge wire or higher (smaller wire size) to
minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC-TT-K-36-36)
is adequate for most applications.
)
R2
R1
(1VV FBOUT +×= (24)
where, VFB = 0.8V. A typical value of R1 can be between
3k and 10k. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small, it will decrease the efficiency of the power supply,
especially at light loads. Once R1 is selected, R2 can be
calculated using:
FBOUT
FB
VV
R1V
R2 −
×
= (25)
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, a IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
Micrel, Inc. MIC26950
July 2011 21 M9999-070111-C
PCB Layout Guidelines
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
The following guidelines should be followed to insure
proper operation of the MIC26950 converter.
IC
• The 2.2µF ceramic capacitor, which is connected to
the VDD terminal, must be located right at the IC. The
VDD terminal is very noise sensitive and placement of
the capacitor is very critical. Use wide traces to
connect to the VDD and PGND pins.
• The signal ground pin (SGND) must be connected
directly to the ground planes. Do not route the
SGND pin to the PGND Pad on the top layer.
• Place the IC close to the point-of-load (POL).
• Use fat traces to route the input and output power
lines.
• Signal and power grounds should be kept separate
and connected at only one location.
Input Capacitor
• Place the input capacitor next.
• Place the input capacitors on the same side of the
board and as close to the IC as possible.
• Keep both the VIN Pin and PGND connections short.
• Place several vias to the ground plane close to the
input capacitor ground terminal.
• Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
• Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
• If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
• In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.
Inductor
• Keep the inductor connection to the switch node
(SW) short.
• Do not route any digital lines underneath or close to
the inductor.
• Keep the switch node (SW) away from the feedback
(FB) pin.
• The CS pin should be connected directly to the SW
pin to accurate sense the voltage across the low-
side MOSFET.
• To minimize noise, place a ground plane underneath
the inductor.
Output Capacitor
• Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
• Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
• The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
RC Snubber
• Place the RC snubber on the same side of the board
and as close to the SW pin as possible.
Micrel, Inc. MIC26950
July 2011 22 M9999-070111-C
Evaluation Board Schematic
Figure 8. Schematic of MIC26950 Evaluation Board
(J13, R13, R15 are for testing purposes)
Micrel, Inc. MIC26950
July 2011 23 M9999-070111-C
Bill of Materials
Item Part Number Manufacturer Description Qty.
C1 B41125A7227M EPCOS(1) 220µF Aluminum Capacitor, SMD, 35V 1
12105C475KAZ2A AVX(2)
C2, C3
GRM32ER71H475KA88L Murata(3)
4.7µF Ceramic Capacitor, X7R, Size 1210, 50V 2
C4, C5 Open
06035C104KAT2A AVX(2)
GRM188R71H104KA93D Murata(3)
C6, C7,
C10
C1608X7R1H104K TDK(4)
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V 3
0805ZC225MAT2A AVX(2)
GRM21BR71A225KA01L Murata(3)
C8,C9
C2012X7R1A225K TDK(4)
2.2µF Ceramic Capacitor, X7R, Size 0805, 10V 2
06035C102KAT2A AVX(2)
GRM188R71H102KA01D Murata(3)
C11
C1608X7R1H102K TDK(4)
1nF Ceramic Capacitor, X7R, Size 0603, 50V 1
06035C223KAZ2A AVX(2)
GRM188R71H223K Murata(3)
C12
C1608X7R1H223K TDK(4)
22nF Ceramic Capacitor, X7R, Size 0603, 50V 1
12106D107MAT2A AVX(2)
C13 GRM32ER60J107ME20L Murata(3)
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V 1
C14 Open
C15 6SEPC560MX SANYO(5) 560µF OSCON Capacitor, 6.3V 1
SD103AWS-7 Diodes Inc(6)
D1
SD103AWS Vishay(7)
Small Signal Schottky Diode 1
D2 CMDZ5L6 Central Semi(8) 5.6V Zener Diode 1
L1 HCF1305-2R2-R
Cooper Bussmann(9) 2.2µH Inductor, 15A Saturation Current 1
Q1 FCX619 ZETEX(6) 50V NPN Transistor 1
R1 CRCW06034R75FKEA Vishay Dale(7) 4.75 Resistor, Size 0603, 1% 1
R2, R16 CRCW08051R21FKEA Vishay Dale(7) 1.21 Resistor, Size 0805, 1% 2
R3, R4 CRCW060310K0FKEA Vishay Dale(7) 10k Resistor, Size 0603, 1% 2
Notes:
1. EPCOS: www.epcos.com.
2. AVX: www.avx.com.
3. Murata: www.murata.com.
4. TDK: www.tdk.com.
5. SANYO: www.sanyo.com.
6. Diode Inc.: www.diodes.com.
7. Vishay: www.vishay.com.
8. Central Semi: www.centralsemi.com.
9. Cooper Bussmann: www.cooperbussmann.com.
10. Micrel, Inc.: www.micrel.com.
Micrel, Inc. MIC26950
July 2011 24 M9999-070111-C
Bill of Materials (Continued)
Item Part Number Manufacturer Description Qty.
R5 CRCW060380K6FKEA Vishay Dale(7) 80.6k Resistor, Size 0603, 1% 1
R6 CRCW060340K2FKEA Vishay Dale(7) 40.2k Resistor, Size 0603, 1% 1
R7 CRCW060320K0FKEA Vishay Dale(7) 20k Resistor, Size 0603, 1% 1
R8 CRCW060311K5FKEA Vishay Dale(7) 11.5k Resistor, Size 0603, 1% 1
R9 CRCW06038K06FKEA Vishay Dale(7) 8.06k Resistor, Size 0603, 1% 1
R10 CRCW06034K75FKEA Vishay Dale(7) 4.75k Resistor, Size 0603, 1% 1
R11 CRCW06033K24FKEA Vishay Dale(7) 3.24k Resistor, Size 0603, 1% 1
R12 CRCW06031K91FKEA Vishay Dale(7) 1.91k Resistor, Size 0603, 1% 1
R13 CRCW06030000FKEA Vishay Dale(7) 0 Resistor, Size 0603, 5% 1
R14 CRCW06035K23FKEA Vishay Dale(7) 5.23k Resistor, Size 0603, 1% 1
R15 CRCW060349R9FKEA Vishay Dale(7) 49.9 Resistor, Size 0603, 1% 1
U1 MIC26950YJL Micrel. Inc.(10) 26V/12A Synchronous Buck DC-DC Regulator 1
Micrel, Inc. MIC26950
July 2011 25 M9999-070111-C
PCB Layout
Figure 9. MIC26950 Evaluation Board Top Layer
Figure 10. MIC26950 Evaluation Board Mid-Layer 1 (Ground Plane)
Micrel, Inc. MIC26950
July 2011 26 M9999-070111-C
PCB Layout (Continued)
Figure 11. MIC26950 Evaluation Board Mid-Layer 2
Figure 12. MIC26950 Evaluation Board Bottom Layer
Micrel, Inc. MIC26950
July 2011 27 M9999-070111-C
Recommended Land Pattern
Micrel, Inc. MIC26950
July 2011 28 M9999-070111-C
Package Information
28-Pin 5mm x 6mm MLF® (YJL)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical impla
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
can nt
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