LM22670EVAL/NOPB - NATIONAL SEMICONDUCTOR

LM22670

October 22, 2010

3A SIMPLE SWITCHER®, Step-Down Voltage Regulator

with Synchronization or Adjustable Switching Frequency

General Description

The LM22670 series of regulators are monolithic integrated

circuits which provide all of the active functions for a step-

down (buck) switching regulator capable of driving up to 3A

loads with excellent line and load regulation characteristics.

High efficiency (>90%) is obtained through the use of a low

ON-resistance N-channel MOSFET. The series consists of a

fixed 5V output and an adjustable version.

The SIMPLE SWITCHER® concept provides for an easy to

use complete design using a minimum number of external

components and National’s WEBENCH® design tool.

National’s WEBENCH® tool includes features such as exter-

nal component calculation, electrical simulation, thermal sim-

ulation, and Build-It boards for easy design-in. The switching

clock frequency is provided by an internal fixed frequency os-

cillator which operates at 500 kHz. The switching frequency

can also be adjusted with an external resistor or synchronized

to an external clock up to 1MHz. It is also possible to self-

synchronize multiple regulators to share the same switching

frequency. The LM22670 series also has built in thermal shut-

down, current limiting and an enable control input that can

power down the regulator to a low 25 µA quiescent current

standby condition.

Features

■Wide input voltage range: 4.5V to 42V

■Internally compensated voltage mode control

■Stable with low ESR ceramic capacitors

■120 mΩ N-channel MOSFET TO-263 THIN package

■100 mΩ N-channel MOSFET PSOP-8 package

■Output voltage options:

-ADJ (outputs as low as 1.285V)

-5.0 (output fixed to 5V)

■±1.5% feedback reference accuracy

■Switching frequency of 500 kHz, adjustable between 200

kHz and 1 MHz

■-40°C to 125°C operating junction temperature range

■Precision enable pin

■Integrated boot diode

■Integrated soft-start

■Fully WEBENCH® enabled

■Step-down and inverting buck-boost applications

Package

■PSOP-8 (Exposed Pad)

■TO-263 THIN (Exposed Pad)

Applications

■Industrial Control

■Telecom and Datacom Systems

■Embedded Systems

■Automotive Telematics and Body Electronics

■Conversions from Standard 24V, 12V and 5V Input Rails

Simplified Application Schematic

30076001

LM22670 3A SIMPLE SWITCHER®, Step-Down Voltage Regulator with Synchronization or

Adjustable Switching Frequency

Connection Diagrams

30076040

8-Lead Plastic PSOP-8 Package

NS Package Number MRA08B

30076002

7-Lead Plastic TO-263 THIN Package

NS Package Number TJ7A

Ordering Information

Output Voltage Order Number Package Type NSC Package

Drawing Supplied As

ADJ LM22670MR-ADJ PSOP-8 Exposed Pad MRA08B 95 Units in Rails

ADJ LM22670MRE-ADJ 250 Units in Tape and Reel

ADJ LM22670MRX-ADJ 2500 Units in Tape and Reel

ADJ LM22670TJE-ADJ TO-263 THIN Exposed Pad TJ7A 250 Units in Tape and Reel

ADJ LM22670TJ-ADJ 1000 Units in Tape and Reel

5.0 LM22670MR-5.0 PSOP-8 Exposed Pad MRA08B 95 Units in Rails

5.0 LM22670MRE-5.0 250 Units in Tape and Reel

5.0 LM22670MRX-5.0 2500 Units in Tape and Reel

5.0 LM22670TJE-5.0 TO-263 THIN Exposed Pad TJ7A 250 Units in Tape and Reel

5.0 LM22670TJ-5.0 1000 Units in Tape and Reel

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LM22670

Pin Descriptions

Pin Numbers

PSOP-8

Package

Pin Numbers

TO-263 THIN

Package

Name Description Application Information

1 3 BOOT Bootstrap input Provides the gate voltage for the high side NFET.

2 - NC Not Connected Pin is not electrically connected inside the chip. Pin

does function as thermal conductor.

3 5 RT/

SYNC

Oscillator frequency adjust pin or

frequency synchronization

A resistor connected from this pin to GND adjusts the

oscillator frequency. This pin can also accept an input

for synchronization from an external clock. Pin can be

left floating and internal setting will be default to 500

kHz.

4 6 FB Feedback pin Inverting input to the internal voltage error amplifier.

5 7 EN Precision enable pin When pulled low regulator turns off.

6 4 GND System ground Provide good capacitive decoupling between VIN and

this pin.

7 2 VIN Source input voltage Input to the regulator. Operates from 4.5V to 42V.

8 1 SW Switch pin Attaches to the switch node.

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LM22670

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VIN to GND 43V

EN Pin Voltage -0.5V to 6V

RT/SYNC Pin Voltage -0.5V to 7V

SW to GND (Note 2) -5V to VIN

BOOT Pin Voltage VSW + 7V

FB Pin Voltage -0.5V to 7V

Power Dissipation Internally Limited

Junction Temperature 150°C

Soldering Information

Infrared (5 sec.) 260°C

ESD Rating (Note 3)

Human Body Model ±2 kV

Storage Temperature Range -65°C to +150°C

Operating Ratings (Note 1)

Supply Voltage (VIN)4.5V to 42V

Junction Temperature Range -40°C to +125°C

Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the

junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical

correlation. Typical values represent the most likely parametric norm at TA = TJ = 25°C, and are provided for reference purposes

only. Unless otherwise specified: VIN = 12V.

Symbol Parameter Conditions Min

(Note 5)

Typ

(Note 4)

Max

(Note 5)Units

LM22670-5.0

VFB Feedback Voltage VIN = 8V to 42V 4.925/4.9 5.0 5.075/5.1 V

LM22670-ADJ

VFB Feedback Voltage VIN = 4.7V to 42V 1.266/1.259 1.285 1.304/1.311 V

All Output Voltage Versions

IQQuiescent Current VFB = 5V 3.4 6mA

ISTDBY Standby Quiescent Current EN Pin = 0V 25 40 µA

ICL Current Limit 3.4/3.35 4.2 5.3/5.5 A

ILOutput Leakage Current VIN = 42V, EN Pin = 0V, VSW = 0V 0.2 2 µA

VSW = -1V 0.1 3 µA

RDS(ON) Switch On-Resistance TO-263 THIN Package 0.12 0.16/0.22 Ω

PSOP-8 Package 0.10 0.16/0.20

fOOscillator Frequency 400 500 600 kHz

TOFFMIN Minimum Off-time 100 200 300 ns

TONMIN Minimum On-time 100 ns

IBIAS Feedback Bias Current VFB = 1.3V (ADJ Version Only) 230 nA

VEN Enable Threshold Voltage Falling 1.3 1.6 1.9 V

VENHYST Enable Voltage Threshold

Hysteresis

0.6 V

IEN Enable Input Current EN Input = 0V 6 µA

FSYNC Synchronization Frequency VSYNC = 3.5V, 50% duty-cycle 1 MHz

VSYNC Synchronization Threshold

Voltage

1.75 V

TSD Thermal Shutdown

Threshold

150 °C

θJA Thermal Resistance TJ Package, Junction to ambient

temperature resistance (Note 6)

22 °C/W

θJA Thermal Resistance MR Package, Junction to ambient

temperature resistance (Note 7)

60 °C/W

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LM22670

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of device reliability

and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or other conditions beyond those indicated in

the recommended Operating Ratings is not implied. The recommended Operating Ratings indicate conditions at which the device is functional and should not be

operated beyond such conditions.

Note 2: The absolute maximum specification of the ‘SW to GND’ applies to DC voltage. An extended negative voltage limit of -10V applies to a pulse of up to 50

ns.

Note 3: ESD was applied using the human body model, a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.

Note 4: Typical values represent most likely parametric norms at the conditions specified and are not guaranteed.

Note 5: Min and Max limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlation using Statistical

Quality Control (SQC) methods. Limits are used to calculate National’s Average Outgoing Quality Level (AOQL).

Note 6: The value of θJA for the TO-263 THIN (TJ) package of 22°C/W is valid if package is mounted to 1 square inch of copper. The θJA value can range from

20 to 30°C/W depending on the amount of PCB copper dedicated to heat transfer. See application note AN-1797 for more information.

Note 7: The value of θJA for the PSOP-8 exposed pad (MR) package of 60°C/W is valid if package is mounted to 1 square inch of copper. The θJA value can

range from 42 to 115°C/W depending on the amount of PCB copper dedicated to heat transfer.

Typical Performance Characteristics Unless otherwise specified the following conditions apply: Vin =

12V, TJ = 25°C.

Efficiency vs IOUT and VIN

VOUT = 3.3V

30076027

Normalized Switching Frequency vs Temperature

30076004

Current Limit vs Temperature

30076003

Normalized RDS(ON) vs Temperature

30076008

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LM22670

Feedback Bias Current vs Temperature

30076005

Normalized Enable Threshold Voltage vs Temperature

30076010

Standby Quiescent Current vs Input Voltage

30076006

Normalized Feedback Voltage vs Temperature

30076007

Normalized Feedback Voltage vs Input Voltage

30076009

Switching Frequency vs RT/SYNC Resistor

30076013

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LM22670

Typical Application Circuit and Block Diagram

30076014

FIGURE 1. 3.3V VOUT, 3A at 600 kHz

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LM22670

Detailed Operating Description

The LM22670 switching regulator features all of the functions

necessary to implement an efficient high voltage buck regu-

lator using a minimum of external components. This easy to

use regulator integrates a 42V N-Channel switch with an out-

put current capability of 3A. The regulator control method is

based on voltage mode control with input voltage feed for-

ward. The loop compensation is integrated into the LM22670

so that no external compensation components need to be se-

lected or utilized. Voltage mode control offers short minimum

on-times allowing short duty-cycles necessary in high input

voltage applications. The operating frequency is fixed at

500kHz to allow for small external components while avoiding

excessive switching losses. The switching frequency can be

adjusted with an external resistor from 200 kHz to 1 MHz or

it can be synchronized to an external clock up to 1 MHz. The

output voltage can be set as low as 1.285V with the -ADJ

device. Fault protection features include current limiting, ther-

mal shutdown and remote shutdown capability. The device is

available in the TO-263 THIN and PSOP-8 packages featur-

ing an exposed pad to aid thermal dissipation.

The functional block diagram with typical application of the

LM22670 are shown in Figure 1.

The internal compensation of the -ADJ option of the LM22670

is optimized for output voltages up to 5V. If an output voltage

of 5V or higher is needed, the -5.0 fixed output voltage option

with an additional external resistive feedback voltage divider

may also be used.

Precision Enable

The precision enable pin (EN) can be used to shut down the

power supply. Connecting this pin to ground or to a voltage

less than typical 1.6V will completely turn off the regulator.

The current drain from the input supply when off is typically

25 µA with 12V input voltage. The power consumed during

this off state is mostly defined by an internal 2 MΩ resistor to

VIN. The enable pin has an internal pull-up current source of

approximately 6 µA. When driving the enable pin, the high

voltage level for the on condition should not exceed the 6V

absolute maximum limit. When enable control is not required,

the EN pin should be left floating. The precision feature en-

ables simple sequencing of multiple power supplies with a

resistor divider from another power supply.

The EN pin can also be used as an external UVLO to disable

the part when input voltage falls below a lower boundary of

operation. This is often used to prevent excessive battery dis-

charge. It can also be used to prevent early turn-on as Vin is

rising which can cause undesirable on-off toggling if Vin

droops below 4.5V during startup. Using EN as en external

UVLO is also recommended to prevent abnormal device op-

eration in applications where the input voltage falls below the

minimum operating voltage of 4.5V, during power down for

example.

Maximum Duty-Cycle / Dropout

Voltage

The typical maximum duty-cycle is 90% at 500 kHz switching

frequency. This corresponds to a typical minimum off-time of

200 ns. When operating at switching frequencies higher than

500 kHz, the 200 ns minimum off-time results in a lower max-

imum duty-cycle limit than 90%. This forced off-time is impor-

tant to provide enough time for the Cboot capacitor to charge

during each cycle.

The lowest input voltage required to maintain operation is:

Where VD is the forward voltage drop across the re-circulating

Schottky diode and VQ is the voltage drop across the internal

power N-FET of the LM22670. The RDS(ON) of the FET is

specified in the electrical characteristics section of this

datasheet to calculate VQ according to the FET current. F is

the switching frequency.

Minimum Duty-Cycle

Besides a minimum off-time, there is also a minimum on-time

which will take effect when the output voltage is adjusted very

low and the input voltage is very high. Should the operation

require an on-time shorter than minimum, individual switching

pulses will be skipped.

Pulse skipping is a normal mode of operation which appears

as a decrease in switching frequency. It has no effect on op-

eration or regulation except for an increase in output ripple

voltage. The pulse skipping function is required to maintain

proper regulation and overcurrent protection under the full

range of operating conditions.

The specified typical minimum on time of 100 ns is based on

the blanking time during current limit operation. During normal

operation, the minimum on-time will also include the effect of

propagation delay. Assume approximately 150 ns as a typical

operating minimum on time.

where D is the duty-cycle.

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LM22670

Current Limit

When the power switch turns on, the slight capacitance load-

ing of the Schottky diode, D1, causes a leading-edge current

spike with an extended ringing period. This spike can cause

the current limit comparator to trip prematurely. A leading

edge blanking time (TBLK) of 100 ns (typical) is used to avoid

sampling the spike.

When the switch current reaches the current limit threshold

the switch is immediately turned off. If TON is larger than the

minimum (100 ns typical) the switcher will hold the output

current flat at the set current limit value. But if TON is at or

decreases to the minimum TON (100 ns typical) the switching

frequency decreases to 1/5 the typical frequency. This effec-

tively causes the output current to fold back to a lower and

safe value. When the current limit condition is removed the

switching frequency is restored to nominal. This 5X frequency

fold back will result in a lower duty cycle pulse of the power

switch to minimize the overall fault condition power dissipa-

tion.

The percentage of output current limit fold back is affected by

duty cycle, inductance, and switching frequency.

See Figure 2 for details.

30076042

FIGURE 2. Output Current in Foldback vs.

Nominal Duty Cycle

The current limit will only protect the inductor from a runaway

condition if the LM22670 is operating in its safe operating

area. A runaway condition of the inductor is potentially catas-

trophic to the application. For every design, the safe operating

area needs to be calculated. Factors in determining the safe

operating area are the switching frequency, input voltage,

output voltage, minimum on-time and feedback voltage dur-

ing an over current condition.

As a first pass check, if the following equation holds true, a

given design is considered in a safe operating area and the

current limit will protect the circuit:

VIN x TBLK x F < VOUT x 0.724

If the equation above does not hold true, the following sec-

ondary equation will need to hold true to be in safe operating

area:

If both equations do not hold true, a particular design will not

have an effective current limit function which might damage

the circuit during startup, over current conditions, or steady

state over current and short circuit condition. Oftentimes a

reduction of the maximum input voltage or a reduction of the

switching frequency will bring a design into the safe operating

area.

When synchronized to an external sync pulse, the LM22670

will not fold back the switching frequency in an over current

condition.

Soft-Start

The soft-start feature allows the regulator to gradually reach

the initial steady state operating point, thus reducing start-up

stresses and surges. The soft-start is fixed to 500 µs (typical)

start-up time and cannot be modified.

Switching Frequency Setting and

Synchronization

There are three different modes for the RT/SYNC pin. It can

be left floating for a 500 kHz switching frequency.

A resistor from the RT/SYNC pin to ground can be used to

adjust the switching frequency between 200 kHz and 1 MHz.

An external synchronization pulse can be applied to the RT/

SYNC pin for switching frequencies up to 1 MHz. The

LM22670 internally sets the RT/SYNC mode during start up.

Many applications are limited to lower switching frequencies

due to minimum on-time and minimum off-time requirements

as well as increased switching losses at switching frequen-

cies higher than 500 kHz. In case of synchronizing to an

external sync pulse, the clock pulse must be in a valid low or

high state when the LM22670 starts up and can be applied

after device regulation. The internal oscillator will synchronize

to the external sync pulse rising edge. If the external sync

pulse is disconnected the LM22670 switching frequency will

return to default 500 kHz (typ.).

When operating at frequencies above 500 kHz, external com-

ponents such as the inductor and the output capacitors can-

not be reduced dramatically. This is due to the fixed internal

compensation and stability requirements. Running the

LM22670 at frequencies higher than 500 kHz is intended for

applications with EMI or beat frequency concerns. The flexi-

bility of increasing the switching frequency above 500 kHz can

be used to operate outside a critical signal frequency band of

an application. This can also be used to set multiple switchers

in an application to the same frequency to reduce beat fre-

quencies and simplify filtering.

See the current limit section of this datasheet for information

about the safe operating area. When synchronizing to an ex-

ternal synchronization pulse, the LM22670 will not fold back

the switching frequency in an over current condition.

The typical curve below shows adjusted switching frequen-

cies with different frequency set resistors from the RT/SYNC

pin to ground.

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LM22670

30076013

FIGURE 3. Switching Frequency vs RT/SYNC Resistor

Self Synchronize

It is also possible to self-synchronize multiple LM22670 reg-

ulators to share the same switching frequency. This can be

done by attaching the RT/SYNC pins together and putting a

1 kΩ resistor to ground. The diagram in Figure 4 illustrates

this setup. The two regulators will be clocked at the same fre-

quency but slightly phase shifted according to the minimum

off-time of the regulator with the fastest running oscillator. The

slight phase shift helps to reduce the stress on the input ca-

pacitors of the power supply.

30076030

FIGURE 4. Self Synchronizing Setup

Boot Pin

The LM22670 integrates an N-Channel FET switch and as-

sociated floating high voltage level shift / gate driver. This gate

driver circuit works in conjunction with an internal diode and

an external bootstrap capacitor. A 0.01 µF ceramic capacitor

connected with short traces between the BOOT pin and the

SW pin is recommended to effectively drive the internal FET

switch. During the off-time of the switch, the SW voltage is

approximately -0.5V and the external bootstrap capacitor is

charged from the internal supply through the internal boot-

strap diode. When operating with a high PWM duty-cycle, the

buck switch will be forced off each cycle to ensure that the

bootstrap capacitor is recharged. See the maximum duty-cy-

cle section for more details.

Thermal Protection

Internal Thermal Shutdown circuitry protects the LM22670 in

the event the maximum junction temperature is exceeded.

When activated, typically at 150°C, the regulator is forced into

a low power reset state. There is a typical hysteresis of 15

degrees.

Internal Compensation

The LM22670 has internal compensation designed for a sta-

ble loop with a wide range of external power stage compo-

nents.

Insuring stability of a design with a specific power stage (in-

ductor and output capacitor) can be tricky. The LM22670

stability can be verified over varying loads and input and out-

put voltages using WEBENCH® Designer online circuit sim-

ulation tool at www.national.com. A quick start spreadsheet

can also be downloaded from the online product folder.

The internal compensation of the -ADJ option of the LM22670

is optimized for output voltages below 5V. If an output voltage

of 5V or higher is needed, the -5.0 option with an additional

external resistor divider may also be used.

The typical location of the internal compensation poles and

zeros as well as the DC gain is given in Table 1. The LM22670

has internal type III compensation allowing for the use of most

output capacitors including ceramics.

This information can be used to calculate the transfer function

from the FB pin to the internal compensation node (input to

the PWM comparator in the block diagram).

TABLE 1.

Corners Frequency

Pole 1 150 kHz

Pole 2 250 kHz

Pole 3 100 Hz

Zero 1 1.5 kHz

Zero 2 15 kHz

DC gain 37.5 dB

For the power stage transfer function the standard voltage

mode formulas for the double pole and the ESR zero apply:

The peak ramp level of the oscillator signal feeding into the

PWM comparator is VIN/10 which equals a gain of 20dB of

this modulator stage of the IC. The -5.0 fixed output voltage

option has twice the gain of the compensation transfer func-

tion compared to the -ADJ option which is 43.5 dB instead of

37.5 dB.

Generally, calculation as well as simulation can only aid in

selecting good power stage components. A good design prac-

tice is to test for stability with load transient tests or loop

measurement tests. Application note AN-1889 shows how to

easily perform a loop transfer function measurement with only

an oscilloscope and a function generator.

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LM22670

Application Information

EXTERNAL COMPONENTS

The following design procedures can be used to design a non-

synchronous buck converter with the LM22670.

Inductor

The inductor value is determined based on the load current,

ripple current, and the minimum and maximum input voltage.

To keep the application in continuous current conduction

mode (CCM), the maximum ripple current, IRIPPLE , should be

less than twice the minimum load current.

The general rule of keeping the inductor current peak-to-peak

ripple around 30% of the nominal output current is a good

compromise between excessive output voltage ripple and ex-

cessive component size and cost. When selecting the induc-

tor ripple current ensure that the peak current is below the

minimum current limit as given in the Electrical Characteris-

tics section. Using this value of ripple current, the value of

inductor, L, is calculated using the following formula:

where F is the switching frequency which is 500 kHz without

an external frequency set resistor or external sync signal ap-

plied to the RT/SYNC pin. If the switching frequency is set

higher than 500kHz, the inductance value may not be reduced

accordingly due to stability requirements. The internal com-

pensation is optimized for circuits with a 500 kHz switching

frequency. See the internal compensation section for more

details. This procedure provides a guide to select the value of

the inductor L. The nearest standard value will then be used

in the circuit.

Increasing the inductance will generally slow down the tran-

sient response but reduce the output voltage ripple amplitude.

Reducing the inductance will generally improve the transient

response but increase the output voltage ripple.

The inductor must be rated for the peak current, IPK+, to pre-

vent saturation. During normal loading conditions, the peak

current occurs at maximum load current plus maximum ripple.

Under an overload condition as well as during load transients,

the peak current is limited to 4.2A typical (5.5A maximum).

This requires that the inductor be selected such that it can run

at the maximum current limit and not only the steady state

current.

Depending on inductor manufacturer, the saturation rating is

defined as the current necessary for the inductance to reduce

by 30% at 20°C. In typical designs the inductor will run at

higher temperatures. If the inductor is not rated for enough

current, it might saturate and due to the propagation delay of

the current limit circuitry, the power supply may get damaged.

Input Capacitor

Good quality input capacitors are necessary to limit the ripple

voltage at the VIN pin while supplying most of the switch cur-

rent during on-time. When the switch turns on, the current into

the VIN pin steps to the peak value, then drops to zero at turn-

off. The average current into VIN during switch on-time is the

load current. The input capacitance should be selected for

RMS current, IRMS, and minimum ripple voltage. A good ap-

proximation for the required ripple current rating necessary is

IRMS > IOUT / 2.

Quality ceramic capacitors with a low ESR should be selected

for the input filter. To allow for capacitor tolerances and volt-

age effects, multiple capacitors may be used in parallel. If step

input voltage transients are expected near the maximum rat-

ing of the LM22670, a careful evaluation of ringing and pos-

sible voltage spikes at the VIN pin should be completed. An

additional damping network or input voltage clamp may be

required in these cases.

Usually putting a higher ESR electrolytic input capacitor in

parallel to the low ESR bypass capacitor will help to reduce

excessive voltages during a line transient and will also move

the resonance frequency of the input filter away from the reg-

ulator bandwidth.

Output Capacitor

The output capacitor can limit the output ripple voltage and

provide a source of charge for transient loading conditions.

Multiple capacitors can be placed in parallel. Very low ESR

capacitors such as ceramic capacitors reduce the output rip-

ple voltage and noise spikes, while higher value capacitors in

parallel provide large bulk capacitance for transient loading

and unloading. Therefore, a combination of parallel capaci-

tors, a single low ESR SP or Poscap capacitor, or a high value

of ceramic capacitor provides the best overall performance.

Output capacitor selection depends on application conditions

as well as ripple and transient requirements. Typically a value

of at least 100 µF is recommended. An approximation for the

output voltage ripple is:

In applications with Vout less than 3.3V, where input voltage

may fall below the operating minimum of 4.5V, it is critical that

low ESR output capacitors are selected. This will limit poten-

tial output voltage overshoots as the input voltage falls below

device normal operation range.

If the switching frequency is set higher than 500 kHz, the ca-

pacitance value may not be reduced accordingly due to sta-

bility requirements. The internal compensation is optimized

for circuits with a 500 kHz switching frequency. See the in-

ternal compensation section for more details.

Cboot Capacitor

The bootstrap capacitor between the BOOT pin and the SW

pin supplies the gate current to turn on the N-channel MOS-

FET. The recommended value of this capacitor is 10 nF and

should be a good quality, low ESR ceramic capacitor.

It is possible to put a small resistor in series with the Cboot

capacitor to slow down the turn-on transition time of the in-

ternal N-channel MOSFET. Resistors in the range of 10Ω to

50Ω can slow down the transition time. This can reduce EMI

of a switched mode power supply circuit. Using such a series

resistor is not recommended for every design since it will in-

crease the switching losses of the application and makes

thermal considerations more challenging.

Resistor Divider

For the -5.0 option no resistor divider is required for 5V output

voltage. The output voltage should be directly connected to

the FB pin. Output voltages above 5V can use the -5.0 option

with a resistor divider as an alternative to the -ADJ option.

This may offer improved loop bandwidth in some applications.

See the Internal Compensation section for more details.

For the -ADJ option no resistor divider is required for 1.285V

output voltage. The output voltage should be directly con-

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LM22670

nected to the FB pin. Other output voltages can use the -ADJ

option with a resistor divider.

The resistor values can be determined by the following equa-

tions:

-ADJ option:

-5.0 option:

Where VFB = 1.285V typical for the -ADJ option and 5V for the

-5.0 option

30076023

FIGURE 5. Resistive Feedback Divider

A maximum value of 10 kΩ is recommended for the sum of

R1 and R2 to keep high output voltage accuracy for the –ADJ

option. A maximum of 2 kΩ is recommended for the -5.0 out-

put voltage option. For the 5V fixed output voltage option, the

total internal divider resistance is typically 9.93 kΩ.

At loads less than 5 mA, the boot capacitor will not hold

enough charge to power the internal high side driver. The

output voltage may droop until the boot capacitor is

recharged. Selecting a total feedback resistance to be below

3 kΩ will provide some minimal load and can keep the output

voltage from collapsing in such low load conditions.

Catch Diode

A Schottky type re-circulating diode is required for all

LM22670 applications. Ultra-fast diodes which are not Schot-

tky diodes are not recommended and may result in damage

to the IC due to reverse recovery current transients. The near

ideal reverse recovery characteristics and low forward volt-

age drop of Schottky diodes are particularly important diode

characteristics for high input voltage and low output voltage

applications common to the LM22670. The reverse recovery

characteristic determines how long the current surge lasts

each cycle when the N-channel MOSFET is turned on. The

reverse recovery characteristics of Schottky diodes mini-

mizes the peak instantaneous power in the switch occurring

during turn-on for each cycle. The resulting switching losses

are significantly reduced when using a Schottky diode. The

reverse breakdown rating should be selected for the maxi-

mum VIN, plus some safety margin. A rule of thumb is to select

a diode with the reverse voltage rating of 1.3 times the max-

imum input voltage.

The forward voltage drop has a significant impact on the con-

version efficiency, especially for applications with a low output

voltage. ‘Rated’ current for diodes varies widely from various

manufacturers. The worst case is to assume a short circuit

load condition. In this case the diode will carry the output cur-

rent almost continuously. For the LM22670 this current can

be as high as 4.2A (typical). Assuming a worst case 1V drop

across the diode, the maximum diode power dissipation can

be as high as 4.2W.

Circuit Board Layout

Board layout is critical for switching power supplies. First, the

ground plane area must be sufficient for thermal dissipation

purposes. Second, appropriate guidelines must be followed

to reduce the effects of switching noise. Switch mode con-

verters are very fast switching devices. In such devices, the

rapid increase of input current combined with the parasitic

trace inductance generates unwanted L di/dt noise spikes.

The magnitude of this noise tends to increase as the output

current increases. This parasitic spike noise may turn into

electromagnetic interference (EMI) and can also cause prob-

lems in device performance. Therefore, care must be taken

in layout to minimize the effect of this switching noise.

The most important layout rule is to keep the AC current loops

as small as possible. Figure 6 shows the current flow of a buck

converter. The top schematic shows a dotted line which rep-

resents the current flow during the FET switch on-state. The

middle schematic shows the current flow during the FET

switch off-state.

The bottom schematic shows the currents referred to as AC

currents. These AC currents are the most critical since current

is changing in very short time periods. The dotted lines of the

bottom schematic are the traces to keep as short as possible.

This will also yield a small loop area reducing the loop induc-

tance. To avoid functional problems due to layout, review the

PCB layout example. Providing 3A of output current in a very

low thermal resistance package such as the TO-263 THIN is

challenging considering the trace inductances involved. Best

results are achieved if the placement of the LM22670, the by-

pass capacitor, the Schottky diode and the inductor are

placed as shown in the example. It is also recommended to

use 2oz copper boards or thicker to help thermal dissipation

and to reduce the parasitic inductances of board traces.

It is very important to ensure that the exposed DAP on the

TO-263 THIN package is soldered to the ground area of the

PCB to reduce the AC trace length between the bypass ca-

pacitor ground and the ground connection to the LM22670.

Not soldering the DAP to the board may result in erroneous

operation due to excessive noise on the board.

www.national.com 12

LM22670

30076024

FIGURE 6. Current Flow in a Buck Application

Thermal Considerations

The two highest power dissipating components are the re-

circulating diode and the LM22670 regulator IC. The easiest

method to determine the power dissipation within the

LM22670 is to measure the total conversion losses (Pin –

Pout) then subtract the power losses in the Schottky diode

and output inductor. An approximation for the Schottky diode

loss is:

P = (1 - D) x IOUT x VD

An approximation for the output inductor power is:

P = IOUT2 x R x 1.1,

where R is the DC resistance of the inductor and the 1.1 factor

is an approximation for the AC losses. The regulator has an

exposed thermal pad to aid power dissipation. Adding several

vias under the device to the ground plane will greatly reduce

the regulator junction temperature. Selecting a diode with an

exposed pad will aid the power dissipation of the diode. The

most significant variables that affect the power dissipated by

the LM22670 are the output current, input voltage and oper-

ating frequency. The power dissipated while operating near

the maximum output current and maximum input voltage can

be appreciable. The junction-to-ambient thermal resistance of

the LM22670 will vary with the application. The most signifi-

cant variables are the area of copper in the PC board, the

number of vias under the IC exposed pad and the amount of

forced air cooling provided. The integrity of the solder con-

nection from the IC exposed pad to the PC board is critical.

Excessive voids will greatly diminish the thermal dissipation

capacity. The junction-to-ambient thermal resistance of the

LM22670 TO-263 THIN and PSOP-8 packages are specified

in the Electrical Characteristics table under the applicable

conditions. For more information regarding the TO-263 THIN

package, refer to Application Note AN-1797 at

www.national.com.

13 www.national.com

LM22670

PCB Layout Example for TO-263 THIN Package

30076025

www.national.com 14

LM22670

PCB Layout Example for PSOP-8 Package

30076041

15 www.national.com

LM22670

Schematic for Buck/Boost (Inverting) Application

See AN-1888 for more information on the inverting (buck-boost) application generating a negative output voltage from a positive

input voltage.

30076026

www.national.com 16

LM22670

Physical Dimensions inches (millimeters) unless otherwise noted

7-Lead Plastic TO-263 THIN Package

NS Package Number TJ7A

8-Lead PSOP Package

NS Package Number MRA08B

17 www.national.com

LM22670

Notes

LM22670 3A SIMPLE SWITCHER®, Step-Down Voltage Regulator with Synchronization or

Adjustable Switching Frequency

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